The 7th International Conference on Power ElectronicsOctober 22-26, 2007 / Exeo, Daegu, Korea

A Half Bridge Flyback Converter with ZVS and ZCS Operations

WEC3·1

Li-MingWuChing Yun University

Department of Electrical Engineering229, Chien-Hsin Rd., Jung-Li, Taiwan 320, R.O.C.

Email: lmwu@cvu.edu.tw

Abstract-An asymmetrical half bridge flyback converterwithout output inductor is presented in this paper, in which thepower switches can be turned on with ZVS and the outputdiode can be turned on and off with ZCS. While the ZVS isobtained from the magnetizing current, the ZCS is achievedfrom the resonant current. The ZVS and ZCS conditions canbe maintained for full load range. The characteristics of theproposed converter rely on the utilization ofresonant inductorand blocking capacitor as resonant energy delivery devicesrather than linearly charging and discharging components.The operational principle and circuit analysis of the proposedconverter is described in this paper. The experimental test of a90W prototype converter is performed and the results arepresented.

I. INTRODUCTION

For reasons of simple structure and robust characteristics,the flyback converter has been one of the most widely useddc-dc power supplies in the small power areas. Themultiple-output and cost effective design makes the flybackconverter attractive for applications such as personalcomputers, assorted home appliances, and various officeequipments, etc. Major disadvantage of flyback converterattributes to the hard switching operation. In general, the hardswitching operation of the power switch results in a highswitching loss, high EMI noise, and high switch voltage stress.To overcome these drawbacks, various kinds ofsoft switchingtechniques have been proposed [1----9]. Among them, theactive-clamp converters [3----5] make use of simple activeclamp network to achieve ZVS operation of the powerswitches. However, the power switch still suffers from highvoltage stresses since the sum of input voltage and thetransformer primary voltage is imposed on them. Therefore,the asymmetrical half bridge flyback converters [6----9], whichutilizes the half bridge structure and a blocking capacitor toachieve ZVS operation of the power switches as well as lowswitch voltage stresses, are gaining popularity. From previousworks [7,8], it had been shown that, by reducing themagnetizing inductor and allowing duty cycle constraintbeyond 50%, the half bridge flyback converter can achieveZVS operation for power switches from no load to full loadconditions. However, the rectifier suffers from high di/dt attum off, which results in significant energy loss from highreverse recovery current.

978-1-4244-1872-5/08/$25.00 © 2008 IEEE

Chen-Yin PongChing Yun University

Department of Electrical Engineering229, Chien-Hsin Rd., Jung-Li, Taiwan 320, R.O.C.

Email: m9512014@cyu.edu.tw

Another research [9] used the same power circuit as in [8] butadded an inductor as output filter to obtain the samesecondary diode current waveforms as the conventionalforward converter does. The output rectifier is complex andhas no soft switching operations as well.

In this research, a half bridge flyback converter withoutusing output inductor is studied, in which both the powerswitches and output power diode can achieve soft switchingoperations. The voltage stress ofpower switches is not greaterthan the input voltage. The power switches can be turned onwith ZVS and the output diode can be turned on and off withZCS. The ZVS and ZCS conditions can be maintained for thefull spectrum of load range. The characteristics of theproposed converter rely on the utilization ofresonant inductorand blocking capacitor as resonant energy delivery devicesrather than a pair of linearly charging and dischargingcomponents.

The operational principles, circuit analysis, and designequations of the proposed converter are described in thispaper. The experimental test of a 90W prototype converter isperformed and the test results are presented to confirm thevalidity of the proposed converter.

II. OPERATIONAL PRINCIPLES

Figure 1 shows the simplified circuit diagram of the proposedhalfbridge flyback converter. The resonant inductor Lk consists oftotal leakage inductance of transfonner Tr and external strayinductances. The ESRs in passive components and internalresistance of semiconductors are not included in the diagram forsimplification. Fig. 2 shows the eight operating modes of theconverter during a switching cycle, and Fig. 3 illustrates the keywaveforms in time intervals corresponding to each of the eightmodes. Prior to describing the basic operational principles, severalassumptions are made first:

• The converter is operating in steady state.

• The resonant inductor Lk is much less than the magnetizinginductor Lm.

• The resonant period ofCb and Lk is comparable with the off-timeofswitch Ql.

• In mode 2 and 6, the circuit is assumed to behave linearly.

• In mode 3, the resonant current is asswned to be the same as themagnetizing current.

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The 7th International Conference on Power ElectronicsOctober 22-26, 2007 I Exeo, Daegu, Korea

Fig. 1Simplified schematic ofthe halfbridge flyback converter

(~) loa. 2(&)loa.1

n

oT

lr

lq30'Do + ++a RL VoL.---_---+- >___-

+ + o:JVc Cb

~ Vin

(clIoae3 (a)lIo4t4

(1 )

To proceed, the operational analysis of the asymmetrical halfbridge flyback converter can be explained as follows.

Mode 1(to"'41): At to , Q1 is on and Q2 is off The output diode Dois reversely biased. Components Cb, Lm and Lk fonn a seriesresonant tank with voltage source V;n providing the input energy.Since the length of time interval (Irlo) is short compared with thetime constant ofthe resonant~ the power source ~':n charges Cb,

Lm and Lk in a linear fashion The leakage and magnetizing currentsstart to decrease from zero. The circuit governing equations are:

di(Lm+Lk)~=-T/~n +VC

dt

(,)Ioa.,

Fig. 2 Halfbridge flyback converter topological states

(3)

(2)

This mode ends when Q1 turns off

Mode 2 (t['--12): At I]. Q1 is turned off and Q2 remains off Thecomponents Cb, Lm, Lkand Cds1 fonn a new series resonant network,and Cb, Lm, Lk and Cds2 fonn another series resonant network Thepower source Vjn charges Cds1 and Cb with halfof the magnetizingcurrent. The other half of the magnetizing current charges Cb

through discharging Cds~. The circuit governing equations are:

diL(Lm +Lk)_k =-V;n +vc +Vdsl

dtiLm

(4)

dvds2 1 . 1 .C -----1 ---Ids2 dt - 2 Lk - 2 Lm

(5)

(6)

ilk

By assuming a linear magnetizing current, the voltage vds1 starts toincreases from zero in a quadratic fashion This mode ends whenVds1 reaches to the value of(Vin + Jljd) and the body diode ofQ2 startsto conduct. The notation vftstands for diode fOlWard voltage drop.

Mode 3 (/1"t3): In this period, both Q1 and Q2 are remaining offThe body diode of Q2 starts to conduct and the voltage across themagnetizing inductor changes polarity. By assuming the resonantcurrent the same as the magnetizing curren~ the governingequations can be written as:

Vol

: . , '

iDo """'1 /~iI~'_/)_' ~/"""I~\__""'I_ tl a a ~ ~ ~ fl ~

Fig. 3 Steady state wavetonns ofthe halfbridge flyback converterwith ZVS and ZCS operations

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The 7th International Conference on Power ElectronicsOctober 22-26, 2007 / EXCO, Daegu, Korea

(7) (18)

(8)This IOOde ends when vckJ drops to -Jtfd and the body diode of QJstarts to conduct.

Since the difference between the resonant current and themagnetizing current is zero at this stage, the output diode is notconducting and no voltage is reflected from the secnodaIy to theprimary side of transfonner. This IOOde ends when Q2 is ZVSturned on and the dead time is over.

(20)

(19)

dvc . .C b -=-IL =-IL

dt k m

(1) Voltage transfer ratio

In steady state, the voltage transfer ratio depends on dutyratio D and turns ratio n of transformer Tr• It can be obtainedusing the flux balance principle in Lm and Lk• The equationsdescribing these relationships are:

kl(f';n - Vc)D - nVo(l- D) = 0 (21)

k2(~n - Vc ) - (Vc - nVo)(1- D) = 0 (22)

where k l = Lm/(Lm+Lk), and k2 =Lk/(Lm+Lk)·

Solving (21) and (22) simultaneously, the voltage transferratio is obtained as follows:

Vo = kiD ~ D (23)~n n n

III. CIRCUIT ANALYSIS

Mode 8 (tlto): At t7, QJ turns on under ZVS. The magnetizingcurrent keeps decreasing in a linear fashion and Cb dischargesenergy back to the voltage SOlU"Ce Vine This mode ends when themagnetizing current decreases to zero. Then the input power SOlU"Cebegins to charge Cb, Lm and Lk again, and another switching cyclerepeats.

This mode ends when the dead time is over and QJ is ZVS turnedon.

In this section, key characteristics of the studied converter areanalyzed based on the above operational modes. However, tosimplify the analysis, the dead time between QJ and Q2 and theswitch output capacitances have been neglected. The simplificationshould not affect the truth of the analysis. Fig. 4 shows thesimplified waveforms of the resonant inductor current, themagnetizing inductor current, the voltage of blocking capacitor,and the output diode current.

Mode 7 (to-47): In this mode, both QJ and Q2are still off At t6, thevoltage across the magnetizing inductor changes polarity. The bodydiode of QJ starts to conduct and the magnetizing current starts todecrease. The governing equations are:

(14)

(13)

(15)

diL (9)Lm-m =nVodt

diL (10)Lk _k =Vc -nVodt

dvc . (11 )Cb-=ILdt *

iDo = n(iLk-iLm ) (12)

Mode 5 (t4 t5): In this IOOde, states of Q2 and QI remains thesame as in mode 4. The resonant network stops resonance becausethe resonant current can not be smaller than the magnetizing current.The output diode is fOlWardly biased with zero current conducted.The governing equations become:

Mode 4 (trt4): In this period, Q2 is on and QI is off The outputdiode is conducting and the converter output voltage is reflected tothe transfonner primary side. The resonant tank fonned by Cb, Lm

and Lk starts to develop a resonant current which is different fromthe magnetizing current. It is the difference between the resonantcurrent and the magnetizing current that detennines the outputdiode current. This mode ends when the resonant current meets themagnetizing current and the output diode turns offwith zero current.The governing equations in this IOOde include:

This mode ends when Q2 is turned off

Mode 6 (t)t6): In this period, QI and Q2are both off At t5, halfof the magnetizing current starts to charge Cck2 from zero voltage,the other half discharges CckJ from voltage V;n. The governingequations are:

dvc . .Cb -=-IL =-IL

dt k m

(16)

(17)

(2) Average voltage across blocking capacitor

The average voltage Vc across the blocking capacitor isproportional to the duty ratio D. The relationship can beobtained from solving (21) and (22) simultaneously:

Vc = D~n (24)

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The 7th International Conference on Power ElectronicsOctober 22-26, 2007 I Exeo, Daegu, Korea

(30)

(31)

Vc Cb ~ischar9ing Cb charging~

Utilizing the charge balance ofCb over one switching cycleyields:

r'iLk (t)dt =0 =>

~[sin(m(1-D)Ts -~)+sinljJ]= ~n(1-D)D2T& i",u DmJ: 2(L", + Lk )

Bringing (30) into (26), one obtains

10 =-i + V;nD(1- D)r:n mu 2(L

m+ L

k)

Substituting (31) into (25) gives the simple relationshipbetween /m and /0:

/ =_ 10

mn

(32)

That is," nV

d cos(m,(I- D)Ts -¢J) ~ 1mL +_0 (1- D)Ts (36)Lm

Since parameter d is a function of ()Jr, (36) is highly nonlinearin ()Jr. Analytically, it is difficult to solve (36) for ()Jr. To findthe resonant frequency, the following empirical formula canbe used:

(5) Resonantfrequency

During the time interval DTs ~ t ~ Ts

' the resonant current

hk(t) is a sinusoidal function and the magnetizing current hm(t)increases linearly. The difference between the resonantcurrent and the magnetizing current generates the outputdiode current. Under full load condition, the output diodeturns offwith ZCS only if the following criterion is satisfied:

iLk (J:) ~ i Lm (J:) (33)

(37)

(34)

(35)

iLk (t) =d cos(mr ( t - Dr: ) - ¢J )

" nViL (I) = ImL +_0 (I - Dr:)

m L",

where

1tm >---

r (1- D)J:

Satisfaction of (37) means that the resonant current willcontinue to develop for at least half of its resonant periodduring the time interval O--(l-D)Ts in order to have ZCS tumoff for the output diode. It is also noted that once the outputdiode turns off with ZCS under full load condition, the ZCSwill continue to effect for light load because satisfaction ofcriterion (33) is guaranteed.

(26)

iDo 0,--------,-----:::,1~iv\----'-----::~~, , ,

(4) Average load current

The output diode current iDo(t) is determined by the turnsratio n of Tr and the difference between the resonant currentirk and the magnetizing current hm. The average load current/0 can be obtained as follows:

1 !l-D)T,1

0=- n(iL (I) - iL (/»dlT

sk",

={~sin(mr (1- D)Ts -ljJ) + sinljJ - j"'L (1- D) - nVo (1- D)2 T]~~ 2~

(3) Average magnetizing current

Referring to Fig. 4, the average magnetizing current /m canbe derived as follows:

1 =~ (8 i (I)dt =i _~nD(1- D)T& (25)", Ts.b Lm mu 2(L", + L

k)

where imu

is the upper peak current ofmagnetizing inductor.

Fig. 4 Simplified waveforms of inductor currents, outputdiode current, and voltage across blocking capacitor.

Equation (23) and (24) reveal that, depends on V;n and Vo,the value of duty ratio D can be larger or less than 0.5. Forfixed Vim the smaller the Vmthe shorter the duty ratio.

where

d2 =i../ + [(nvo -i(DT')f Z=l~b ' (27)

m, =1/~LkCb ' (2

8)

ljJ=tan-1(nJ.: -~c(DJ:»+7r . (29)ZI",L

and imL is the lower peak current ofmagnetizing inductor.

(6) ZVS conditions

The ZVS operation of power switches in the studiedconverter depends on the magnitude of magnetizing current.The ZVS conditions of Q2 can be easily reached because themagnetizing current at time t1 is at its lower peak value imL,

which is large enough in magnitude to fully discharge theenergy stored in the output capacitor Cds2• According to (31),1mL can be obtained as follows:

i - _V;n(1- D)DJ: 10 (38)mL - 2(L", + L

k) n

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(42)

(45)

The 7th International Conference on Power ElectronicsOctober 22-26, 2007 I EXCO, Daegu, Korea

and the ZVS condition for Q2 is

1 "2 1 2 (39)"2LmlmL ~ "2Cds2~n .

The discharged energy from Cds2 is charged back to theblocking capacitor Cb• The dead time between QJ and Q2 mustbe long enough to allow Cds2 fully discharged.

The ZVS conditions of QJ depends on the upper peakcurrent imu of the magnetizing inductor. This peak currentmust be positive and large enough in magnitude to fullydischarge the energy stored in the output capacitor CdsJ '

Repeating (31), one is requiring

j = ~nD(l- D)'rs 10 ~ 0 ' (40)mu 2(Lm+L

k) n

which leads to

(41)

Inequality constraint (41) must be satisfied for QJ to haveZVS. Another necessary condition for QJ to have ZVS is

1 "2 1 2"2Lmlmu ~ "2 Cdst V;n

It should be noted that the time used to discharge CdsJ

(mode 6) is longer than that of Cds2 (mode 2). This is becausethe upper peak current imu of the magnetizing inductor issmaller in magnitude compared with that of the lower peakcurrent imL• The minimum dead time between QJ and Qb

therefore, should be determined by the time required to fullydischarge CdsJ .

(IV) DESIGN CONSIDERATIONTo validate the characteristics of the studied converter, a

prototype converter has been designed based on the followingspecifications:

• Input voltage Vin: 156V DC• Output voltage Vo: 12V• Maximum output power Pmax: 90W• Switching frequency Is: 50kHz

(1) Duty ratioThe duty ratio D can be determined according to (23).

Depending on the transformer turns ratio n, the value ofD canexceed 0.5 when voltage mode control is applied. The largerthe D value, the more energy stored in Lm, Lk and Cb, whichimplies a bigger input capacitor with longer hold-up time isrequired. On the other hand, the voltage stress on output diodeis smaller when the duty ratio is larger.(2) Transformer turns ratio

The transformer turns ratio n is related to duty ratio D andmagnetizing inductance Lm through (23) and (41). A largervalue of n reduces the current on the primary side oftransformer and increases the current on the secondary side.This is beneficial to the reduction of conduction losses on theprimary side. The drawback is a higher current stress on therectifier. A compromise between selection of n, D and Lm arerecommended.(3) Transformer magnetizing inductor

The design formula used to determine the magnetizinginductor for conventional flyback converter operating incontinuous conduction mode can be referred here. It isdescribed as

L ~ [B X 10-8 - V;n D(1- DT.s]x N A ~ (43)m sat 2N

pA

ep e 1

0

where Bsat is the saturated flux density ofthe transformer core,Np is the number of turns of primary winding, and Ae is theeffective cross section area ofthe transformer core. However,Eq. (43) does not guarantee the satisfaction ofZVS conditions.It is recommended that, after using (43), Eq. (40) and (42)should be checked to ensure the existence of ZVS operationfor switch QJ. A preliminary checking formula for ZVS is(41).(4) Resonant inductor

The resonant inductor Lk is mainly used to develop aresonant current during the period energy being transferredfrom the primary to the secondary side of the transformer. Toensure the ZCS operation of the output diode, the resonantfrequency wr=t/~LkCh can be estimated from satisfying Eq.

(37). Since the transformer leakage inductor is the major partofthe resonant inductor, and the solution of(37) usually leadsto a higher frequency compared with switching frequency, theleakage inductor should be determined as small as possible. Ithas been noted that the resonant inductor Lk is closely relatedto the ZCS operation of the rectifier and has nothing to dowith the ZVS operations of the power switches.(5) Blocking capacitor

Depending onLh the block capacitor Cb can be determinedby

1Cb ~ 2

(J(44)

Lt (1-~)T,Limiting the voltage ripple of the blocking capacitor smallerthan the average voltage Vc generates another constraint forthe determination of Cb :

Cb 2(~)2Lt,DVin

where parameter d comes from (27).

IV. EXPERIMENTAL RESULTS

Based on the design consideration in the preceding section,a prototype converter operated at frequency of 50kHz hasbeen constructed. Table 1 lists the components that are used inthe prototype of the studied converter. Fig. 5 shows the keywaveforms ofa simulated converter at full load condition. Fig.6 shows the key waveforms measured from the prototypeconverter under full load condition. The resonant inductorcurrent hk(t) and the output diode current ido(t) in Fig. 6 are ingood agreement with the theoretical and simulation analysisin Fig. 3 and Fig. 5, respectively. The ZVS operations of QJ

and Q2 can be observed from zero to full load, as indicated inthe waveforms shown in Fig. 7.

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The 7th International Conference on Power ElectronicsOctober 22-26, 2007 / EXCO, Daegu, Korea

Vgs1 (12V)

(a) QI at zero load

Table 1. Components used in the prototype converterMagnetizin~ inductor Lm O.3mH

Resonant inductor Lk 2.58JlHTurns ratio 7:1

Blocking capacitor Cb O.68JlF1250VFilter capacitor Cr 4700JlF/35V

Power switch Ql & Q2 K2843x2Power diode Do MOSPEC S30C40C

Switching frequency Is 50kHz

Load R/oad 1.6 nDriver ICs UC3843+IR2III

Opto-coupler 4N25

iLk(2.5A)

Vgs2 (l2V)

(b) QI at full load

(C) {!2 at zero load

(d) Q2 at full load

Fig. 7 Experimental waveforms of ZVS operations

resonant current. The ZVS and ZCS conditions can bemaintained for the full load range. The operational principleshave been presented in the mode analysis, and the designequations have been derived in steady state analysis. The

iDo(10A)

Vgs1(5V)

Fig. 6 Key waveforms from experiments

Fig. 5 Key waveforms from simulation

fl~ r---:-----------------

~o~ l::====:..........;.......;.......;.......;.--=:======~.......J

V. CONCLUSION

This report has presented the analysis and experimentalresults of an asymmetrical half bridge flyback converterwithout output inductor. The power switches can be turned onwith ZVS and the output diode can be turned on and off withZCS. It has been found that the ZVS operation of the powerswitches is obtained from the magnetizing current, the ZCSoperation of the output power diode is achieved from the

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The 7th International Conference on Power ElectronicsOctober 22-26, 2007 / Exeo, Daegu, Korea

necessary conditions for ZVS and ZCS operations have beenobtained as well. Based on the design equations, a 90Wprototype converter is constructed and tested by experiment.The experimental results of the prototype converter haveshown the characteristics that are in good agreement withsimulation and theoretical analysis.

ACKNOWLEDGMENT

This work was supported under the contract number NSC95-2221-E-231-043 by the National Science Council, Taiwan,R.O.C.

REFERENCES

[1] A. Aqik and 1. Cadirci, "Active clamped ZVS forward converter withsoft-switched synchronous rectifier for high efficiency, low outputvoltage applications," lEE Proc. EIectr. Power Appl., Vol. 150, No.2,March 2003.

[2] Yilei Gu, Zhengyu Lu, etc., "A Novel ZVS Resonant Reset DualSwitch Forward DC-DC Converter," IEEE Transactions on PowerElectronics, Vol. 22, No.1, January 2007.

[3] Robert Watson. Fred C. Lee, Guichao C. Hua, "Utilization of anActive-Clamp Circuit to Achieve Soft Switching in FlybackConverters," IEEE Trans. on Power Electronics, Vol. 11, No. I, 1996,pp.162-169.

[4] 11, H.K., and KIM, H.J., "Active clamp forward converter withMOSFET synchronous rectification," IEEE Power Electron. Spec.Conf. Rec., 1994, pp. 895-901.

[5] WATSON, R., LEE, F.C., and HUA, G.C.: 'Utilization of an activeclamp circuit to achieve soft switching in flyback converters," IEEETran. On Power Electron., 1996,11, (I), pp. 162-169.

[6] YOSHIDA, K., ISHII, T., and NAGAGATA, N.: "Zero voltageswitching approach for flyback converter," Proceedings of IEEEINTELEC, 1992, pp. 324-329.

[7] SEO, D.H., LEE, OJ., LIM, S.H., and PARK, 1.S.: 'AsymmetricalPWM flyback converter," Proceedings of IEEE PESC, 2000, pp.848-852.

[8] T. M. Chen, C. L. Chen, "Analysis and Design of Asymmetrical HalfBridge Flyback Converter," Electric Power Applications, lEE Proc.,Vol. 149, 2002, pp. 433-420.

[9] Sung-Sae Lee, Sang-Kyoo Han, and Gun-Woo Moon, "Analysis andDesign of Asymmetrical ZVS PWM Half Bridge Forward Converterwith Flyback Type Transformer," Proceeding of the 35th Annual IEEEPower Electronics Specialists Conference, 2004, p. 1525-1530.

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