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Transcript of Wide Band Linearly Tapered Slot
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THE UNIVERSITY OF QUEENSLAND
School of Information Technology and Electrical Engineering
Submitted for the degree of Bachelor of Engineering (Honours)
in the division of Electrical Engineering
May 2003
Wide Band Linearly Tapered SlotAntenna
ByJ ustin J oseph Paul
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Justin Joseph Paul
8/260 Sir Fred Schonell Drive
St. Lucia, QLD 4067
Australia
Tel. (07) 32177851
23rd May 2003
The Dean
School of Information Technology and Electrical Engineering
University of Queensland
St. Lucia, QLD 4067
Dear Sir,
In accordance with the requirements of the degree of Bachelor in Engineering (Honours) in
the division of Electrical Engineering at the University of Queensland, I present the
following thesis entitled A Wide Band Linearly Tapered Slot Antenna. This work was
performed under the supervision of Associate Professor Nicholas Shuley.
I declare that all the work submitted in this thesis is my own, except as acknowledged in
the text, and have not been submitted for a degree at the University of Queensland or any
other institution.
Yours sincerely,
Justin Joseph Paul
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Abstract
i
Abstract
A Wide Band Linearly Tapered Slot Antenna (LTSA) acting as a receiving antenna used
for identification of objects through measurement of pulses was experimentally designedand constructed. Since a simulated design along with an identical prototype was completed,
results obtained for the S11 return loss, bandwidth and radiation patterns of both the
simulated and measured results were evaluated.
The simulation of the Wide Band LTSA was done on a program called FEKO. A return
loss of 48dB and a usable bandwidth of 66% was obtained for the simulated design
whereas the prototype obtained a return loss of 39dB and a usable bandwidth of only 15%.
There seem to be an alarming difference in the usable bandwidth and thus, we can conclude
that a wide bandwidth was apparent only for the simulated design and not the prototype.
The radiation patterns, however, produce almost similar results for both the simulated and
measured designs. This proved that the signal is being transmitted in the same directions in
both cases.
At the end of the thesis, a simulated design of a Wide Band LTSA was successfully
constructed into a prototype and tested. The overall results obtained agreed, to a certain
extent, with the research and theoretical background found in several journals on this thesis
topic.
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Acknowledgements
ii
Acknowledgements
I wish to express sincere thanks to my supervisor, Associate Professor Nicholas Shuley, for
his patience and guidance throughout the course of this project. This thesis would not havebeen completed without his invaluable time and advice given during the design process of
this thesis.
I would like to thank my parents, who gave me unlimited love and support throughout these
years, even though they are all the way back home in Singapore. This would not have been
possible without them.
Finally, many thanks go out to all those who have helped me throughout the course of the
project. I would like to thank Keith and Denis from the UQ Electronics Laboratory, for
rendering their professional skills to the completion of the prototype. And also to Russell
Clark from the Microwave Laboratory, who provide assistance during the setting up and
testing of the antenna in the Anechoic Chamber.
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Contents
iii
Contents
Abstract i
Acknowledgments ii
L ist of Figures vi
List of Tables ix
1 Introduction 1
1.1 Aims and Objectives of the Thesis 1
1.2 Overview of Thesis 2
1.3 Simulation Program: FEKO 3
2 The Tapered Slot Antenna 5
2.1 Characteristics of a Tapered Slot Antenna 5
2.1.1 Radiation Characteristics 6
2.1.2 Bandwidth Characteristics 6
2.2 Design Considerations 6
2.3 Taper Profiles 7
1.3.1 Effect of Curvature on Taper Profile 9
2.4 Feeding Techniques 10
2.4.1 Coaxial Line Feed 11
2.4.2 Microstrip Line Feed 13
2.5 Summary 15
3 Microstrip Transmission Line 16
3.1 Microstrip Principles 16
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Contents
iv
3.2 Substrate Materials 18
3.3 Microstrip Design Formulas 19
3.3.1 Effective Dielectric Constant 20
3.3.2 Wavelength 21
3.3.3 Characteristic Impedance 22
3.4 Quarter-wave Microstrip Transformer 22
3.4.1 Design on FEKO 23
3.4.2 Simulated Results Using FEKO 25
3.5 Discussion of Results 26
3.6 Summary 27
4 Microstrip to Slot Transition 28
4.1 Microstrip to Slot Transition 28
4.1.1 Microstrip to Slot Transition 28
Using a Double Y Balun
4.2 A Back-to-Back Microstrip to Slot Transition 30
4.2.1 Design on FEKO 30
4.2.2 Simulated Results 33
4.2.3 Sketch of Prototype 34
4.2.4 Measured Results 35
4.3 Discussion of Results 37
4.4 Summary 38
5 Design and Simulated Results of the Wide Band LTSA 39
5.1 Features of LTSA 39
5.2 Design Considerations 39
5.3 Design on FEKO 40
5.4 Simulated Results 41
5.4.1 S11 Return Loss and Bandwidth 42
5.4.2 Radiation Patterns 42
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Contents
v
5.5 Discussion of Results 45
5.6 Summary 47
6 Prototype and Measured Results of the Wide Band LTSA 48
6.1 Sketch of Prototype 48
6.2 Measured Results 48
6.2.1 S11 Return Loss and Bandwidth 50
6.2.2 Radiation Patterns 50
6.3 Discussion of Results 53
6.4 Summary 54
7 Evaluation 55
7.1 Evaluation of Simulated and Measured 55
S11 Return Loss and Bandwidth
7.2 Evaluation of Simulated and Measured 56
Radiation Patterns
7.3 Summary 57
8 Conclusion 58
8.1 Future Work 59
Appendix A FEKO Programs 61
Appendix B Sketches 69
References 71
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List of Figures
vi
List of Figures
Chapter 2:
Figure 2.1: Top view of Taper Slot Antenna 7
Figure 2.2: Cross-sectional View of Slotline 7
Figure 2.3: Different Taper Profiles of a TSA: (a) exponential, 8
(b) tangential, (c) parabolic, (d) linear, (e) linear-constant,
(f) exponential-constant, (g) step-constant, (h) broken linear
Figure 2.4: Schematic of TSA Taper Profiles 10
Figure 2.5: Different Feeding Techniques of a TSA: (a) coaxial line, 11
(b) microstrip line, (c) CPW, (d) air-bridge/GCPW,
(e) FCPW/centre-strip, (f) FCPW/notch
Figure 2.6: Model of Coaxial Line to Slot Transition 12
Figure 2.7: Equivalent Circuit of Coaxial Line to Slot Transition 12
Figure 2.8: Model of Microstrip to Slotline Transition 14
Figure 2.9: Equivalent circuit of a Microstrip to Slotline Transition 15
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List of Figures
vii
Chapter 3:
Figure 3.1: Structure of Microstrip Transmission Line 17
Figure 3.2: Wide and Narrow Microstrip Line 20
Figure 3.3: Equivalent Circuit of a Quarter Wave Microstrip Transformer 23
Figure 3.4: Top View of Design in WinFEKO 24
Figure 3.5: Return Loss obtained from GraphFEKO 25
Figure 3.6: Voltage Standing Wave Ratio obtained from GraphFEKO 26
Chapter 4:
Figure 4.1: Microstrip to Slot Transition Using a Double Y-balun 29
Figure 4.2: Equivalent Circuit of Back-to-Back Microstrip to Slot Transition 31
Figure 4.3: Top View of Back-to-Back Microstrip to Slot 32
Transition in WinFEKO
Figure 4.4: Simulated Return Loss, S11, of a Back-to-Back 33
Microstrip to Slot Transition
Figure 4.5: Top View of Finished Prototype 35
Figure 4.6: Bottom View of Finished Prototype 36
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List of Figures
viii
Figure 4.7: Measured Return Loss, S11, of a Back-to-Back 37
Microstrip to Slot Transition
Chapter 5:
Figure 5.1: Top View of Wide Band LTSA in WinFEKO 40
Figure 5.2: Simulated Return Loss, S11, of the Wide Band LTSA 42
Figure 5.3: Radiation Patterns for E-plane 44
Figure 5.4: Radiation Patterns for H-plane 45
Chapter 6:
Figure 6.1: Top View of Finished Prototype 49
Figure 6.2: Bottom View of Finished Prototype 49
Figure 6.3: Measured Return Loss, S11, of the Wide Band LTSA 50
Figure 6.4: General Orientation of Wide Band LTSA for E-plane and H-plane 51
Figure 6.5: Radiation Patterns for E-plane 52
Figure 6.6: Radiation Patterns for H-plane 52
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List of Tables
ix
List of Tables
Table 3.1: Properties of Microwave Dielectric Substrates 19
Table 4.1: Parameters of Back-to-Back Microstrip to Slot Transition 31
Table 5.1: Parameters for the Wide Band LTSA 41
Table 7.1: Compiled Simulated and Measured Results 55
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Chapter 1: Introduction
1
Chapter 1: Introduction
This chapter highlights the importance of the Wide Band Linearly Tapered Slot Antenna
(LTSA). After defining the aims and objectives of the thesis, this chapter closes with an
overview of the thesis. The overview briefly summarizes the topics discussed in every of
the following chapters in this thesis.
1.1 Aims and Objectives of the Thesis
The aim of this thesis is to successfully design and construct a wide band LTSA that is able
to identify objects through measurement of pulse responses. Aiming to complete the design
and construction of this project on schedule is given the highest priority. Also, obtaining
similar simulated and measured results is another objective to fulfil. Throughout the course
of the thesis, gaining a comprehensive understanding of travelling wave antennas, matching
capabilities as well as microstrip and slotline characteristics will definitely provide useful
knowledge for the future.
The antenna and all the designs carried out in this thesis were carried out using ROGERSdielectric substrate with relative permittivity of 2.0 and thickness of 0.5mm. All the
simulated and measured results were done over a frequency range of 1 GHz to 8 GHz.
FEKO is a new program introduced due to the need to design and simulate the wide band
LTSA. Being able to gain knowledge on FEKO will prove to be a useful tool for the
present and future. This, together with learning and understanding how basic antennas
generally work, are also aims of this project.
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Chapter 1: Introduction
2
1.2 Overview of Thesis
Chapter 1begins with some aims and objectives set at the commencement of the thesis. An
overview of the thesis will also be given at the end of the chapter.
Chapter 2begins with an introduction of a basicTapered Slot Antenna. Characteristics and
design considerations will give a better insight into the operations of the antenna. Most
importantly, the effects of the taper profile and two feeding techniques will be discussed.
Chapter 3 will provide an in depth look into the principles of a Microstrip Transmission
Line. Different substrate materials that can be used for antennas are mentioned along with a
table highlighting their properties. Microstrip design formulas are also included in this
chapter to provide useful equations for designing. Finally, all the above are implemented in
a Quarter Wave Microstrip Transformer design that is done using FEKO.
Chapter 4 begins with an introduction of the most important factor leading to the
completion of the wide band linearly tapered slot antenna, the Microstrip to Slot Transition.
An extremely useful implementation of this transition is through a Double Y-Balun. The
first prototype designed and built during this thesis was a Back-to-Back Microstrip to Slot
Transition. Steps that were taken while approaching this design as well as simulated results
obtained are also covered in this chapter. These simulated results are then compared with
measured results obtained from the laboratory.
Chapter 5will cover theDesign and Simulated Results of theWide Band Linearly Tapered
Slot Antenna. A detailed look into the features and the design considerations of the antenna
is available in this chapter. Simulated results obtained for the antenna design will be
discussed at the end of the chapter.
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Chapter 1: Introduction
3
Chapter 6 will focus mainly on the Prototype and Measured Results of the Wide Band
Linearly Tapered Slot Antenna. In this chapter, information through sketches of the
prototype and measured results will be provided. The chapter will conclude with
discussions done on the measured results.
Chapter 7 will contain the Evaluation of the results obtained for both the simulated and
measured results of the wide band linearly tapered slot antenna. Comparisons will be made
between both the results and explanations of similarities and differences will be done as
well.
Chapter 8 will give an overall Conclusion by summarising all the work done during the
course of it and future prospects for the wide band linearly tapered slot antenna.
1.3 Simulation Program: FEKO
The program FEKO is based on the Method of Moments (MOM). Electromagnetic fields
are obtained by first calculating the electric surface currents on conducting surfaces and
equivalent magnetic and electric surface currents on the surface of a dielectric solid. The
currents are calculated using a linear combination of basis functions, where the coefficients
are obtained by solving a system of linear equations. Once the current distribution is
known, further parameters such as near and far field, radar cross sections, directivity of
input impedance can be found. Only time domain harmonic sources are supported in the
current version and calculation is done in the frequency domain. FEKO uses ejt time
conversion.
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Chapter 1: Introduction
4
WinFEKO is the main user interface module and is used to control the solution of a
problem. The geometry is defined in terms of high level commands in the *pre input file
which also sets the solution parameters. The customised text editor, EditFEKO, assists the
user in creating and editing the input file. The processor/mesher, PREFEKO, processes this
file and prepares the input file, *fek, for the program FEKO which is actually the field
calculation code. The PREFEKO enables the user to create complex geometries with a
single command. The output file, *out, of FEKO contains all the solution information. The
resulting fields and/or currents can be displayed in 3D in WinFEKO or as 2D plots in
GraphFEKO.
The above description of the simulation program FEKO is a summarised version obtained
from the User Manual of FEKO. For more detailed information on FEKO, this manual is
recommended. To put the above in laymans terms, EditFEKO is used to write the code for
the simulation program. Upon running PREFEKO, we are then able to obtain a
visualisation of the outlook of the design in WinFEKO. Finally, all the 2D plots for the
simulation can be obtained through GraphFEKO.
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Chapter 2: The Tapered Slot Antenna
5
Chapter 2: The Tapered Slot Antenna
The chapter begins with a review on some characteristics of the Tapered Slot Antenna. This
is followed by design aspects to be taken into consideration in the process of constructing
the antenna. In general, designs differ only in the taper profile of the slot and the feeding
technique. Some of the common taper profiles and feeding techniques will be presented in
this chapter.
2.1 Characteristics of a Tapered Slot Antenna
The tapered slot antenna (TSA) belong to the general class of end-fire travelling wave
antennas and consist of a tapered slot etched onto a thin film of metal. This is done either
with or without a dielectric substrate on one side of the film. Besides being efficient and
lightweight, the more attractive features of TSAs are that they can work over a large
frequency bandwidth and produce a symmetrical end-fire beam with appreciable gain and
low side lobes [1]. An important step in the design of the antenna is to find suitable feeding
techniques for a slotline excited TSA.
Understanding the characteristics of the TSA is fundamental and would help a great deal in
designing the antenna. From research journals on the TSA, we can confirm that TSAs
generally have wider bandwidth, higher directivity and are able to produce symmetrical
radiation patterns [2].
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Chapter 2: The Tapered Slot Antenna
6
2.1.1 Radiation Characteristics
As the TSA is a travelling wave antenna, the phase velocity and the guide wavelength, g,
varies with the change in thickness, dielectric constant and taper shape. Having the gain
proportional to L/g, parameters such as length, width and taper profiles also have direct
impact on the radiation patterns, directivity and cross-polarisation level of the antenna. The
radiation characteristics of the antenna are also affected by the substrate thickness and
ground plane. [4]
2.1.2 Bandwidth Characteristics
The TSA is capable of having an operating bandwidth within a frequency range of 2 GHz
to 90 GHz. To achieve a wider bandwidth, it is ideal for the TSA to have a perfect
impedance match at both the feed transition and the slot termination. Different methods for
bandwidth broadening depend on the feed methods chosen. This will be described further
in section 2.4. The bandwidth is normally proportional to the change in frequency,f. [4]
2.2 Design Considerations
A TSA is formed by slowly increasing the width of a slot from the point of its feed to an
open end of width generally greater than O/2 [3]. This is illustrated in figure 2.1.
Experimental results done in various journals have confirmed that the impedance,
bandwidth and radiation patterns are greatly affected by parameters such as length, width
and taper profile of a TSA. The dielectric substrates thickness and relative permittivity are
also important as they contribute to the efficiency of the antenna. Figure 2.1 and 2.2 show
the top view and cross-sectional view of a slotline on a dielectric substrate with its
important parameters illustrated. The shaded area in both the figures represents the
remaining copper on the dielectric substrate after etching is done.
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Chapter 2: The Tapered Slot Antenna
7
Figure 2.1: Top view of Taper Slot Antenna
Figure 2.2: Cross-sectional View of Slotline
2.3 Taper Profiles
Many taper profiles exist for a normal TSA. Figure 2.3 shows different planar designs and
we can observe that each antenna differs from one another only in the taper profile of the
slot. Of all the designs illustrated in figure 2.3 [4], only the Vivaldi [5] and linearly tapered
profile [3] have been thoroughly studied over the past few years.
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Chapter 2: The Tapered Slot Antenna
8
Planar tapered slot antennas have two common features. The radiating slot acts as the
ground plane for the antenna and the antenna is fed by a balanced slotline. However,
drawbacks for a planar TSA come in the form of using a low dielectric constant substrate
and obtaining an impedance match for the slotline. By fabricating on a low dielectric
constant substrate, relatively high impedance is obtained for the slotline. If a microstrip
feed is chosen, it makes matching very difficult. Thus, the microstrip to slot transition will
limit the operating bandwidth of the TSA.
(a) (e)
(b) (f)
(c) (g)
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Chapter 2: The Tapered Slot Antenna
9
(d) (h)
Figure 2.3: Different Taper Profiles of a TSA: (a) exponential, (b) tangential, (c)
parabolic, (d) linear, (e) linear-constant, (f) exponential-constant, (g) step-constant,
(h) broken linear
2.3.1 Effect of Curvature on Taper Profile
Tapered slot antennas with linear, exponential or constant taper profile are commonly
reported and their journals can be easily found. However, information on the effects of the
curvature on a taper profile is not readily available. From the authors of [6], we are able to
obtain experimental investigation and results on the effects. The important points will be
briefly summarized in this sub-section. For more detailed explanation and illustrations, it is
recommended that the particular journal be referred to.
Figure 2.4 shows the schematic of linear (1) and exponential (2), (3) and (4) taper profiles
of a TSA. As seen in the figure, four TSAs of same length and terminating slot width, but
with different taper profiles, were fabricated and tested. Fabrication was done on the same
type of substrate with the same relative permittivity.
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Chapter 2: The Tapered Slot Antenna
10
Figure 2.4: Schematic of TSA Taper Profiles
The cross polarization is generally improved with the decrease in the radius of the curvature
except for the E-plane, which will not show any improvement. More importantly, thedecrease on the radius of the curvature also reduces the bandwidth of the antenna. This is
not ideal as the later part of the thesis focuses on designing an antenna with a wide
bandwidth.
2.4 Feeding Techniques
A slot generally always excites a TSA. In order to test and design slotline circuits, it is
necessary to have a transition between a slot and another transmission medium [7]. These
transitions should be very compact and have low loss. Some feeding techniques and their
transitions are shown in the figure 2.5 [4]. The commonly used methods are the coaxial line
feed and the microstrip line feed. These will be illustrated and discussed in the next two
sub-sections.
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Chapter 2: The Tapered Slot Antenna
11
(a) (b)
(c) (d)
(e) (f)
Figure 2.5: Different Feeding Techniques of a TSA: (a) coaxial line, (b) microstrip
line, (c) CPW, (d) air-bridge/GCPW, (e) FCPW/centre-strip, (f) FCPW/notch
2.4.1 Coaxial Line Feed
A coaxial line feed provides a direct path for coupling of fields across the slot [4]. A
commonly used coaxial line to slot transition is shown in figure 2.6. The transition consists
of a coaxial line placed perpendicular at the end of an open circuited slot. The outer
conductor of the cable is electrically connected to the ground plane on one side of the slot
while the inner conductor of the coaxial line forms a semicircular shape over the slot as
shown in figure 2.6. This transition has been analysed in [8]. An equivalent circuit, also
from [8], is shown in figure 2.7.
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Chapter 2: The Tapered Slot Antenna
12
Figure 2.6: Model of Coaxial Line to Slot Transition
Figure 2.7: Equivalent Circuit of Coaxial Line to Slot Transition
From the equivalent circuit, we can predict that the slot impedance will be transform to a
lesser value, by a factor of n, so as to match a 50 coaxial cable. To do this, a slot
impedance of around 75 is needed. However, in practice, it is difficult to obtain a slot
impedance of around 75 because a slotline impedance of less than 100 will have a
very small width and this makes fabrication with etching difficult and inaccurate.
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Chapter 2: The Tapered Slot Antenna
13
2.4.2 Microstrip Line Feed
A microstrip to slot transition consists of a slot, etched on one side of the substrate,
crossing an open circuited microstrip line, located on the opposite side, at a right angle. The
slot extends to one quarter of a wavelength beyond the microstrip and the microstrip
extends one quarter of a wavelength beyond the slot as shown in figure 2.8. The latters
wavelength has to minus a length extension ofL. The length extension is due to fringing
at the end of the open circuited line, which makes the line appear electrically longer [4].
The length extension can be approximated using the following expression [9]:
( )
( )
+
++=
8.0258.0
264.03.0412.0
,
,
h
wh
wh
L
effr
effr
(2.1)
An equivalent circuit of the microstrip to slot transition is shown in figure 2.9 [4]. An
impedance match between the microstrip and slotline can be obtained at a given frequency
by applying equation (2.2). The equation can also be applied to the coaxial to slot
transition.
ms ZnZ2= (2.2)
where ( )
=
oo
utqutn
2sincot2cos (2.2.1)
)vu
out
q
1
tan2
+=
(2.2.2)
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Chapter 2: The Tapered Slot Antenna
14
,
2
=
s
oru
1
2
=
s
ov
(2.2.3)
where Zs = characteristic impedance of the slot, Zm = characteristic impedance of themicrostrip, o =free space wavelength ands =slotline wavelength
To achieve proper impedance match, multi-step quarter wave transformers are sometimes
used. By terminating the microstrip with a radial stub and the slot with an elliptical shaped
cavity, the bandwidth can be broadened. Also, when terminating the slot with the elliptical
cavity, the operating bandwidth of the transition tends to shift down in frequency [4].
Figure 2.8: Model of Microstrip to Slotline Transition
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Chapter 2: The Tapered Slot Antenna
15
Figure 2.9: Equivalent circuit of a Microstrip to Slotline Transition
2.5 Summary
From this chapter, a better understanding on the characteristics and design considerations
inevitably helps in the designing and constructing of a tapered slot antenna. Various taper
profiles and feeding techniques were described and illustrated to give the reader different
options while designing a TSA. The effects the angle of the taper profile has on the antenna
were also highlighted. Finally, the transitions of the two more common feeding techniques,
coaxial line and microstrip line, were explained.
The overall design of the wide band LTSA was closely modelled after some of the figures
presented in this chapter. Figures 2.1 and 2.2 were taken into consideration when designing
the linearly taper profile of the slot. Figure 2.8 illustrates the most important considerations
that the microstrip to slot transition was designed after. However, a good transition at one
particular frequency does not work well over all the frequencies as the wavelength changes
with the frequency.
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Chapter 3: Microstrip Transmission Line
16
Chapter 3: Microstrip Transmission Line
This chapter will provide an overview on the basic principles and operation of a microstrip
transmission line. By providing definitions on important parameters and a detailed
description on its operation, one is able to gain a better knowledge and understanding of the
design specifications of a microstrip transmission line. This is essential in the latter
chapters as the microstrip transmission line plays an important part in this thesis.
3.1 Microstrip Principles
The microstrip transmission line is the most commonly used Microwave Integrated Circuit
(MIC) transmission medium and is also one of the most popular type of planar transmission
line. A planar configuration implies that the dimensions in a single plane can determine the
characteristics of the element. For example, the width, w, of a microstrip line on a dielectric
substrate can be adjusted to control its impedance.
The microstrip transmission line is popular due to the fact that the mode of propagation on
microstrip is almost TEM. This allows easy approximate analysis and yields wide bandcircuits [7].
The structure of a microstrip transmission line is shown in the figure 3.1. The most
important dimension parameters of a microstrip circuit design are the width, w, of the
microstrip line and the height, h, which is equivalent to the thickness of the dielectric
substrate [10]. The relative permittivity, r, of the substrate is also another important
parameter. The fabrication of a microstrip transmission line is often done through etching
on a microwave substrate material.
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Chapter 3: Microstrip Transmission Line
17
Figure 3.1: Structure of Microstrip Transmission Line
There are generally two types of dielectric substrates, soft and hard. Soft substrates are
normally used as they are flexible, cheap and can be easily fabricated. Hard substrates have
better reliability and lower thermal expansion coefficients. However, they are more
expensive and not flexible. Substrates materials will be mentioned in the next section.
From the above, we can conclude that the microstrip line has many advantages, such as low
cost, small in size and use of photolithographic method for fabrication that leads to good
repeatability, reproducibility and ease of mass production. However, the microstrip line
does have its disadvantages that include higher loss, lower power-handling capability and
greater temperature instability. The thickness of the strip, t, and the conductivity, , are not
important parameters and are often neglected.
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Chapter 3: Microstrip Transmission Line
18
3.2 Substrate Materials
The choice of dielectric substrate plays an important role in the design and simulation of
the microstrip transmission line as well as any other antennas. Some important dimensions
of the dielectric substrate are:
The dielectric constant.
The dielectric loss tangent that sets the dielectric loss.
The thermal expansion and conductivity.
The cost.
The manufacturability.
The thickness of the copper surface.
There are numerous types of substrates that can be used for the design of antennas. They
often have different characteristics and their dielectric constants normally range from 2.2
r 12. Thick substrates with low relative dielectric constants are often used as they
provide better efficiency and a wider bandwidth. However, using thin substrates with high
dielectric constant would result in smaller antenna size. But this also results negatively on
the efficiency and bandwidth. Therefore, there must be a design trade-off between antenna
size and good antenna performance [11].
Material Relative
Dielectric
Constant
Loss Tangent at
10 GHz (tan )
Thermal
Conductivity, K
(W/cm/C)
Dielectric
Strength
(kV/cm)
Sapphire 11.7 10-4 0.4 4 x 103
Alumina 9.7 2 x 10-4 0.3 4 x 103
Quartz (fused) 3.8 10-4 0.01 10 x 103
Polystyrene 2.53 4.7 x 10-4 0.0015 280
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Chapter 3: Microstrip Transmission Line
19
Beryllium Oxide
(BeO)
6.6 10-4 2.5 -
GaAs 12.3 16 x 10-4 0.3 350
Si 11.7 50 x 10-4 0.9 300
3M 250 type GX 2.5 19 x 10-4 0.0026 200
Keene DI-clad
527
2.5 19 x 10-4 0.0026 200
RT Duriod 5870 2.35 12 x 10-4 0.0026 200
3M Cu-clad 233 2.33 12 x 10-4 0.0026 200
Keene DI-clad
870
2.33 12 x 10-4 0.0026 200
RT Duriod 5880 2.20 9 x 10-4 0.0026 200
3M Cu-clad 217 2.17 9 x 10-4 0.0026 200
Keene DI-clad
880
2.20 9 x 10-4 0.0026 200
RT Duriod 6010 10.5 15 x 10-4 0.004 160
3M epsilon IU 10.2 15 x 10-4 0.004 160
Keene DI-clad
810
10.2 15 x 10-4 0.004 160
Air 1.0 0 0.00024 30
Table 3.1: Properties of Microwave Dielectric Substrates
3.3 Microstrip Design Formulas
To design a basic microstrip transmission line, one must be able to obtain dimensions such
as effective dielectric constant, wavelength and characteristic impedance. This can be
calculated through some simple equations that will be shown in the next few sub-sections.
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Chapter 3: Microstrip Transmission Line
20
3.3.1 Effective Dielectric Constant
One might think that the effective dielectric constant, r,eff, is the same as the dielectric
constant, r, of the substrate. This appears to be true only for a homogeneous structure and
not for a non-homogeneous structure. For microstrip structures, we are able to calculate the
effective dielectric constant that comes in two different cases. These two cases are
illustrated in figure 3.2 whereby the top diagram shows a microstrip with width, w, greater
than the thickness, h, of the substrate (wh). The opposite can be said about the bottom
diagram [10].
Figure 3.2: Wide and Narrow Microstrip Line
By looking at the diagram with wh, we can conclude that the circuit performs similar to
having two parallel planes as most of the fields as kept under the wide microstrip width.
Thus, reff is approximately equivalent to r. When wh, half of the fields will be in air with
r =1, while the other half of the fields will be confined to the substrate with r,eff=(r +
1). Therefore, the range of a dielectric constant can be said to be:
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Chapter 3: Microstrip Transmission Line
21
( ) reffrr + ,12
1(3.1)
The following equations can be used to obtain a precise value ofr,eff. Equations (3.2) and
(3.3) take into consideration negligible thickness of the microstrip.
1104.012
12
1
2
122
1
,
+
+
++=
h
wfor
h
w
hw
rreffr
(3.2)
112
12
1
2
12
1
,
+++=
h
wfor
hw
rreffr
(3.3)
3.3.2 Wavelength
For a propagating wave in free space, the wavelength of that medium is equal to the speed
of light divided by its operating frequency. To obtain the wavelength of a given wave-guide
or antenna, the free space wavelength is simply divided by the square root of the effective
dielectric constant of the wave-guide. These are shown in equations (3.4) and (3.5).
o
of
c= (3.4)
effr
og
,
= (3.5)
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Chapter 3: Microstrip Transmission Line
22
where c =speed of light, fo =operating frequency, o =free space wavelength andg=the
guide wavelength.
3.3.3 Characteristic Impedance
The characteristic impedance, Zo, of any line is the function of its geometry and dielectric
constant. For a microstrip transmission line, the characteristic impedance is defined as the
ratio of voltage and current of a travelling wave. For a microstrip line with width, w, we are
able to calculate the characteristic impedance through the following two equations [12]:
125.08
ln60
,
+=
h
wfor
h
w
hw
Zeffr
o
(3.6)
( )1
444.1ln667.0393.1
120
, +++
=h
wfor
hw
hw
Zeffr
o
(3.7)
Note: Negligible microstrip thickness is taken into consideration
3.4 Quarter Wave Microstrip Transformer
A quarter wave microstrip transformer was modelled and designed in FEKO. Figure 3.3
shows an equivalent circuit that is simple and useful for matching a real load impedance to
a transmission line. A step in width will exist at the junction of two microstrip lines due to
both lines having different impedance. This is commonly encountered when designing
transitions. Designing a single section quarter wave microstrip transformer will prove to be
useful in the latter part of this thesis involving the transition from microstrip to slotline.
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Chapter 3: Microstrip Transmission Line
23
Figure 3.3: Equivalent Circuit of a Quarter Wave Microstrip Transformer
One drawback of the quarter wave transformer is that it can only match a real load
impedance. However, by using an appropriate length of transmission line between the load
and the transformer, a complex load can always be changed into a real impedance.
3.4.1 Design on FEKO
A microstrip line with a characteristic impedance, Zo, of 50 is matched to a real load
impedance, ZL, of 100 by a single section quarter wave transformer. The characteristicimpedance of the quarter wave matching section can be obtained by equation (3.8).
LoZZZ =1 (3.8)
where Zo, Z1 and ZL represent the given characteristic impedances as seen in figure 3.3
Programs such as PUFF and PICAARD can easily determine the length, l, of the singlesection quarter wave transformer.
Z Z1 ZL
l
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Chapter 3: Microstrip Transmission Line
24
From equation (3.8), we are able to calculate the characteristic impedance, Z1, of the
quarter wave matching section to be 70.71. By setting a design frequency, fo, we are also
able to obtain the electrical length of the quarter wave section, g/4, of the matching section
through the wavelength equations given previously. However, the electrical length of the
matching section will definitely differ at other frequencies and thus, a perfect match can no
longer be achieved.
Figure 3.4 shows the top view of the microstrip design as seen in WinFEKO. The pink area
surrounding the design represents the substrate with a dielectric constant of 2.0 and
thickness of 0.5mm. From the figure, we can clearly see a length of microstrip transmission
line, in orange and yellow, stepping down to a single section quarter wave transformer, in
green. This looks similar to the one seen in figure 3.3. The FEKO code for this design is
attached in the Appendix A.
Figure 3.4: Top View of Design in WinFEKO
/4
ZoZ1
Terminated
withZL
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Chapter 3: Microstrip Transmission Line
25
3.4.2 Simulated Results Using FEKO
Figure 3.5 and 3.6 are results of the return loss, S11, and voltage standing wave ratio,
VSWR, of the quarter wave microstrip transformer respectively. Figure 3.5 shows the
transformer to have a very good return loss over a frequency range of 1 GHz to 8 GHz and
figure 3.6 shows a VSWR of approximately less than 2 for the same frequency range.
Figure 3.5: Return Loss obtained from GraphFEKO
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Chapter 3: Microstrip Transmission Line
26
Figure 3.6: Voltage Standing Wave Ratio obtained from GraphFEKO
3.5 Discussion of Results
From the results obtained for the return loss and standing wave ratio of the quarter wave
microstrip transformer, we can conclude that the 50 transmission line matches well with
the terminated load impedance of 100 through the 70.71 single section quarter wave
transformer. Normally, antennas generally require a return loss of at least 10dB or a
VSWR of less than 2 for it to work effectively. Both the graphs confirm that these criteria
are met and that the design works excellently over a wide frequency range of 1 GHz to 8
GHz.
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Chapter 3: Microstrip Transmission Line
27
3.6 Summary
Understanding the principles and operation of a quarter wave microstrip transformer helped
during the course of this thesis project. This is because the basic knowledge of how to
match impedance was learnt through design and simulation of this transformer. Matching
of impedances will play an important part in the latter part of the thesis because the wide
band linearly tapered slot antenna requires a microstrip to slot transition.
Many problems were faced during the first design stage of the thesis because learning the
use of FEKO proved to be difficult task. As the quarter wave microstrip transformer was
the first design done on FEKO, a better understanding of how to operate FEKO and the
various geometrical and control options was obtained.
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Chapter 4: Microstrip to Slot Transition
28
Chapter 4: Microstrip to Slot Transition
The microstrip to slot transition is the most important factor when it comes to the wide
band linearly tapered slot antenna. The point at which the microstrip crosses the slot has to
be almost exact for an antenna to produce a good performance. This chapter will further
describe the implementation of the transition through a double Y-balun and a back-to-back
transition.
4.1 Microstrip to Slot Transition
By feeding a slot with a microstrip line, we are creating a transition between the two. This
transition has been described and illustrated in chapter 2 under section 2.4. An improved
microstrip to slot transition has been proposed using a double Y-balun. Theoretically, this
would prove to be extremely useful for the microstrip to slot transition. However, due to
time constrains of this thesis, the double Y balun was not implemented during the design of
the wide band linearly tapered slot antenna. Hence, only a brief description of the balun
will be given in the next sub-section to give the reader an outlook as to how the transition is
done using a double Y balun.
4.1.1 Microstrip to Slot Transition Using a Double Y Balun
The double Y-balun is an extremely effective method while doing a microstrip to slot
transition. Double Y-baluns are based on the 6 port double Y junction, which consists of 3
balanced and 3 unbalanced lines placed alternately around the centre of the structure.
Figure 4.1 illustrates how a double Y-balun looks like when used in a microstrip to slot
transition [13].
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Chapter 4: Microstrip to Slot Transition
29
.
Figure 4.1: Microstrip to Slot Transition Using a Double Y-balun
In order to make a structure work as a balun with perfect transmission between an opposite
balanced and unbalanced ports, opposite pairs of lines should have reflection coefficients
with opposite phases. This means that one pair of lines should be short circuit and the other,
open circuit. The electrical lengths of the lines from the open or short circuits to the centre
of the junction should be equal:
ssmm ll = (4.1)
Wheremands are the phase constants and lmand ls are the line lengths for microstrip and
slot, respectively. In experimental realization, the values of length, l, are made small. To
avoid radiation effects, the following should be observed:
42
l (4.2)
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Chapter 4: Microstrip to Slot Transition
30
4.2 Back-to-Back Microstrip to Slot Transition
By designing a back-to-back microstrip to slot transition, we are basically creating a two-
port network. A microstrip line, following the same guidelines stated in the microstrip to
slot transition, feeds the slot antenna that is terminated by an elliptical shaped cavity. This
part will be used for the wide band linearly tapered slot antenna as well. A back-to-back
microstrip to slot transition is created by simply duplicating this design symmetrically at
the end of the slot.
4.2.1 Design on FEKO
The back-to-back microstrip to slot transition was again designed using FEKO. Figure 4.2
shows a equivalent circuit of the transition and figure 4.3 shows the top view of it in
WinFEKO. The design incorporates a microstrip transmission line with a single section
quarter wave transformer that was described in Chapter 3. The design was created using the
measurements of a ROGERS dielectric substrate with a dielectric constant of 2.0 and
thickness of 0.5mm. This particular substrate was use because of its mass availability in the
laboratory. Other substrates of different dielectric constants were available but not in
quantity. This might prove to be a problem in repeated fabrication as the design might have
to be redesigned when the substrate runs out.
The FEKO code for this design can be found in the Appendix A. Included in this code are
the parameters for the back-to-back microstrip to slot transition. They are as follows:
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Chapter 4: Microstrip to Slot Transition
31
Length of Substrate 132mm
Width of Substrate 61mm
Width of Slot, Ws 3mm
Length of Slot, Ls 50mm
Width of Microstrip Transmission Line, Wm 2.4mm
Length of Microstrip Transmission Line, Lm 20mm
Width of Quarter Wave Section, Wq 1.2mm
Length of Quarter Wave Section, L q 21.5mm
Radius of Elliptical Cavity 20mm
Table 4.1: Parameters of Back-to-Back Microstrip to Slot Transition
The dimensions given in the previous page were a combination of analysis of designs used
in journals as well as through trial and error. Through the dimensions used, we were able to
obtain the best available results.
Figure 4.2: Equivalent Circuit of Back-to-Back Microstrip to Slot Transition
Slotline
O/C
Line of Symmetry
50 Load
Excitation
MicrostripMicrostrip
Slotline
O/C
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Chapter 4: Microstrip to Slot Transition
32
Figure 4.3: Top View of Back-to-Back Microstrip to Slot Transition in WinFEKO
Similar to the microstrip design in Chapter 3, the pink surface represents the dielectric
substrate. On the surface, we are able to see many triangles that represent the metallic
surface of the substrate. The slot is terminated by an elliptical cavity on both sides. There
are also two strips crossing the slot that represents the location of the microstrips located at
the bottom of the substrate. One side of the transition is excited by a 50 microstrip line
while the microstrip line on the other half is terminated by a 50 real load impedance.
This can be interchanged due to the design being symmetrical.
Line ofSymmetry
EllipticalCavity
MicrostripLine(bottom ofsubstrate)
Slot
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Chapter 4: Microstrip to Slot Transition
33
4.2.2 Simulated Results
When simulating the back-to-back microstrip to slot transtion in FEKO, only the result for
return loss, S11, was really looked at and taken into consideration. This is because the
design is done just to prove the matching capabilities of a symmetrical microstrip to slot
transition through a two-port network. If it matches well, we are then able to use half of the
design and attach it to a linear taper profile.
Figure 4.4: Simulated Return Loss, S11, of a Back-to-Back Microstrip to Slot
Transition
Figure 4.4 shows the return loss of the transition obtained from GraphFEKO after running
the simulation in FEKO. From it, we can observe that the results of the return loss were
taken over a frequency range of 1 GHz to around 8 GHz.
UsableBandwidth
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Chapter 4: Microstrip to Slot Transition
34
By observing the result of the simulated return loss, S11, for the back-to-back microstrip to
slot transition, it can be seen that the return loss is about 30dB at the resonant frequency of
3.4 GHz. As for the usable bandwidth of this transition, it is defined over the frequency
range at which the S11 is at 10dB. Figure 4.4 shows the user bandwidth to be
approximately from 3 GHz to 7.25 GHz. Thus, the bandwidth can be calculated through the
equation:
%100)( 12 =
of
ffBW (4.3)
where fo is the frequency at which S11 is minimum
f1 and f2 are the frequencies at which S11 is at 10dB
By applying the above equation, the usable bandwidth for the back-to-back microstrip to
slot transition is approximately 125%. This result is considered to be satisfactory as the
objective of obtaining a wide bandwidth and impedance match of this transition is
achieved.
4.2.3 Sketch of Prototype
The sketch of the prototype, with all its dimensions, can be found in the Appendix B and it
shows how the antenna would look like physically. This sketch looks similar to the design
in FEKO as illustrated in figure 4.3.
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Chapter 4: Microstrip to Slot Transition
35
4.2.4 Measured Results
A physical version of the back-to-back transition was needed in order to verify the
simulated results obtained from FEKO. The program Protel was used to draw out an exact
design of both the top and bottom side of the transition with precise dimensions. Upon
completion, the Protel design was submitted to the Electronics Laboratory located in S309,
Hawken Engineering Building. The fabrication took only two hours and the etching came
out according to the given dimensions. The top and bottom view of the finished product can
be seen in figure 4.5 and 4.6 on the next page. As seen from the figures, a SMA connector
is soldered onto each of the microstrip lines. Soldering them to the side with the slot also
grounds the connectors.
Figure 4.5: Top View of Finished Prototype
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Chapter 4: Microstrip to Slot Transition
36
Figure 4.6: Bottom View of Finished Prototype
The result for the return loss of the back-to-back microstrip to slot transition was done
through a Network Analyser stationed in S507, Hawken Engineering Building. Since the
design is symmetrical, either connector can be connected to a coaxial cable which
calculates the return loss while the other is terminated by a 50 load. This results in the
design working as a one-port network.
From figure 4.7, the plot for the measured results, though not a clear one, can be seen. The
measured return loss for the transition was also taken over a frequency range of 1GHz to
8GHz on the x-axis, scaled at 0.7 GHz per division. The return loss on the y-axis was
scaled at 10dB per division. The same results were obtained when the load and excitation
were switched around.
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Chapter 4: Microstrip to Slot Transition
37
1 2.7 3.8 5.6 6.6 8 (GHz)
Figure 4.7: Measured Return Loss, S11, of a Back-to-Back Microstrip to Slot
Transition
From the plot, we can observe that the usable bandwidths for the measured results are
obtained over two sets of approximate frequency ranges: 2.7 GHz to 3.8 GHz and 5.6 GHz
to 6.6 GHz. This works out to a calculated bandwidth of around 34% for the first range and
17% for the second range. This shows that the transition to have only a narrow bandwidth
over two resonant frequencies of 3.2 GHz and 5.9 GHz. The return losses at these two
frequencies are observed to be at 37dB and 30 dB respectively.
4.3 Discussion of Results
It is normally difficult to obtain similar simulated and measured results when it comes to
testing of antennas. In the case of the back-to-back microstrip to slot transition, this proves
likewise.
S11ReturnL
oss(dB) UsableBandwidths
0
3.2 5.9
-10
-30-37
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Chapter 4: Microstrip to Slot Transition
38
For the simulated results obtained from FEKO, one has to realise that the program
considers the transition to be in an optimised environment unless stated. FEKO will only
take into consideration the dimensions and control parameters that the user includes in the
code. Other parameters, such as losses, mutual inductance and capacitance, if not included,
will be assumed to be at an optimised level. Therefore, because the transition is simulated
in an ideal case, the plot in figure 4.4 can be said to produce a good result.
By looking at figure 4.7, we can conclude that the measured result of the transition did not
produce a wide bandwidth as obtained for the simulated result. This could be due to the
factors as mentioned in the previous paragraph. However, the two peaks for the simulated
and measured return losses are located at around the same frequency point with a slightly
better return loss for the measured result. This proves that the transition works at the same
simulated and measured frequency. By testing the back-to-back transition physically, the
product tends to be subjected to other physical factors affecting it. For example, there are
bound to be losses in the cables used for testing and even a slight bend in a cable tends to
shift the result a little.
4.4 Summary
At the start of this project, the major concern was that the simulated results would not tally
with the measured results. Designing the back-to-back microstrip to slot transition allowed
the comparison of these results to be done. As expected, both the results were not similar
except for the resonant frequency. The reasons behind this have been discussed in the
previous section.
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Chapter 5: Design and Simulated Results of the Wide Band LTSA
39
Chapter 5: Design and Simulated Results of the Wide Band LTSA
A wide band linearly tapered slot antenna incorporates a slot antenna with a linear tapered
profile and is fed by a microstrip line. Both taper profiles and feeding techniques have
already been described in chapter 2 and further description of the linear taper profile and
microstrip feed can also be found in that chapter. This chapter will describe the design and
simulation processes of the wide band LTSA.
5.1 Features of LTSA
The LTSA generally has a lot of features including narrow beam width, high element gain
and wide bandwidth. A major disadvantage of the LTSA is that it requires either a
microstrip to slot transition or a coplanar waveguide to slot transition as part of its feeding
network. Due to this, the antenna design complexity increases and there is also a limit to
the wideness of the bandwidth of which the antenna can achieve.
5.2 Design Considerations
To design a wide band LTSA, it is important to obtain the required free space wavelength
and guide wavelength of the antenna. The effective dielectric constant is also required for
the guide wavelength. To obtain these parameters, it is recommended that section 3.3 of
Chapter 3 be referred to. In that section, equations are given to calculate these parameters.
Also, programs like PCAARD and slotline or microstrip calculators available on the
Internet can be use to obtain these parameters. The results from these programs tend to be
not as accurate. Thus, calculations through equations are still strongly recommended.
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Chapter 5: Design and Simulated Results of the Wide Band LTSA
40
5.3 Design on FEKO
Similar to the back-to-back microstrip to slot transition, the design for the wide band LTSA
was also done using FEKO. From the results observed in section 4.4 of chapter 4, we can
conclude that the simulated back-to-back transition works well. Thus, as mentioned before,
half of the transition is used for the wide band LTSA. A linearly taper profile is simply
attached to the end of the slot. Figure 5.1 shows how the top view of the wide band LTSA
would look like using the program WinFEKO.
Figure 5.1: Top View of Wide Band LTSA in WinFEKO
Since the measurements concerning the microstrip to slot transition of the antenna has
already been calculated in Chapter 4, all we need to do is alter some of the measurements
and include the dimensions of the linear taper profile. The overall dimensions for the wide
band LTSA are tabulated in the next page.
EllipticalCavity
MicrostripLine(bottom ofsubstrate)
14
Slot
TaperAngle
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Chapter 5: Design and Simulated Results of the Wide Band LTSA
41
Length of Antenna 133.5mm
Width of Antenna 61mm
Width of Slot, Ws 3mm
Length of Slot, Ls 25mm
Width of Microstrip Transmission Line, Wm 2.4mm
Length of Microstrip Transmission Line, Lm 20mm
Width of Quarter Wave Section, Wq 1.2mm
Length of Quarter Wave Section, L q 21.5mm
Radius of Elliptical Cavity 20mm
Taper Angle, 14
Table 5.1: Parameters for the Wide Band LTSA
Again, although the dimensions of the antenna can be calculated through given equations,
some of them are implemented into the design through the trial and error process. The
dimensions of the design should only be adjusted a little at a time when testing because
every little change matters when it comes to designing of antennas. The code written using
FEKO can be found in Appendix A.
5.4 Simulated Results
The wide band LTSA was simulated using FEKO and the results were plotted out using
GraphFEKO. For this particular antenna, the results taken into considerations were the
return loss, S11, and the radiation patterns of the E-plane and H-plane. The E-plane
represents radiation with respect to the vertical plane and the H-plane represents radiation
with respect to the horizontal plane. Similar to the back-to-back microstrip to slot
transition, the wide band LTSA was design to operate over a frequency of 1 GHz to 8 GHz.
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Chapter 5: Design and Simulated Results of the Wide Band LTSA
42
5.4.1 S11 Return Loss and Bandwidth
Figure 5.2: Simulated Return Loss, S11, of the Wide Band LTSA
Figure 5.2 shows the plot of the return loss of the wide band linearly tapered slot antenna as
obtained from GraphFEKO.
5.4.2 Radiation Patterns
The plots shown in figure 5.3 and 5.4 illustrate the E-plane and H-plane radiation patterns
obtained for four frequencies, 2.4 GHz, 3.38 GHz, 5.34 GHz and 6.46 GHz. Thesefrequencies were chosen because the return losses peak at that particular point. The peaks
can be observed from figure 5.2.
UsableBandwidth
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Chapter 5: Design and Simulated Results of the Wide Band LTSA
43
At 2.4 GHz
At 3.38 GHz
At 5.34 GHz
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Chapter 5: Design and Simulated Results of the Wide Band LTSA
44
At 6.46 GHz
Figure 5.3: Radiation Patterns for E-plane
At 2.4 GHz
At 3.38 GHz
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Chapter 5: Design and Simulated Results of the Wide Band LTSA
45
At 5.34 GHz
6.46 GHz
Figure 5.4: Radiation Patterns for H-plane
5.5 Discussion of Results
From the simulated return loss, S11, for the antenna, we are able to obtain a minimum
return loss of about 48dB at the resonant frequency of 6.46 GHz. A return loss of 10dB
can be obtained over the frequency of 3.25 GHz to 7.5 GHz. By referring to figure 5.2 and
using the equation (4.3), the usable bandwidth of the antenna is calculated to be
approximately 66%. Thus, the antenna can be considered to be wide band.
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Chapter 5: Design and Simulated Results of the Wide Band LTSA
46
As mentioned previously, the radiation patterns were taken at four different frequency
points of 2.4 GHz, 3.38 GHz, 5.34 GHz and 6.46 GHz with the bore sight of the antenna at
90. We will first discuss the radiation patterns taken with respect to the E-plane. At 2.4
GHz, a main lobe can be seen at the angle of 135. A couple of side lobes at the angles of
45 and 195 exist. The side lobes are negligible as they prove to be only very small. At
3.38 GHz, the main lobe can again be seen at the angle of 135. However, there are
considerable back lobes existing at 225 and 315 and a side lobe at 30. As the frequency
increases to 5.34 GHz, the main lobe at the bore sight is reduced significantly and is shifted
to the angle of 120. There are also more back lobes with a large back lobe appearing at the
angle of 240. At 6.46 GHz, we basically get the same radiation pattern as for 5.34 GHz
except there seems to be the existence of more back lobes but at a smaller scale. Judging
from the radiation patterns obtained at all four frequencies, we can conclude that more back
lobes are evident as the frequency increases. These extra lobes are undesirable because they
represent energy wasted from the antenna.
Looking at the radiation patterns on the H-plane, we can conclude the antenna eludes
almost symmetrical patterns. At a lower frequency of 2.4 GHz, we can observe that there is
a sufficiently large main lobe at the bore sight with no side lobes, although a couple of back
lobes do exist. As the frequency increases, one can observe the main lobe at the bore sightto have narrowed. However, there is an appearance of side and back lobes with
considerable gain appearing at the angles of 225 and 315 approximately. These are again
considered to be undesirable as there will be a loss of power from the antenna.
Judging from the discussion of the radiation patterns, the wide band linearly tapered slot
antenna tends to perform better at lower frequencies because there is less power loss due to
side and back lobes and thus the antenna can transmit a stronger signal. Normally, both the
free space and guide wavelengths are affected by frequency. An increase in frequency
would lead to a decrease in wavelength. Since antennas are design to operate better over a
given frequency, operating it at a different frequency requires a change in dimension as
well.
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Chapter 5: Design and Simulated Results of the Wide Band LTSA
47
5.6 Summary
This chapter provides a detailed design and simulation process of the wide band linearly
tapered slot antenna. The exact parameters used for the designing of the antenna in FEKO
were given with illustrations along the way.
From the results for return loss and bandwidth, the criteria set at the start of this project of
obtaining a wide band antenna was relatively achieved. By taking the vertical and
horizontal radiation patterns over four points from 1 GHz to 8 GHz, we were able to
observe the performance of the antenna and thus, can conclude that the antenna performs
better at lower frequencies.
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Chapter 6: Prototype and Measured Results of the Wide Band LTSA
48
Chapter 6: Prototype and Measured Results of the Wide Band LTSA
A prototype of the wide band LTSA was built with the help of the Electronics Laboratory
located at S309, Hawken Engineering Building. The design for the prototype was done in
Protel before submitting to the laboratory for fabrication. This chapter will provide a sketch
of the prototype but primarily focuses on the measured results obtained from the Network
Analyser located in room S507, Hawken Engineering Building.
6.1 Sketch of Prototype
A sketch of the prototype was done in order to have a visualisation on how the antenna
would look like if built. It also gives us an idea on how big the antenna would be and if the
design will come out as planned. The sketch of the prototype can be found in Appendix B
and looks similar to figure 5.1 of chapter 5.
6.2 Measured Results
Similar to the back-to-back microstrip to slot transition, the same process was taken to
design and fabricate the antenna on the same ROGERS dielectric substrate of 2.0 dielectric
constant and thickness of 0.5mm. This process proved to be as efficient as before with the
help of the personnel at the Electronics Laboratory. The top and bottom view of the
finished product can be seen in figure 6.1 and 6.2. From the figures, we can clearly see a
slot terminated by an elliptical cavity on one side while the other side opens up to a linear
taper with an angle of 14. A microstrip line can also be seen crossing the slot at the bottom
of the substrate. A SMA connector is soldered to the microstrip in order to excite it.
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Chapter 6: Prototype and Measured Results of the Wide Band LTSA
49
Figure 6.1: Top View of Finished Prototype
Figure 6.2: Bottom View of Finished Prototype
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Chapter 6: Prototype and Measured Results of the Wide Band LTSA
50
6.2.1 S11 Return Loss and Bandwidth
1 5 5.8 8 (GHz)
Figure 6.3: Measured Return Loss, S11, of the Wide Band LTSA
Figure 6.3 shows the measured return loss obtained through the network analyser. The x-
axis is scaled at 0.7 GHz per division and the y-axis is scaled at 10 dB per division.
6.2.2 Radiation Patterns
The radiation patterns, E-plane and H-plane, for the wide band linearly tapered slot antenna
was taken through testing in the Anechoic Chamber. A mount for the antenna had to be
built in order to securely attach the antenna to the rotational stand in the chamber. This
mount was built with the help of the Electrical Engineering Workshop. During testing, the
antenna radiates a signal sent from a source horn antenna. The different radiation planes
can be changed by rotating the LTSA or the source antenna 90 accordingly.
S11ReturnLoss(dB
)
UsableBandwidth0
-39
5.3
-10
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Chapter 6: Prototype and Measured Results of the Wide Band LTSA
51
Figure 6.4 illustrates the general orientation of the antenna from which the E-plane and H-
plane are taken. The orientations for the radiation patterns at all the frequencies are not
illustrated like in chapter 5 because the Network Analyser, unlike FEKO, is unable to
produce the sketches. The measured results for the radiation patterns were taken at the same
frequency points as the simulated results. Figures 6.5 and 6.6 show the measured results for
the radiation patterns for the E-plane and H-plane respectively.
Figure 6.4: General Orientation of Wide Band LTSA for E-plane and H-plane
90 90
270
180
270
0 180
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Chapter 6: Prototype and Measured Results of the Wide Band LTSA
52
Figure 6.5: Radiation Patterns for E-plane
Figure 6.6: Radiation Patterns for H-plane
9090
180 180
270270
00
00
9090
180180
270 270
0 0
270 270
180 180
90 90
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Chapter 6: Prototype and Measured Results of the Wide Band LTSA
53
6.3 Discussion of Results
By observing the measured return loss, S11, in figure 6.3, we are able to obtain a minimum
return loss of 39dB at the resonant frequency of 5.3 GHz. A return loss of 10dB can be
obtain over a frequency of 5 GHz to 5.8 GHz. Referring to figure 6.3 and applying equation
(4.3), the usable bandwidth can be calculated to be approximately 15%. From this, we can
conclude that the antenna can only work over a very narrow band.
Figures 6.4 and 6.5 show the radiation patterns for the E-plane and H-plane taken at the
frequencies of 2.4 GHz, 3.38 GHz, 5.34 GHz and 6.46 GHz. The bore sight of the antenna
was taken at 90. Again, the plots were taken from the Network Analyser.
Looking at the radiation pattern for the E-plane at 2.4 GHz, we notice a main lobe at the
angle of 45. There are minor small side and back lobes that may be neglected due to the
size of them. At 3.38 GHz, the main lobe has shifted towards the bore sight and there are
still the same side and back lobes. At the frequencies of 5.34 GHz and 6.46 GHz, a larger
main lobe is seen at the bore sight. However, more side and back lobes do exist.
For the radiation patterns obtained for the H-plane, we are able to notice almost
symmetrical patterns at all the frequencies. At all the frequencies, we are able to see a main
lobe at the bore sight that narrows as the frequency increases. However, as the frequency
increases, there are also side and back lobes appearing. These are apparent at the higher
frequencies of 5.34 GHz and 6.46 GHz with obvious side lobes around the angles of 0 and
180.
Judging from the plots for both the E-plane and H-plane taken at the four frequencies, we
can conclude that the wide band linearly tapered slot antenna works better at lower
frequencies. This is because the antenna is able to transmit a stronger signal, as there is less
power loss due to the appearance of side and back lobes.
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Chapter 6: Prototype and Measured Results of the Wide Band LTSA
54
6.4 Summary
From this chapter, we are able to draw a conclusion to the physical performance of the wide
band linearly tapered slot antenna. Although the measured return loss was not obtained as
what was expected, the radiation patterns were almost identical to what was predicted.
Experience was also gain in the form of hands on experience when it comes to the setting
up and operation of the Network Analyser and Anechoic Chamber. This might prove to be
useful in the working environment.
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Chapter 7: Evaluation
55
Chapter 7: Evaluation
This chapter will compare and evaluate all the simulated and measured results obtained for
the wide band LTSA. Judging from chapter 4, the simulated results of the back-to-back
microstrip to slot transition tend to differ a lot from the measured ones. Thus, this chapter
will discuss the similarities and differences of both sets of results.
7.1 Evaluation of Simulated and Measured S11 Return Loss and Bandwidth
The simulated and measured results of the S11 return loss and bandwidth can be seen in
figures 5.2 and 6.3 in chapters 5 and 6 respectively. Both results were taken over the same
frequency and were supposedly taken under the same design and physical considerations as
well. The comparisons will be made on the minimum return loss, resonant frequency and
usable bandwidth. These are tabulated below.
Simulated Measured
Minimum Return Loss -48dB -39dB
Resonant Frequency 6.46 GHz 5.3 GHzUsable Bandwidth 66% 15%
Table 7.1: Compiled Simulated and Measured Results
Judging from the tabulated results, we can conclude that the antenna designed in FEKO has
a better return loss of 48dB and operates at a higher resonant frequency of 6.46 GHz. The
physical design has its resonant frequency shifted down to 5.3 GHz and a slightly worse
return loss of 39dB. The change in the return loss and resonant frequency for the physical
design could be due to the fabrication and testing process.
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Chapter 7: Evaluation
56
As mentioned before, the fabrication of antenna was done by submitting a Protel design to
the Electronics Laboratory. Although the design done through Protel should produce an
antenna with exact parameters, it is still bound to have a slight percentage of error. For
example, the Keep Out Layer in Protel is formed by having a border of certain thickness.
This thickness is bound to overlap into the actual dimensions of the antenna, thus making
the length or the width slightly off the required measurements. A slight change in
dimensions, be it 0.1mm, will make a difference in the output of the antenna. This, together
with reasons mentioned in section 4.3 of chapter 4 concerning the testing of the antenna,
tends to have at least some effect on performance of the antenna.
7.2 Evaluation of Simulated and Measured Radiation Patterns
Radiation patterns of the wide band LTSA helps us understand how much signal is being
transmitted and in which direction. Comparisons will be made on the E-plane and H-plane
of both the simulated and measured results to see if the antenna designed in FEKO
performs similarly as to the one built.
By observing figure 5.3 of chapter 5 and 6.5 of chapter 6, we are able to draw a comparison
between the simulated and measured radiation patterns for E-plane. Unlike the difference in
the return losses, the radiation patterns for both the simulated and measured results seem to
have almost identical patterns at the bore sight of 90. The main lobe, taken at all the
frequencies, are all facing the same direction although there seems to be the existence of
more side and back lobes for the measured results. Also, there seems to be a lot of noise in
the measured results but this is due to the reflections off objects such as the rotational stand
or the source antenna in the Anechoic Chamber. Having absorbers located around this
objects can reduce this noise but still, not all the noise can be subdued.
Referring to figures 5.4 and 6.6 of chapter 5 and 6 respectively, we are also able to
conclude that the simulated and measured radiation patterns for the H-plane produce almost
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Chapter 7: Evaluation
57
similar outcomes. All the main lobes for both sets of results are pointing at the bore sight of
90.The only difference is that the main lobe of the simulated results tends to be wider and
larger. There are generally the same side and back lobes located in the same direction.
There is again noise similar to those obtain for the E-plane.
7.3 Summary
This chapter enables us to compare both sets of radiation patterns obtained for the wide
band linearly tapered slot antenna. From the comparisons made, we can conclude that the
actual antenna performs according to the design specification made in FEKO. However, a
great difference in bandwidth is apparent where the simulated results obtained a 66%
bandwidth whereas the measured ones only had a 15% bandwidth. This shows the actual
antenna can operated well only over a narrow bandwidth and thus, not meet the criteria of
the antenna being wide band. Steps that can be taken to improve this will be mentioned in
section 8.1 of chapter 8.
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Chapter 8: Conclusion
58
Chapter 8: Conclusion
The main focus of this thesis was to successfully design and construct a Wide Band
Linearly Tapered Slot Antenna and at the same time obtain almost satisfactory results. An
important objective was to ensure that the simulated results of the antenna done in FEKO
and the measured results obtain from the actual testing of the antenna produce almost
similar outcomes.
On the whole, the antenna was completed on time. The simulated usable bandwidth of the
antenna was calculated to be 66% while the measured usable bandwidth was calculated to
be only 15%. Therefore, the simulated results met the criteria of obtaining a wide band for
the antenna although the measured results did not. The radiation patterns for both thesimulated and measured results were almost the same to prove that the signal of the antenna
in FEKO and actual antenna radiates the signal in similar directions.
Upon the completion of this thesis, all the aims and objectives made at the start of the
project were met. The wide band linearly tapered slot antenna was completed on time and a
great deal was learnt about the program FEKO. Finally, knowledge on the microwave field
and travelling wave antenna was gain and this will be an invaluable asset for the future.
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Chapter 8: Conclusion
59
8.1 Future Work
Although this thesis was considered to be a minor success, several factors of the wide band
linearly tapered slot antenna can still be improved. As we have seen during the course of
this project, the simulated and measured results were not similar at times and it was also
difficult to obtain a wide bandwidth for the antenna. Also, due to time constrains, different
designs of the antenna were researched on but further work was not carried out.
Listed below are three factors that should be research on and implemented to ensure better
performance and results if future work was to be done on this thesis.
Implementation of the Double Y-balun
When designing the antenna, one of the major problems was the microstrip to slot
transition that is crucial in obtaining a wide bandwidth for the antenna.
Implementing a double y-balun by following closely to the steps mentioned in
section 4.1 of chapter 4 would result in the bandwidth of the antenna improving a
great deal.
Changing of parameters to observe differences in results
Design and construction of the wide band linearly tapered slot antenna was done on
a fixed set of parameters. Although satisfactory results were obtained, one has to
wonder how different the results would have been if some of the parameters were
changed. Therefore, comparisons of results should be made on the same type of
antenna with a series of different parameters.
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Chapter 8: Conclusion
60
Measurement of discontinuities of the antenna
When comparing the simulated and measured return loss of the antenna, the
simulated result obtained a much better return loss and bandwidth than the
measured result. The reason behind this can be found out by measuring the
discontinuities of the antenna in the time domain through the Network Analyser.
By experimenting and implementing the recommended future work, this thesis project can
be use as a base for the next student to build a solid foundation on. Hence, this will ensure
that an improved wide band linearly tapered slot antenna will perform better and exceed
more requirements.
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Appendix A
61
Appendix A
FEKO code for Quarter Wave Microstrip Transformer
** Scaling factor since all dimensions below in mm
SF 1 0.001
#h = 1.0
#epsr = 2.0 ** Relative permittivity
#effepsr = 1.983 ** Effective permittivity
#freq = 8.0e9
#lam = 1000 * #c0 / #freq / sqrt(#effepsr)
#x = 20
#lam1 = #lam/4 + #x
#a = 1.20
#b = 2.40
** Segmentation parameters#tri_len = #lam / 12
#fine_tri = #lam / 16
IP 0 #tri_len
** Define Points
DP A 0 0 #h
DP B 0 #x #h
DP C #a #x #h
DP D #a 0 #h
DP E #a 0 #h
DP F #a #x #h
DP G #b #x #h
DP H #b 0 #h
DP 1 0 #x #h
DP 2 0 #lam1 #h
DP 3 #a #lam1 #h
DP 4 #a #x #h
** Microstrip Feed
LA 1
BP A B C D #fine_tri #fine_tri
LA 2
BP E F G H #fine_tri #fine_tri
LA 3
BP 1 2 3 4 #fine_tri #fine_tri
** End of geometry input
EG 1 0 0 0 0 1 1GF 10 1 0 1 1 0
#h
1.0 #epsr 1 0.001
LE 2 3 3 100 0
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Appendix A
62
** Frequency
FR 51 0 1.0e9 #freq
** Voltage source at the wire centre with impressed power
PW 3 0 1.0 50 0
AE 0 A H 3 1.0 0 0
** Far-field pattern
FF 1 181 1 1 0 0 2
** Far-field pattern
FF 1 181 1 1 0 90 2
** End
EN
FEKO code for Back-to-Back Microstrip to Slot Transition
** Scaling factor since all dimensions below in mm
SF 1 0.001
#h = 0
#t = -0.5
#epsr = 2.0
#freq = 8.0e9
#lam = 1000 * #c0 / #freq / sqrt(#epsr)
#x = -45
#y = -65
#z = -66
#xx = 45
#yy = 65#zz = 66
#a = 78
#b = 81
#c = 90.5
#d = 110
#1 = -20.5
#2 = -18.1
#3 = 20.5
#4 = 18.1
#5 = -19.3
#6 = 19.3
#L = 49
#tl = #lam/4
#tl1 = min(#tl, 2) ** Comparable to slot width
#tl2 = #tl/2
IP 0 #tl
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Appendix A
63
** Define Points of Slotline
DP A -25 #L #h
DP B -25 #a #h
DP C 0 #a #h
DP D 0 #L #h
DP E -25 #b #h
DP F -25 #d #h
DP G 0 #d #h
DP H 0 #b #h
** Define Points of Microstrip
DP 1 #1 #l #t
DP 2 #1 69 #t
DP 3 #5 69 #t
DP 4 #5 #l #t
DP 1.1 #5 #l #t
DP 2.1 #5 69 #t
DP 3.1 #2 69 #t
DP 4.1 #2 #l #t
DP 1.2 #1 69 #t
DP 2.2 #1 #c #t
DP 3.2 #5 #c #t
DP 4.2 #5 69 #t
DP 11 #1 #L #h
DP 12 #1 #d #h
DP 13 #2 #d #h
DP 14 #2 #L #h
DP 22 #1 #a #h
DP 23 #2 #a #h
DP 32 #1 #b #h
DP 33 #2 #b #h
** Elliptical Open Circuit
DP I #x #L #h
DP J #x #a #h
DP K #y #L #h
DP L #y #a #h
DP M #x #b #h
DP N #x #d #h
DP O #y #b #h
DP P #y #d #h
DP Q #z #L #h
DP R #z #a #h
DP S #z #b #h
DP T #z #d #h
DP M1 #x 101 #h
** Define Points of Slotline2
DP AA 0 #L #h
DP BB 0 #a #h
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Appendix A
64
DP CC 25 #a #h
DP DD 25 #L #h
DP EE 0 #b #h
DP FF 0 #d #h
DP GG 25 #d #h
DP HH 25 #b #h
** Define Points of Microstrip2
DP 1a #3 #l #t
DP 2a #3 69 #t
DP 3a #6 69 #t
DP 4a #6 #l #t
DP 1a1 #6 #l #t
DP 2a1 #6 69 #t
DP 3a1 #4 69 #t
DP 4a1 #4 #l #t
DP 1a2 #3 69 #t
DP 2a2 #3 #c #t
DP 3a2 #6 #c #t
DP 4a2 #6 69 #t
DP 11a #3 #L #h
DP 12a #3 #d #h
DP 13a #4 #d #h
DP 14a #4 #L #h
DP 22a #3 #a #h
DP 23a #4 #a #h
DP 32a #3 #b #h
DP 33a #4 #b #h
** Elliptical Open Circuit2
DP Ia #xx #L #h
DP Ja #xx #a #h
DP Ka #yy #L #h
DP La #yy #a #h
DP Ma #xx #b #h
DP Na #xx #d #h
DP Oa #yy #b #h
DP Pa #yy #d #h
DP Qa #zz #L #h
DP Ra #zz #a #h
DP Sa #zz #b #h
DP Ta #zz #d #h
DP M1a #xx 101 #h
LA 0
PH J L K I L #tl1
PH M N P O M1 #tl1
PH J B A I B #tl1
PH M E F N E #tl1
BP L R Q K
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Appendix A
65
BP P T S O
BP O S R L #tl1
PM A 11 22 B
#tl2 #tl1
BP 22 11 14 23 #tl2 #tl1
PM 23 14 D C
#tl2 #tl1
BP 32 33 13 12 #tl1 #tl2
PM 32 12 F E
#tl2 #tl1
PM 13 33 H G
#tl2 #tl1
LA 1
BP 1 2 3 4 #tl2 #tl1
BP 1.1 2.1 3.1 4.1 #tl2 #tl1
BP 1.2 2.2 3.2 4.2 #tl2 #tl1
** Add feed strip
BP 1 4.1 14 11 #tl1 #tl1
LA 2
PH Ja La Ka Ia La #tl1
PH Ma Na Pa Oa M1a #tl1
PH Ja CC DD Ia CC #tl1
PH Ma HH GG Na HH #tl1
BP La Ra Qa Ka
BP Pa Ta Sa Oa
BP Oa Sa Ra La #tl1
PM AA 14a 23a BB
#tl2 #tl1
BP 22a 11a 14a 23a #tl2 #tl1
PM 22a 11a DD CC
#tl2 #tl1
BP 32a 33a 13a 12a #tl1 #tl2
PM 33a 13a FF EE
#tl2 #tl1
PM 12a 32a HH GG
#tl2 #tl1
** Add feed strip
BP 1a 4a1 14a 11a #tl1 #tl1
LA 3
BP 1a 2a 3a 4a #tl2 #tl1
BP 1a1 2a1 3a1 4a1 #tl2 #tl1
BP 1a2 2a2 3a2 4a2 #tl2 #tl1
** End of geometry input
EG 1 0 0 0 0
GF 11 2 0 1 1
#h
0.5 #epsr 1 0.001
1 1
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Appendix A
66
** Frequency
FR 51 0 1.0e9 #freq
** Voltage source at the wire centre with impressed power
PW 1 1.0
AE 0 0 1 0 1 0
LE 2 3 0 50
OS 1 1
** Far-field pattern
FF 1 181 1 1 0 0 2
** Far-field pattern
FF 1 181 1 1 0 90 2
** End
EN
FEKO code for Wide Band Linearly Tapered Slot Antenna
** Scaling factor since all dimensions below in mm
SF 1 0.001
#h = 0
#t = -0.5