Power Electronic Converter for 30kw,270v Dc

Detailed Power Converter Design for a 30kW Switched Reluctance Starter/Generator Used in Aircraft

Song Shoujun1,2 , Liu Weiguo1, Uwe Schaefer2

1School of Automation Northwestern Polytechnical University

Xi’an 710072, China E-Mail: [email protected]; [email protected]

URL: http://www.nwpu.edu.cn/en/9.html

2Institute of Energy and Automation Technical University of Berlin

D-10587 Berlin, Germany E-Mail: [email protected]

URL: http://www.iee.tu-berlin.de/

Keywords «Airplane», «Switched reluctance drive», «Converter circuit», «Design», «Thermal design»

Abstract Some general considerations for the design of the power converter used in an undergoing 30kW

switched reluctance starter/generator system are presented. The most suitable converter topology and power semiconductor are found firstly based on features of the application and comparisons among many possibilities. Then, the frequency stress, voltage ratings, peak and rms current ratings of the power semiconductor are calculated analytically and verified by the simulation results with PC-SRD. Finally, detailed thermal analysis of the converter is presented to check the applicability of the adopted power semiconductor products and select feasible heatsink.

Introduction Many studies have indicated that More/All Electric Aircraft (M/AEA) technology is the future

development trend of the aerospace industry and Integral engine Starter/Generator (IS/G) is a key subsystem of this novel technology[1-3].

Among kinds of machines that can be used as an IS/G in M/AEA, the Switched Reluctance Starter/Generator (SR S/G) is considered as a prime candidate technology to meet the requirements and constraints well. In [4], the design, implementation and test validation of a switched reluctance starter/generator system for aircraft engine application are presented.

During the operation of SR S/G system, power converter receives power from the power bus during engine starting and provides power to the bus during generating. It is a critical component of the SR S/G system and its design is directly related to the performance and cost of the whole system.

In this paper, the design process of a power converter used in an undergoing 30kW SR S/G system is considered in detail from four aspects: Converter topology and power semiconductor selection, frequency stress calculation, current and voltage rating estimation and thermal analysis.

The organization of this paper is as follows. In Section II, the performance requirements of the undergoing SR S/G system are described. The selections of the converter topology and power semiconductor are discussed in Section III. The detailed calculation of frequency stress, voltage and current ratings are presented in Section IV and V, respectively. Section VI shows the thermal analysis of the power converter. Finally, conclusions are drawn in Section VII.

Authorized licensed use limited to: INDIAN INSTITUTE OF TECHNOLOGY MADRAS. Downloaded on August 17,2010 at 04:21:34 UTC from IEEE Xplore. Restrictions apply.

Performance Requirements of the System The designed SR S/G is a 270-Vdc system that performs two primary functions. The first function

(0-27000rpm) is to start a gas turbine engine with at least 15Nm torque using an available source of 270-Vdc power. The second function (27000rpm-50000rpm) is to extract power from the gas turbine engine to generate 30kW 270-Vdc. Both the starting requirement and generating requirement are summarized by the torque speed curve in Fig.1. It should be noted that the SR S/G is connected to the aircraft engine through a step-down gearbox. The Switched Reluctance Machine (SRM) turns at a higher speed than the aircraft engine. The abscissa in Fig.1 represents the shaft speed of the SRM; it is three times as fast as that of the engine. By the way, the SRM speed 27000rpm and 50000rpm correspond to the flight idle and maximum speed of the aircraft engine respectively.

0 5 10 15 20 25 30 35 40 45 50-20

-10

0

10

20

generatingstarting

30kW

15Nm

Fig.1. Performance requirements indicated by torque speed curve

Selections of Converter Topology and Power Semiconductor With the development of technology and research, many kinds of converter topologies and power

semiconductors exist for SR S/G applications. All of them have their own merits and faults. Finding out the most suitable option is the foremost work for the design of the power converter.

Converter topology selection

In [5], a large number of power converter topologies developed for Switched Reluctance Drives (SRD) were divided into six basic categories by the means in which the energy stored in the magnetic field associated with the phase winding is recovered at the end of each stroke.

The converter semiconductor VA rating gives a useful rule-of-thumb of the total drive cost, since the semiconductor devices comprise much of the variable cost in a SRD. In [5], a direct initial comparison of the related converter topologies was made in terms of semiconductor component minimum VA ratings. According to the results of the comparison, there are seven kinds of topologies have relatively low VA ratings which are: Asymmetric half-bridge circuit, Shared switch circuit, Split DC circuit, Bifilar circuit, Dissipative circuit, Triac circuit, and H-Bridge circuit. The most suitable topology to the designed SR S/G should be generated among them.

There are many special features and requirements in the designed SR S/G system: 1. The number of phase windings is low and there is a strong possibility that the current overlap will

exist; As mentioned above, a 3-phase 6/4 pole SRM is adopted. To enhance the output performance and

reduce the torque ripple, it’s possible to excite two adjacent phases synchronously. 2. The supply voltage is 270V, it’s relatively high;

The High Voltage DC concept is generally accepted as the developmental tendency in the aircraft electric power system, so this system adopts 270Vdc as well. 3. High power and efficiency requirements;

The undergoing system is designed to provide 30kW electrical power to the engine and other electrical loads in the aircraft. What’s more, the system will get its maximum power, about 42.4kW, at 27000rpm under motoring. 4. High reliability requirement

Torq

ue (N

m)

Speed (krpm)


Because the converter is used in the aircraft, so its reliability is a critical issue. In recent years, many topologies are published to further improve the performance of whole system, but most of them are very complex in structure and control method. What’s more, they need more active and passive devices to implement their functions; this will further reduce the reliability of the system.

The final choice of the topology should be based on the characteristics and requirements of the designed system discussed above. Two examples are given as follows:

The Shared switch topology can reduce the number of devices required per phase winding. However, this will cause higher voltage, current, or power dissipation ratings in common switches or/and diodes. Particularly, an increase in current rating may dramatically increase the package cost beyond any savings that may have been made from a reduced number of switches [5]. This disadvantage becomes more evident along with the decrease of the phase numbers. To most of the Shared switch topologies, current overlap in the conventional sense is not allowed. To avoid current overlap, the dwell period should be narrowed; this will lead to poor converter and motor utilization. What’s more, the common devices may decrease the reliability of the system. So the Shared switch topology is not the most suitable one.

In Bibilar circuit, the energy stored in the magnetic field of the phase winding can be transferred to a closely coupled second winding. From there, the energy can be returned to the dc link or used to energize another phase winding. The voltage rating of the switches and diodes in this topology is twice the supply voltage [5]. If the supply voltage is relatively high, the selection of the appropriate devices is more difficult. The perfect coupling is impossible in such systems, so snubber circuits are required which will reduce the reliability of the system. What’s more, the additional winding will reduce the copper area available to the principle winding which will lead to an increased phase resistance. Consequently, the copper losses will be increased and the efficiency of the system will be reduced. So the Bibilar circuit also isn’t suitable.

Finally, based on the above and further analysis, the Asymmetric half-bridge circuit is adopted which is shown in Fig.2.

Phase Diode

Phase Diode Phase Switch

Phase Switch

Phase Winding

Fig.2. Asymmetric half-bridge converter circuit (one phase of three)

Power semiconductor selection The power semiconductors used in the converter of SR S/G system must have the ability to turn-off

current, so the thyristor can’t be used since it is a latching device which is turned on by a gate pulse and then remain conducting until the current falls to zero. In order to reduce the size and weight of the system, the gate drive requirements of the switch should be minimized. Hence, only MOS-gated semiconductors need to be considered. This eliminates bipolar darlingtons and gate turn-off thyristors. According to the analysis above, there are only three kinds of power semiconductors need to be considered: MOS field-effect transistor (MOSFET), the insulated-gate bipolar transistor (IGBT) and the MOS-controlled thyristor (MCT).

According to the analysis in [6], the MOSFET can be eliminated due to its large silicon area and great forward drop and consequently high conduction losses. Compared with the MOSFET and IGBT, the MCT has the smallest silicon area, the lowest forward drop, and thus the lowest power dissipation, it seems to be the best choice for high power applications. However, the IGBT is also usable for high power applications, even though its performance is second to that of the MCT’s. There are many specifications of IGBT covering a very wide range of power levels; it’s relatively easy to find an appropriate product. What’s more, the IGBT is much cheaper than the MCT. Hence, the IGBT is adopted for the undergoing system finally.


Frequency Stress Calculation The maximum allowable switching frequency of an IGBT has progressively increased. It is limited

by the maximum heat it can dissipate before the junction temperature reaches rated temperature under specific thermal conditions.

The purpose of this section is to calculate the maximum switching frequency of the switches in the power converter through out the whole speed range, and thus lay a foundation for the selection of IGBT and the calculation of the its power losses.

As mentioned above, the SRM will operate under two modes: motoring and generating. Under motoring, the power converter will operate in the Current Chopping Control (CCC) mode. Under generating, it will operate in Angle Position Control (APC) mode. So the maximum switching frequencies under these two operating modes need to be calculated respectively to achieve the final result.

Maximum switching frequency under motoring During system operation under motoring, the calculation of the converter switches chopping

frequency can be started from the following two voltage equations,

bkph

phdc Edt

diLV += (1)

bkph

ph Edt

diL +=0 (2)

where, Vdc is the converter dc link voltage, Lph is the phase inductance, iph is the phase current, and Ebk is the back emf. It should be noted that the phase resistance is neglected in (1) and (2).

When the two switches in one phase are turned on, the phase is excited by Vdc and its current will rise. The time for the current to rise by the hysteresis current band value can be obtained by integration of (1),

bkdc

hysphr EV

iLT

−= (3)

where, Tr is the rise time of the current, ihys is the hysteresis current band value. To minimize the converter’s chopping frequency, freewheel chopping is adopted. When the phase

current reaches the upper boundary of the hysteresis band, only one switch of this phase is turned off, the phase winding is shorted, and the phase current will be driven down by the back emf. The time for the current to decrease by the hysteresis current band value can be obtained by integration of (2),

bk

hysphf E

iLT = (4)

where, Tf is the fall time of the current. Based on (1) and (2), the phase chopping frequency can be calculated as follows,

( )dchysph

bkdcbk

frch ViL

EVETT

F −=

+=

1 (5)

According to [7], the back emf can be written as

0ωω

dcbk VE = (6)

where, ω is the speed of the machine and ω0 is the rated speed. For IS/G machine ω0 is approximately flight idle speed which is 27000rpm in the system being

developed. Using (6), (5) can be rewritten as

⎟⎟⎠

⎞⎜⎜⎝

⎛−=

00

1ωω

ωω

hysph

dcch iL

VF (7)

However, (7) is not the final equation. The real quantity of interest is the average chopping frequency since this is what produces temperature rise in the inverter. In practice, the converter does not chop continuously but rather only when the rotor poles are in the correct angular position relative to the stator poles. According to this, (7) can be rewritten assuming that the converter chops for a fraction of the phase period equal to α [7].


⎟⎟⎠

⎞⎜⎜⎝

⎛−=

00

1ωω

ωωα

hysph

dcchavg iL

VF (8)

Here, α can be calculated as follows, ( )

πθθ

α2

onoffrN −= (9)

where, Nr is the number of rotor poles, θoff and θon are the turn-off and turn-on angle respectively. From (8), it can be found that the average chopping frequency will reach its maximum value when

the rotor speed is equal to one-half of slight idle speed. Based on the analysis above, it can be concluded that the switches used in the power converter of

the undergoing system will reach their maximum switching frequency 8.971kHz at the speed of 13500rpm with 0.5 for α, the unaligned inductance (0.0001254mH) for Lph, 30A for ihys and 270V for Vdc.

Fig.3 shows the switching frequency curve of the IGBT versus the machine speed under motoring obtained by PC-SRD [8]. It can be found that the IGBT will get its maximum switching frequency 8.667kHz at the speed of 13000rpm. It verified the above analytical calculations well.

0 5 10 15 10 252

3

4

5

6

7

8

9

speed (krpm)

switc

hing

freq

uenc

y (k

Hz)

25 30 35 40 45 50

1.5

2

2.5

3

3.5

4

speed (krpm)

switc

hing

freq

uenc

y (k

Hz)

Fig.3. Switching frequency under motoring Fig.4. Switching frequency under generating

Maximum switching frequency under generating Under generating, the power converter will operate in square wave mode. The semiconductor

switches will switch at the phase frequency of the SRM which can be obtained as follows,

rph NF60ω

= (10)

Because the maximum speed of the SRM is 50000rpm, the maximum phase frequency is 3.33kHz. This also can be verified by PC-SRD.

Fig.4 shows the switching frequency curve of the IGBT versus the machine speed under generating obtained by PC-SRD. It can be found that the maximum switching frequency of IGBT is 3.33kHz, it occurs at 50000rpm.

Based on the above calculations, it’s clear that the maximum switching frequency of the power converter is 8.667kHz, it occurs at the speed of 13000rpm under motoring.

Voltage and Current Ratings Estimation Voltage and current ratings are important criterions for the selections of IGBT and diode modules.

Voltage ratings According to the adopted converter topology, both the forward blocking voltage of the IGBT and

the reverse voltage of the diode are equal to the converter dc link voltage Vdc which is 270V in the considered application. In power converter, there are voltage spikes due to line or stray inductance, so a modest safety margin should be taken into account. The voltage rating can be obtained as

Vrat ≈Vdc + 300=600V (11)

Current ratings Two types of current ratings have to be observed for IGBTs and diodes: the pulse current rating and

the continuous current rating. The pulsed current rating has to be at least equal to the maximum peak


current of the component. The continuous current rating has to be equal to or higher than the maximum rms current of the component. Hence, the peak and rms currents of the IGBT and diode should be estimated under both motoring and generating mode.

Under motoring, the peak currents of the IGBT and diode can be determined directly, they are equal to the prescribed upper limit of the current chopping which is 260A in the considered system.

The peak current ratings under generating can be determined easily by analytical method. Fig.5 shows the phase current curves versus rotor position under generating.

0 20 40 60 80 100

0

50

100

150

200

250

300

rotor position (deg.)

phas

e cu

rrent

(A)

phase current 1phase current 2phase inductance

27 35 40 45 500

50

100

150

200

250

300

speed(krpm)

peak

pha

se c

urre

nt (A

)

PC-SRDanalytical method

Fig.5. Phase current under generating Fig.6. Comparison between PC-SRD and analytical method

It can be found that there are two situations which are represented by “phase current 1” and “phase current 2” respectively. In the first situation, the machine speed is not high enough, the back emf is smaller than the applied dc link voltage, and the phase current will fall down after the turn-off angle θoff. Hence, the peak current occurs at θoff. In the second situation, the machine speed is high, the back emf is bigger than the applied dc link voltage, and the phase current will keep increasing until θd where the back emf becomes zero due to the constant phase inductance. Hence, the peak current occurs at θd.

The current at θc and θd can be calculated by (12) and (13) respectively, which are deduced from the linear model of SRM.

( )( )[ ] ( )offc

c

ondcph kL

Vi θθθθθω

θθ≤≤

−−−

= ,max

(12)

( )( )[ ] ( )doff

c

onoffdcph kL

Vi θθθ

θθωθθθ

≤≤−−−−

= ,2

max

(13)

where, Lmax is aligned position inductance, θ is rotor position, θon is turn-on angle and k is the slope of the phase inductance.

According to (12) and (13), the maximum peak phase current 278.1A will occur at 27000rpm which is the start point of the generating mode. According to the simulation results with PC-SRD, the maximum peak phase current is 268.3A, it also occurs at 27000rpm. Fig.6 shows the comparison between the results of the analytical calculation and simulation. It can be seen that the analytical method is well verified, the error is due to the linearization of the SRM model. What’s more, it can be found that the peak phase current at 27000rpm occurs at θoff (the first situation). Hence, both the maximum peak current of IGBT and diode are equal to that of the phase.

According to above calculations, the peak current ratings for both IGBT and diode are 268.3A. Considering appropriate safety margin of 1.5～2,

Ipeak=(1.5～2)×268.3=402.45～536.6A (14) According to (14), the IGBT modules in the 400A or 600A rated current class should be considered. The rms current ratings of IGBT and diode also can be calculated by analytical method. For

example, the analytical determination of the IGBT rms current under motoring is given as below. Fig.7 shows the simulated current curve of IGBT under motoring by PC-SRD. If θon is considered as

the zero point of the rotor position and current chopping is neglected, the rms current of the IGBT can be easily obtained by its definition.

⎥⎥⎦

⎤

⎢⎢⎣

⎡+⎟⎟

⎠

⎞⎜⎜⎝

⎛= ∫ ∫

a off

ach

a

chrrms dIdINI

θ θ

θθθθ

θπ 0

2

2

2 (15)

where, Ich is the upper limit of the current chopping which is 260A in this paper, θa is the rotor position where the current reaches the upper limit of the current chopping for the first time, it can be given by

θoff θd θc


dc

cha V

IL ωθ min= (16)

where, Lmin is the unaligned position inductance. Solving (15), it is obtained,

⎟⎠⎞

⎜⎝⎛ −= aoff

rrms

NI θθπ 3

22

(17)

-20 0 20 40 60 80 1000

50

100

150

200

250

300

rotor position (deg.)

IGB

T cu

rrent

(A)

phase currentphase inductance

0 5 10 15 20 25

0

50

100

150

200

250

300

350

400

speed(krpm)

IGB

T rm

s cu

rrent

(A)

PC-SRDanalytical method

Fig.7. IGBT current under motoring Fig.8. IGBT rms current comparison

The IGBT rms current calculated by (17) and obtained by PC-SRD simulation are compared in

Fig.8. It shows a very good match between the two sets of results. Current chopping is neglected in the derivation of (17), so the calculation results are slightly bigger than that obtained by PC-SRD.

According to Fig.8, the maximum IGBT rms current is 176.2A which occurs at 1000rpm. By PC-SRD simulation, it’s found that the maximum rms current of diode is equal to 173.8A, and it also occurs at 1000rpm.

Further simulations by PC-SRD indicate that the rms currents of IGBT and diode under generating are much smaller than those under motoring, so they are not considered here.

Based on the voltage and peak current ratings given above, the IGBT module SEMiX603GAL066HDs and its respective GAR type are adopted, they are produced by Semikron. Fig. 9 shows the circuit diagrams of these two modules.

a) SEMiX603GAL066HDs b) SEMiX603GAR066HDs

Fig.9. Circuit diagrams of the selected IGBT modules

It can be seen that both of them consist of IGBT, freewheeling diode and inverse diode. Compared with Fig. 2, it’s clear that one SEMiX603GAL066HDs and one SEMiX603GAR066HDs can constitute one phase of the converter. To construct the whole converter, three SEMiX603GAL066HDs modules and three SEMiX603GAR066HDs modules are needed. A photograph of the adopted IGBT module is given in Fig.10.

Fig. 10. Photograph of the IGBT module

θoff θa θon

Freewheeling diode

Inverse diode IGBT Freewheeling

diode

Inverse diode IGBT

DriverEvaluation board IGBT


Thermal Analysis There are two purposes of the thermal analysis presented in this section: Checking the thermal

applicability of the adopted power semiconductor products and selecting feasible heatsink. As we know that the temperature rise is caused by the power losses. So, before the thermal analysis

of the converter, the power losses in it must be calculated firstly. The power losses can be obtained in two ways: The first one is called instantaneous power losses

calculated by the product of the real time values of voltage and current. The second one is called average power losses.

In the considered converter, the power losses are discontinuous. This will cause the fluctuation of the junction temperature. It’s necessary to evaluate this fluctuation based on the instantaneous power losses.

Fig.11 shows △Tjc, the transient temperature rise between the junction and the case, of the IGBT and the diode at their worst-cases of heat dissipation respectively. It can be seen that the temperature fluctuation is much smaller than the steady-state temperature rise, and thus negligible. So in this paper, the average power losses are adopted.

0 0.1 0.2 0.3 0.4 0.50

5

10

15

20

25

time (s)

IGBT @ 19000rpm

Diode @ 8000rpm

5 10 15 20 25

0

50

100

150

200

250

300

350

speed(krpm)

pow

er lo

ss (W

)

Ptotal-IGBTPtotal-diode

worst-case

Fig.11. △Tjc of the IGBT and diode Fig. 12. Average power losses of the IGBT and diode

Fig. 12 shows the average power losses of the IGBT and diode at several speeds. According to simulation and analysis, the worst-case of heat transfer occurs under motoring mode, so Fig.12 only shows the power losses under motoring mode.

Thermal applicability of the selected IGBT Besides the constraints on voltage and current ratings, the thermal considerations are equally

important in the selection of the power semiconductors. Under the maximum power losses, the junction temperature of the IGBT and diode mustn’t reach or exceed the maximum allowable value given by the datasheet.

The junction temperature of the IGBT and diode can be easily estimated by cjcj TTT +Δ= (18)

where, Tj is the junction temperature, ΔTjc is the temperature rise between the junction and the case, Tc is the temperature of the case. ΔTjc can be calculated by

)( cjthxtotaljc RPT −−=Δ (19) where, Ptotal-x is the average power losses (can be change to Ptotal-IGBT or Ptotal-diode as shown in Fig. 12) and Rth(j-c) is the thermal resistance from junction to the case which can be directly got from the datasheet.

According to Fig.12, the maximum power loss of the IGBT in SEMiX603GAL066HDs module is 296.2W which occurs at 19000rpm, and the thermal resistance from junction to case is 0.087K/W. By applying these to (19), it can be found that the temperature rise between the junction and the case is 25.8K. If the temperature of the case is assumed to be equal to the ambient temperature which normally is 313K (40oC), the junction temperature will reach 338.8K (65.8oC). The junction temperature of the diode in SEMiX603GAL066HDs module can be got by the same process, the result is 334.7K (61.7oC). These junction temperatures are much smaller than the maximum allowable value, 423K (150oC), got from the datasheet, so SEMiX603GAL066HDs and its respective GAR type can be used in the considered power converter.

△T j

c (K

)


Selection of the heatsink In order to ensure the heat dissipation ability of the converter, the thermal resistance of the selected

heatsink should be considered carefully. Normally, there is a limitation on the thermal resistance value of the heatsink.

In the considered power converter, all the IGBTs and diodes are mounted on the same heatsink. Fig.13 shows the complete transient thermal model of the converter. There are six identical power modules, Module1 to Module6, mounted on the same heatsink, the details of Module1 are shown. The thermal models of IGBT, Diode, Interface and Heatsink in Fig.13 are so called π -thermal model [9], the values of the thermal resistances and thermal capacitances can be extracted from the datasheet or provided by the related manufacturers. Ptotal-IGBT and Ptotal-diode are the average power losses of IGBT and diode as shown in Fig. 12. Ta is the ambient temperature.

The normal maximum allowable junction temperature for IGBT and diode is 423K (150oC). In this project, 30K is considered as safety margin, so the maximum junction temperature should not exceed 393K (120oC).

RDth1

CD1Ptotal-diode

RDth2 RDthn

CD2 CDn

RTth1

CT1Ptotal- IGBT

RTth2 RTthn

CT2 CTnRch 1

Cch 1

IGBT

Diode

Module 1

Module 2

Module 6

Rha

Cha +-

Ta

Heatsink

Interface

0 0.2 0.4 0.6 0.8 1

300

320

340

360

380

400

time (s)

junc

tion

tem

pera

ture

(K)

IGBTdiode

Fig.13. Thermal RC network for the converter Fig.14. Transient junction temperature(Ta=313K)

According to several simulations based on the thermal model shown in Fig.13, to keep the junction temperature below the maximum value, the thermal resistance of the heatsink adopted in this project should not exceed 0.018K/W. Fig. 14 shows the transient junction temperature of the IGBTs and diodes at the worst-case of power losses (Fig.12) with the thermal resistance of heatsink Rha=0.018K/W.

Finally, according to above limitation on the thermal resistance of the heatsink, together with some dimensions of the converter, the heatsink, LA20, produced by the company named Fisher is selected. Fig.15 shows its photograph and thermal characteristic in terms of thermal resistance. It can be found that the thermal resistance of the selected heatsink (400mm in length) is 0.015K/W, it’s smaller than the limitation.

a) Photograph b) Thermal characteristic

Fig.15. Photograph and thermal characteristic of the selected heatsink

Conclusion In this paper, the design of the power converter used in a 30kW SR S/G system is presented in detail.

Based on analysis and comparisons, the asymmetric half-bridge circuit is adopted as the topology of the converter and the IGBT is selected as the switches used in the converter. The maximum switching frequency of the switches in the converter is obtained by analytical calculation and PC-SRD

400mm


simulation, the results of the two methods match pretty well. The voltage and current ratings for both IGBT and diode are obtained. Some of the ratings are calculated by analytical methods, and all the analytical methods are well verified by the simulation results with PC-SRD and Matlab. Based on these, the power semiconductors suitable for the application are selected. Finally, the thermal analysis of the converter is presented, the thermal status of the selected power semiconductor modules is checked and the heatsink feasible for the considered power converter is selected.

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Power Electronic Converter for 30kw,270v Dc

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