FADING CHANNEL SIMULATOR -...
Transcript of FADING CHANNEL SIMULATOR -...
Department of Electrical Engineering
IIT Delhi
FADING CHANNEL SIMULATOR
(B.Tech Project)
Submitted by
Gaurav Sharma (98185) Mehakdeep Singh (98197)
Under the guidance of
Prof. H.M. Gupta
Certificate
We, Mehakdeep Singh and Gaurav Sharma, are submitting this report detailing work
done during our B.Tech Project in the year 2001-2002. The work done and the results
contained are our own and are genuine and all material taken from other sources (books,
manuals, internet, other thesis, etc.) has been fully acknowledged.
Mehakdeep Singh Gaurav Sharma (98197) (98185)
Date: April 26, 2002
Mr. Gaurav Sharma and Mr. Mehekdeep Singh have worked under my direct supervision
during this semester. I have read this report. It meets my expectations and it accurately
reflects the bonafide work done by the students.
Prof. H.M. Gupta (Supervisor)
Date: April 26, 2002
2
Acknowledgement
We hereby express our gratitude to our supervisor, Prof. H.M. Gupta of the Dept. of
Electrical Engineering, for his guidance in the execution of the project. It was indeed a
great honor to work under such unerring teacher, perfectionist and a person of an ever-
helping nature. We are especially grateful for all the help he provided and the resources
that he made available to us without which the project would not have reached its current
stage.
We would also like to thank our parents whose good wishes and silent blessings always
remained with us throughout the course of the project.
Mehakdeep Singh Gaurav Sharma
3
Table of Contents
Certificate 2
Acknowledgement 3
List of Figures 6
1. Introduction
1.1 Motivation 8
1.2 Problem Definition 9
1.3 Relevant Background 9
2. Theoretical Background
2.1 Multipath Propagation 11
2.2 Multipath Fading 12
2.3 Types of Fading 13
2.3.1 Rayleigh Fading 13
2.3.1.1 Effect of Motion 14
2.3.1.2 Phasor Representation 14
2.3.1.3 Probability Distribution Function 15
2.3.1.4 Power Spectrum 16
2.3.2 Rician Fading 16
2.3.3 Nakagami Fading 18
3. Design of Rayleigh Fading Simulator
3.1 Noise Generation Circuit 20
3.2 Shaping Filter Design 20
3.2.1 Peaking Amplifier Design 21
3.2.2 Low Pass Filter Design 22
3.3 Balanced Modulator Design 23
4
3.3.1 Using LM1496 Chip 23
3.3.2 Using AD630 Chip 24
3.4 RC Phase Shifter 25
3.5 LC Band Pass Filter 26
3.6 Adder using Op-amp RCA3142 27
3.7 PCB Design 28
3.8 Results 31
4. Verification Of The Output
4.1 Demodulation of Output 33
4.1.1 Squarer 33
4.1.2 Low Pass Filter 34
4.2 Sampling using 80196 Processor 34
4.3 Histogram Plotting 34
4.4 Results 35
5. Demonstration of Flooring
5.1 Flooring 36
5.2 PSK Modulation and Demodulation 37
5.3 Determination of BER 43
5.4 Results 43
6. Problems Faced 44
7. Future work 45
Appendix 1 46
Appendix 2 48
References 49
5
LIST OF FIGURES
Fig 1: Typical Multipath Propagation Environment
Fig 2: Multipath Fading, Shadowing and Path Loss phenomena
Fig 3: Phasor diagram of a set of scattered waves (in blue), resulting a Rayleigh-fading
envelope (in black)
Fig 4: Power density spectrum of a sine wave suffering from a Doppler spread
Fig 5: Block Diagram of Rayleigh Fading Simulator
Fig 6: Experimental Setup for noise generation
Fig 7: Experimental Setup for Shaping Filter
Fig 8: Peaking Amplifier Circuit
Fig 9: Low Pass Filter Circuit
Fig 10(a): LM1496 Chip and its Internal Circuit
Fig 10(b): LM1496 Chip and its Internal Circuit
Fig 11: AD630 Chip and its Internal Circuit
Fig 12: RC Phase Shifter
Fig 13: LC Band pass Filter
Fig 14: Op-amp RCA3142 Internal Structure
Fig 15: Complete Circuit Diagram of Rayleigh Simulator
Fig 16: PCB Layout
Fig.17 (a): Shaping filter frequency response
Fig 17(b): Band pass filter frequency response
Fig 18: AD630JN Chip Circuit
6
Fig 19: Low Pass Filter Circuit
Fig 20: Histogram of Output Samples
Fig 21: Comparison of Output with Rayleigh Distribution
Fig 22(a): (C/N) vs. Pe for different severity of Fades to demonstrate Fading
Fig 22(b): Set up to demonstrate Flooring
Fig 23: Set up for PSK Modulation
Fig 24: Set up for PSK Demodulation
7
1. INTRODUCTION
1.1 Motivation
Mobile radio channel simulators are essential for repeatable system tests in the
development, design, or test laboratory. Field tests in a mobile environment are
considerably more expensive and may require permission from regulatory authorities.
Because of the random, uncontrollable nature of the mobile propagation path, it is
difficult to generate repeatable field test results. The Rayleigh Fading Simulator can be
used to test the performance of radios in a mobile environment in the lab, without the
need to perform measurements whilst actually mobile. The mobile fading simulation can
also if required be replicated, and the effects can be varied according to the ‘velocity’ of
the mobile receiver. This allows the comparison of the performance of different receivers
under standardized conditions that would not normally be possible in actual mobile
testing situations.
Network Fading Simulator can also be used for the performance analysis of different
modulation-demodulation schemes. Random binary Sequences are generated and
modulated using the desired modulation scheme (generally QPSK or PSK). This binary
sequence is then detected in the presence of additive white Gaussian noise and
multiplicative noise. This multiplicative noise is often Rayleigh Distributed and is
generated by the network-fading simulator. Thus a plot between the C/N ratio and the Bit
Error Ratio (BER) can be obtained. This plot can be used to demonstrate a well-known
effect called flooring.
8
1.2 Problem Definition
The objectives of our project can be listed as follows:
1. Hardware design of a Rayleigh Fading Channel Simulator
2. Using the Simulator for performance analysis of PSK modulation scheme
3. To demonstrate the effect of Flooring using the PSK modulation and the
Simulator
1.3 Relevant Background
The following courses offered by the Department of Electrical Engineering have helped
us to understand the concepts involved in our project:
1. EE206N: This basic course on Communication Engg. familiarized us with the
various modulation schemes, signal and system representation schemes and the
concept of coherent and non-coherent detection. Moreover, the course provided
an overview on sampling, quantization and A/D conversion. 2. EE330N: This course furthered the concepts taught in EE206N and helped us to
gain insight into the digital modulation schemes like PSK, QPSK, FSK, ASK etc.
An overview of the performance analysis of the various digital schemes and their
detection methods in presence of noise was also given. 3. EE432N: This course on Satellite communications provided a general overview
and technical characteristics of various communication systems, multiple access
techniques, synchronization techniques and reception systems.
9
2. THEORETICAL BACKGROUND
2.1 Multipath Propagation
A typical model of a land mobile radio, including PCS and digital cellular transmission
link, consists of an elevated base-station antenna (or multiple antennas) and a relatively
short distance line-of-sight (LOS) propagation path, followed by many NLOS reflected
propagation paths and a mobile antenna or antennas mounted on the vehicle or more
generally on the transmitter/receiver (T/R) or transceiver of the mobile or portable unit.
In most applications, no complete, direct LOS propagation exists between the base-
station antenna, also known as the access point, and the mobile antenna because of the
natural and constructed obstacles. In such environment the radio transmission path, or
radio link, may be modeled as a randomly varying propagation path. In many instances
there may exist more than one propagation path, and this situation is referred to as
multipath propagation. The propagation path changes with the movement of the mobile
unit, the base unit, and/or the movement of the surroundings and environment.
Even the smallest, slowest movement causes time variable multipath, thus random time
variable signal reception. For example, assume that the cellular user is sitting in an
automobile in a parking lot, near a busy freeway. Although the user is relatively
stationary, part of the environment is moving at 100km/hr. The automobiles on the
freeway become “reflectors” of the radio signals. If during transmission or reception the
user is also moving, (for example, driving at 100km/hr), the randomly reflected signals
vary at a faster rate. The rate of variations of the signal is frequently described as Doppler
Spread.
10
Fig 1: Typical Multipath Propagation Environment
Three partially separable effects known as multipath fading, shadowing, and path loss
characterize radio propagation in such environments.
• Multipath propagation leads to rapid fluctuations of the phase and amplitude of the
signal if the vehicle moves over a distance in the order of a wavelength or more.
Multipath fading thus has a 'small-scale' effect.
• Shadowing is a 'medium-scale' effect: field strength variations occur if the antenna is
displaced over distances larger than a few tens or hundreds of meters.
• The 'large-scale' effects cause the received power to vary gradually due to signal
attenuation determined by the geometry of the path profile in its entirety. This is in
contrast to the local propagation mechanisms, which are determined by terrain
features in the immediate vicinity of the antennas.
• Path loss models describe the signal attenuation between transmit and receive
antenna as a function of the propagation distance and other parameters. Some models
can include many details of the terrain profile to estimate the signal attenuation.
11
The large-scale effects determine a power level averaged over an area of tens or hundreds
of meters and therefore called the 'area-mean' power. Shadowing introduces additional
fluctuations, so the received local-mean power varies around the area-mean. The term
'local-mean' is used to denote the signal level averaged over a few tens of wavelengths,
typically 40 wavelengths. This ensures that the rapid fluctuations of the instantaneous
received power due to multipath effects are largely removed.
2.2 Multipath Fading
Multipath fading is characterized by envelope fading (non-frequency-selective amplitude
distribution), Doppler spread (time selective or time variable random phase noise), and
time-delay spread (variable propagation distance of reflected signals causes time
variations in the reflected signals). These signals cause frequency-selective fades. These
phenomena are summarized in Fig.2.
Doppler Spread is defined as the spectral width of a received carrier when a single
sinusoidal carrier is transmitted through the multipath channel. If a carrier wave (an
unmodulated sinusoidal tone) having a radio frequency fc is transmitted, then because of
Doppler Spread fd we receive a smeared signal spectrum with spectral components
between fc – fd and fc + fd. This effect may be interpreted as a temporal decorrelation
effect of the random multipath-faded channel and is known as time-selective fading. The
effect of Time-delay Spread can be interpreted as a frequency-selective fading effect.
This effect may cause severe waveform distortions in the demodulated signal and may
impose a limit on the bit-error-ratio (BER) performance of high-speed radio systems.
12
Fig 2: Multipath Fading, Shadowing and Path Loss phenomena
2.3 Types of Fading
Many models for the probability distribution function of the signal amplitude exposed to
mobile fading have been given. Out of these models Rayleigh fading, Rician Fading and
Nakagami fading models are most widely used. We will now discuss these three models
in brief and in the following chapters Rayleigh fading will be discussed in detail.
2.3.1 Rayleigh Fading
Rayleigh fading is caused by multipath reception. The mobile antenna receives a large
number, say N, reflected and scattered waves. Because of wave cancellation effects, the
instantaneous received power seen by a moving antenna becomes a random variable,
13
dependent on the location of the antenna. Let us discuss the basic mechanisms of mobile
reception. In case of an unmodulated carrier, the transmitted signal has the form
.
.
2.3.1.1 Effect of Motion
Let the n-th reflected wave with amplitude c_n and phase arrive from an angle
relative to the direction of the motion of the antenna.
The Doppler shift of this wave is
,
where v is the speed of the antenna.
2.3.1.2 Phasor representation
Fig 3: Phasor diagram of a set of scattered waves (in blue),
resulting a Rayleigh-fading envelope (in black)
14
The received unmodulated signal r(t) can be expressed as
An in phase-quadrature representation of the form
can be found with in-phase component
and quadrature phase component
.
Provide that N is sufficiently large and that all the cn are equal, then by the central limit
theorem, I(t) and Q(t) can be shown to be zero mean stationary baseband Gaussian
processes.
2.3.1.3 Probability Distribution Function
If we express r (t) as:
r(t) = R(t) cos2л fct + θ(t)
then, the probability density functions of R and θ are given respectively by
p(R) = R/σ2 exp(-R2/2σ2) R≥0
p(θ) = 1/2л -л≤θ≤л
The preceding equations indicate that the received faded carrier e(t) has a Rayleigh
distributed envelope r(t) and a uniformly distributed phase θ(t).
15
2.3.1.4 Doppler Power Spectrum
Assuming that many waves arrive each with its own random angle of arrival (thus with
its own Doppler shift), which is uniformly distributed within [0, 2 pi], independently of
other waves. This allows us to compute a probability density function of the frequency of
incoming waves. Moreover, we can obtain the Doppler spectrum of the received signal.
This leads to the U-shaped power spectrum for isotropic scattering,
Fig 4: Power density spectrum of a sine wave suffering from a Doppler spread
2.3.2 Rician Fading
The model behind Rician fading is similar to that for Rayleigh fading, except that in
Rician fading a strong dominant component is present. This dominant component can for
instance be the line-of-sight wave. Refined Rician models also consider
• that the dominant wave can be a phasor sum of two or more dominant signals, e.g.
the line-of-sight, plus a ground reflection. This combined signal is then mostly
treated as a deterministic (fully predictable) process, and
• that the dominant wave can also be subject to shadow attenuation. This is a
popular assumption in the modeling of satellite channels.
16
Besides the dominant component, the mobile antenna receives a large number of
reflected and scattered waves.
The derivation is similar to the derivation for Rayleigh fading. In order to obtain the
probability density of the signal amplitude we observe the random processes I(t) and
Q(t) at one particular instant t_0. If the number of scattered waves is sufficiently large,
and is i.i.d., the central limit theorem says that I(t_0) and Q(t_0) are Gaussian, but, due to
the deterministic dominant term, no longer zero mean. Transformation of variables shows
that the amplitude and the phase have the joint pdf
Here, is the local-mean scattered power and is the power of the dominant
component. The pdf of the amplitude is found from the integral
,
where is the modified Bessel function of the first kind and zero order, defined as
17
2.3.3 Nakagami Fading
Nakagami fading occurs for instance for multipath scattering with relatively large delay-
time spreads, with different clusters of reflected waves. Within any one cluster, the
phases of individual reflected waves are random, but the delay times are approximately
equal for all waves. As a result the envelope of each cumulated cluster signal is Rayleigh
distributed. The average time delay is assumed to differ significantly between clusters. If
the delay times also significantly exceed the bit time of a digital link, the different
clusters produce serious intersymbol interference, so the multipath self-interference then
approximates the case of co-channel interference by multiple incoherent Rayleigh-fading
signals. Following are some important facts related to Nakagami Fading.
• If the envelope is Nakagami distributed, the corresponding instantaneous power is
gamma distributed.
• The parameter m is called the 'shape factor' of the Nakagami or the gamma
distribution.
• In the special case m = 1, Rayleigh fading is recovered, with an exponentially
distributed instantaneous power
• For m > 1, the fluctuations of the signal strength reduce compared to Rayleigh
fading.
18
3. DESIGN OF RAYLEIGH FADING SIMULATOR
A detailed implementation concept of the Rayleigh fade simulator is illustrated in Fig 5.
The theoretical foundation and justification for the implementation of such simulators has
been explained already in previous sections. From our derivations and results, we note
that the in-phase i(t) and quadrature phase q(t) baseband control signals must be band-
limited Gaussian sources with a power spectral density W(f). This density is proportional
to:
W(f) = C. 1/( fD2- f2)1/2 |f|≤ fD
= 0 |f|≥ fD
where fD is the maximum Doppler Frequency and C is proportionality constant.
The Rayleigh envelope statistics are obtained by adding two noise sources in quadrature.
The theoretical spectrum of the received signal is approximated by shaping the spectrum
of the noise sources with filters.
19
Fig 5: Block Diagram of Rayleigh Fading Simulator
3.1 Noise Generation Circuit
Experimental setup for the generation of noise sources is given in Fig6. The zener diode
used was Z122. It had breakdown voltage of 8.7V. A voltage of 12V was applied to the
zener diode through a resistance of 20KΩ. The noise so obtained was amplified using an
operational amplifier circuit. This noise was then fed to the shaping filter to appropriately
shape its spectrum.
Fig 6: Experimental Setup for noise generation
20
3.2 Shaping Filter Design
Fig 7: Experimental Setup for Shaping Filter
Shaping filter circuit shown in Fig 7 can be divided into two parts:
• Low pass filter
• Peaking amplifier
3.2.1 Peaking Amplifier Design
Fig 8: Peaking Amplifier Circuit
The peaking amplifier can be represented as a Twin-T network as shown in Fig 8 and its
transfer function is given by:
21
=iV
V0
1)()(1)()(
331212
213232313131213132213
321321
2132
213213
321321
++++++++++++++
sCRCRCRsCCRRCCRRCCRRCCRRCCRRsCCCRRRsCCRsCCRRRsCCCRRR
In our case,
R1=R2=R3/2=R and C1=C2=2C3=C
=iV
V0
R3C3s3+R2C2s2+RCs+1 R3C3s3+5R2C2s2+7/2RCs+1
The desired peaking frequency was 100KHz.
The peak frequency is given by the formula, fp = 1/(2лRC)
We chose the value of C = 1nF, for this value of C the value of R obtained from the
above formula is 1.59KΩ.
From these values of R and C we obtained the values of R1, R2, R3, C1, C2 and C3.
3.2.2 Low Pass Filter Design
Experimental setup for the low pass filter is given in Fig 9. The low pass filter chosen
was Butter worth low pass filter of second order. The reason why we chose Butter worth
filters is that they offer the flattest pass band and provide a fast initial falloff and
reasonable overshoot. The design procedure is given below:
1. The corner frequency (fc) was chosen to be 100KHz.
2. The value of C was chosen to be 1nF.
3. R2 was calculated as, R2 = 1/(2πfcC) = 1.6 KΩ.
22
4. R1 was calculated as R1 = XR2, where X is dependent on the gain that you desire. The
gain was chosen to be 2 and for this value of gain X was obtained to be 0.5. This
gives you the value of R1 = 800 Ω.
Fig 9: Low Pass Filter Circuit
3.3 Balanced Modulator
The Balanced Modulation has been tried using two different circuits:
• Using LM1496 chip
• Using AD630 chip
3.3.1 Balanced Modulator Design using LM1496
The chip circuit is shown in Fig 10(a) and Fig 10(b). The LM1496 is a double balanced
modulator, which produce an output voltage proportional to the product of an input
(signal) voltage and a switching (carrier) signal. The LM1496 has adjustable gain and
signal handling, fully balanced inputs and outputs, low offset and drift and a wide
frequency response up to 100 MHz. The input to Pins 8 and 10 is the Carrier Input and to
Pins 1 and 4 is the Modulator Input. Pins 2 and 3 are for gain adjustment and Pin 14 is
connected to –V. The output is obtained on Pins 6 and 12. The internal structure of the
pin is also shown in Fig 10.
23
Fig 10(a): LM1496 Chip and its Internal Circuit
Fig 10(b): LM1496 Chip and its Internal Circuit
3.3.2 Balanced Modulator Design using AD630JN
The chip circuit is shown in Fig 11. The AD630JN is a high precision balanced
modulator that combines a flexible commutating architecture with the accuracy and
temperature stability afforded by laser wafer trimmed thin-film resistors. AD630JN is a
24
20-pin chip. The Modulation input is given to Pin 16 and the Carrier Input to Pin 9. Pins
2 and 3 and Pins 5 and 6 are for gain adjustment. The output is obtained on Pin 13. The
internal structure of the pin is also shown in Fig 11.
The configuration of AD630JN makes it ideal for signal processing applications such as
balanced modulation and demodulation. The 100dB dynamic range of AD630JN exceeds
that of any IC Balanced Modulator and is comparable to that of costly signal processing
instruments. Moreover, the op amp format of AD630JN ensures easy implementation of
high gain applications with no additional parts.
Fig 11: AD630 Chip and its Internal Circuit
25
3.4 RC Phase Shifter
To obtain the required Rayleigh Fading spectrum, the two noise sources have to be added
in quadrature. This shift of 90 degrees between two RF sources is obtained using the
circuit shown in Fig 12. Here the idea is to make the impedance of capacitor and
resistance same at the desired frequency of operation.
f = 1/ (2 ΠRC) = 1 MHz
C = 1nF => R = 160 Ω
Fig 12: RC Phase Shifter
3.5 LC Band Pass Filter
The basic circuit of LC bandpass filter is shown in Fig 13. It consists of two transistor
stages. The basic idea is that LC circuit shows very high impedance at resonance and
small resistance otherwise. Since the ac gain of the common collector configuration is
proportional to the resistance put at the collector, so by putting a LC circuit there we can
26
get a peaking at the desired frequency. The capacitors used in the two stages were both
variable capacitors. By properly adjusting the value of the capacitors you can make the
two circuits to peak at slightly different frequencies. This makes the transfer function
almost constant in the in between region and falls sharply after that. Hence a bandpass
filter close to ideal can be obtained. The reason why we did not use an active RC
bandpass filter is that in this case the resonant frequency is quite a complicated function
of all the resistances and capacitances. And also it is very sensitive to changes in any of
the component values. Slight changes in the component value can alter the passband and
center frequency by a large value.
Fig 13: LC Band pass Filter
3.6 Adder using Op-amp RCA3140
The two noise sources produced in quadrature cannot be added using the normal
operational amplifier HA17741 due to its low bandwidth. HA17741 adds a lot of noise
and gives distorted output at high frequencies. To avoid this problem, RCA3142 chip is
used. Its pin structure is shown in Fig 14. RCA3142 is a wideband monolithic amplifier
with improved speed and stability and low power consumption. It has a bandwidth of
10MHz and thus is ideally suited for our application. RCA3142 has the same number of
27
pins as HA17741 (8pins). Pins 7 and 4 are connected to +12V and –12V respectively.
Pins 2 and 3 are the input pins and Pin7 is the output pin.
Fig 14: Op-amp RCA3142 Internal Structure
3.7 PCB Design
Initially all the components of the Rayleigh fading channel simulator (as described
above) were made on the bred board. The complete circuit diagram is shown in Fig 15.
After verifying that all the components were working as desired we made a PCB of the
complete circuit. The PCB layout is given in Fig 16. Following were the additional
features of the PCB:
• The noise produced by the two zener diode is generally not equal in amplitude.
This noise is then passed through the amplifier, low pass filter and peaking
amplifier. The response of these can also be slightly different in the two cases.
But before combining the two noise sources we must make sure that their
amplitudes are equal. To achieve this we added an additional variable gain stage
in each of the two circuits. By varying the gain of this stage we can make the
amplitudes of the two noise sources equal.
28
• Finally the summer, which sums the outputs of two AD630JN chips, was also
provided with variable gain. By adjusting the gain of the adder, we could change
the overall output of the Rayleigh Fading Simulator.
• Some additional capacitors were added to reduce the effect of interference. And
the results were quite good, with all these capacitors and some other design
features interference was almost removed at the output.
29
30
Fig 16: PCB Layout
31
3.8 Results
The following were the results obtained:
• The power spectrum of the noise generated by the zener diode was found to be
flat over the entire frequency range of 10Hz to 1MHz. After passing the noise
through the amplifier circuit it was found that the higher frequency components
were slightly attenuated. This was expected because the operational amplifiers
used had a finite bandwidth.
• The shaping filters frequency response obtained is shown in Fig 17(a). Here the
value on the y-axis is the normalized gain obtained as:
Gn(f) = G(f) / G(0)
where Gn(f) is the normalized gain at frequency f Hz, G(f) is the actual gain at
frequency f Hz and G(0) is the gain at 0 Hz.
Normalized Gain vs Frequency
-25
-20
-15
-10
-5
0
5
10
15
0 50 100 150 200 250
Frequency (in KHz)
Nor
mal
ized
Gai
n (in
dB
)
Fig.17 (a): Shaping filter frequency response
The response above is the combined response of the low pass filter as well as the
peaking amplifier.
32
• The desired phase shift of 90 degrees was obtained between the two outputs
taken from the phase shifter.
• The response of the Band pass filter was found to be flat for frequency range
900KHz to 1.1MHz and response was quite small for frequencies below
500KHz and above 1.5MHz that is what was required. This is shown in Fig
17(b).
00.5
11.5
22.5
33.5
4
0 500 1000 1500 2000 2500
FREQUENCY (IN KHz)
GA
IN
Fig 17(b): Band pass filter frequency response
The above results were also obtained on the PCB and were found to be the same.
However, by the visual inspection of the output one could not verify whether the
amplitude was Rayleigh distributed or not. So for that we demodulated the output and
plotted the histogram of the output samples. This is illustrated in the next section.
33
4. VERIFICATION OF THE OUTPUT
4.1 Demodulation of Output
Since the output of the Rayleigh Fading channel simulator does not have any carrier and
has random phase component this makes the use of envelope detector and coherent
demodulation schemes impossible. So the only way it can be demodulated is using a
squarer followed by a low pass filter. The output of the squarer will be a term
proportional to the square of the output amplitude and some high frequency component
centered on twice the carrier frequency. The low pass filter removes the high frequency
component and gives the output proportional to the square of the received signal.
4.1.1 Squarer
The squarer is implemented using AD630JN chip. The output of the Fading Channel
Simulator is applied to the carrier as well as the modulating input. The output obtained is
fed to the low pass filter.
Fig 18: AD630JN Chip Circuit
34
4.1.2 Low Pass Filter
The low pass filter used was the same as the one used in the shaping filter. The cutoff
frequency was 100 KHz.
Fig 19: Low Pass Filter Circuit
4.2 Sampling using 80196 Processor
The demodulated output was fed to the input of 80196 microprocessor. This analog input
was then sampled and the corresponding 8-bit samples were stored in memory. The
program for Sampling followed by A/D conversion is given in Appendix 1.
4.3 Histogram Plotting
The above-mentioned program stores the histogram of the output samples in memory
locations. These values were then manually read and written into a file called input.txt.
The histogram of the output samples was then plotted and compared with Rayleigh
distributions with varying σ. The Matlab program for this is given in Appendix 2.
35
4.4 Results
The histogram of the output samples is shown in the Fig 20. The histogram was than
compared with Rayleigh distribution for different values of σ as shown in Fig 21. It was
found that the output was closest to Rayleigh distribution with σ equal to 0.8.
-2000
0
2000
4000
6000
8000
10000
12000
14000
16000
0 0.5 1 1.5 2 2.
Demodulated Output (in V)
No.
of s
ampl
es
5
Fig 20: Histogram of Output Samples
0 0.5 1 1.5 2 2.50
2000
4000
6000
8000
10000
12000
14000
16000
18000
VOLTAGE
NO
. OF
SA
MP
LES
Rayleigh (sigma = 0.7) Rayleigh (sigma = 0.8) Rayleigh (sigma = 0.9) Actual Output
Fig 21: Comparison of Output with Rayleigh Distribution
36
5. DEMONSTRATION OF FLOORING
5.1 Flooring
After verification of the output, our next objective is to demonstrate flooring. For that we
will be we will be generating a pseudo random binary sequence and modulating and
demodulating it in the presence of AWGN and multiplicative noise generated through our
fading channel simulator. For a given severity of the multiplicative noise as you keep on
decreasing the AWGN what you observe is that bit error rate does not fall after some time
as shown in Fig. 22(a). This phenomenon is called flooring. Different curves are obtained
for different severities of the fade. The Block Diagram of the circuit to be used to
demonstrate the Flooring effect is shown in Fig 22(b).
Fig 22(a): (C/N) vs. Pe for different severity of Fades to demonstrate Fading
Fig 22(b): Set up to demonstrate Flooring
37
The modulation scheme used to demonstrate Flooring is PSK and its explanation is given
below.
5.2 PSK Modulation and Demodulation
Phase-shift keying (PSK) means altering the phase of a (usually sinusoidal) carrier signal
with respect to a reference phase, in accordance with the value of the base-band signal.
PSK, to an extent often greater than with frequency-shift keying, has the advantage that
amplitude variations can be suppressed by limiting. It is also beneficial in reducing noise.
However it is more susceptible to sudden changes in transmission delay of a channel,
such as can happen with radio links, and the modulation and demodulation processes tend
to be more complex. The amount of phase-shift can for a binary (two-valued) baseband
signal can be ±90 or it can be less.
In this experiment modulation will be achieved by adding, to a constant carrier by phasor
C, a quadrature component represented by Q that may be reversed in phase to give
resultant signal of phase ±ø.
The method of demodulation to be used must depend on reconstructing an equivalent of
C. as a reference phase, with which the phase of the received signal can be compared.
This is easily done if the data format is such as to give equal periods of 0 and 1 signal in a
short interval, such as bi-phase code. It is then only necessary to use a phase lock loop
whose action is slow enough to hold the oscillator at the mean of the two signal phases.
38
For the PSK modulation and demodulation, the kit available in the lab was used and it
had the following components:
1 DCS297A Data Source
1 DCS297B Data Format
2 DCS297C Double Balanced Modulators
1 DCS297D Carrier Phase-shifter
1 DCS297E Voltage-Controlled Oscillator
1 DCS297F Data Clock Regeneration
1 DCS297G Data Recovery
1 DCS297H Data Receiver
1 DCS297K Audio Module
1 DCS297L Tuned Circuit
1 DCS297M Power Supply
1 DCS297N set of connecting leads
Dual-beam oscilloscope
Modulation
Connect the equipment as shown in Fig 23. Note that the bi-phase data waveform from
the Data Format module DCS297B always has the same dc component, averaged over
any one bit-time. This can be verified by measuring its dc component with a voltmeter.
Because of this, the capacitor-feed to the modulator can reject the dc and produce equal
positive and negative inputs to the modulator proper. Check this with the oscilloscope at
input b. Use the oscilloscope to verify that the Carrier Phase-shifter DCS297 D produces
39
carrier waveforms in mutual quadrature at links 10 and 12. The associated gain control
may be turned fully clockwise and the phase control should be adjusted initially to give
approximately equal output voltages. These two voltages are fed respectively to two
modulators. The lower modulator has a constant bias as its second input, so that it’s
output corresponds to the C phasor. The bias value controls the magnitude (and sign) of
this output. It should be set to a positive value by turning the control clockwise. Link 10
supplies the quadrature carrier, which is reversed in phase as the modulating signal (link
9) changes sign at the 'b' terminal. The output currents of the two modulators are
combined in the common load, producing the phase-modulated signal. The phase-
modulated signal can be examined as follows:
• Increase the time base speed to say 2 microseconds per division.
• Synchronize (using the external sync/trigger connection for later convenience) to
the 1.28MHz carrier, link 6, to which also the Y1 channel should be connected.
• Finally display the output, link 14, on Y2.
This will show the two output phases superimposed. The amount of modulation can be
varied by adjusting the 'phase' control. It should be set less then ±90.
Demodulation
Without disturbing the equipment already set up connect up the remainder as shown in
Fig 24. The links 14 and 15 are the communication links and are therefore the same as the
links 14 and 15 shown in Fig 23. With a signal on link 14, transfer the Y1 lead (but not
the sync/trigger lead) to the VCO module, link 16. This should show the carrier recovered
from the signal by the phase-lock loop. It will vary in phase somewhat, but not much,
40
because, although the loop tries to lock it to the changing phase of the incoming signal, it
is slow acting. Note that its phase is in quadrature with the mean phase of the received
signal. The latter's quadrature components (Q in fig 3.8) are therefore at 0° or 180° to it.
A sinusoid multiplied by another of the same phase produces a dc component in the
output:
(2sin2ω= 1 - cos2 ω t)
When one of them is shifted 180° this component changes sign. The modulator output on
links 20, 21 therefore contains a component that represents the original bi-phase data.
The shunt capacitor passes most of the component of frequency 2ω t, keeping the ripple
voltage small. Check this with the oscilloscope, Y1 showing the bi-phase data link 9, and
Y2 the recovered data link 20. The settings should be restored to the usual:
• Y1 and Y2; dc-coupled
• 5V/division time base
• 10µs/division, externally triggered by +ve going edge
41
Fig 23: Set up for PSK Modulation
42
Fig 24: Set up for PSK Demodulation
43
5.3 Determination of BER
In our case the delay between the input stream, i.e. the sequence of bits generated by the
pseudo random generator and the output stream (demodulated bit stream) was quite less.
So we can simply use a XOR gate followed by a counter to count the number of errors.
Obviously the setup should be run for small enough time so that the counter does not
overflow.
5.4 Results
• Initially we were making our own PRBS generator using LFSR’s (Linear
Feedback Shift Registers). But the PSK modulation and demodulation kit
available in the lad had its own PRBS generator. So we used the PRBS generator
available with the kit.
• Since some of the modules in the PSK demodulator kit were not working, so we
could not set up the entire circuit as intended. Right now we are working on how
we can get the signal, which will be obtained, at the receiver side in case of fading
channel. For this we are trying to multiply the demodulated output of our
Rayleigh Fading channel simulator with the modulated signal.
44
6. PROBLEMS FACED
• The desired frequency of operation was 10.7 MHz, but the operational amplifiers
available had maximum possible bandwidth of 5 MHz.
• At frequencies in the range of few MHz or higher there was lot of interference
due to radiation. We used all the probes available in the lab, but still interference
could not be avoided. This was the reason why we could not properly test the
output of the balanced modulator chip and the phase shifter.
• The spectrum analyzer available in the lab is capable of showing the frequency
spectrum only up to 50 KHz. After that we had to find the amplitude response
using CRO to obtain the power spectrum density.
• There was only one RF source available in the lab, that was capable of working
up to 20 MHz. But the output of the source was distorted in the frequency range
of about 5 MHz to 14 MHz.
Because of all the above-mentioned problems we shifted our frequency of operation
down to 1 MHz. In the case of bred board there was a lot of interference due to radiation
etc. even at this frequency. But in the case of PCB this interference was almost zero.
Another problem was that the demodulator part of the PSK modulator and demodulator
kit available in the optical communications lab was not working.
45
7. FUTURE WORK
1. The power spectrum of the output can be observed using the new power spectrum
analyzers, which are going to be available in the communications lab soon. These
new spectrum analyzers can work from a frequency of 400KHz to 1GHz as
compared to the previous spectrum analyzers that work only up to 50KHz.
2. BER detection can be done pretty easily, with the BER detection equipment,
which will be available in one or two months in the optical communications lab.
3. Rician fading can also be obtained with slight modification in the circuit.
46
APPENDIX 1
Program for A/D Conversion and Sampling using 80196 microprocessor
ORG 4000H AX EQU 1CH AL EQU 1CH AH EQU 1DH BX EQU 1EH BL EQU 1EH ES EQU 24H AD_COM EQU 02H ADRESL EQU 02H ADRESH EQU 03H USERADC EQU 0D2H INTPEND EQU 09H INTMASK EQU 08H PBCD_BCD EQU 22F0H LED_DISP EQU 21B0H MAIN: LD USERADC, #4100H CLRB INTPEND LDB INTMASK, #02H EI LDB AD_COM, #18H LOOP: SCALL AD_DIS SJMP LOOP ORG 4100H AD_INT: LDB 38H, ADRESL LDB 39H, ADRESH LDB 3AH, 38H ANDB 3AH, #07H ANDB 38H, #0C0H SHRB 38H, #06H EI LDB AD_COM, #18H RET AD_DIS:
47
LDB 0EFH, #0AH LDB 0EEH, #0DH LDB 0EDH, #14H LDB AH, 39H LD BX, #6000H MUL AH, #02H ADD AX, BX MOV BX, AX LD AX, [BX] INC AX ST AX, [BX] RET END
Explanation of the Program: The program samples the analog input available at Channel
1 of Connector C6 at 10 KHz and converts it into a digital output of 8 bits. The number of
samples corresponding to each digitized value is stored in memory locations starting from
6000H as a 16-bit value. The total number of samples taken is 10,00,000.
48
APPENDIX 2
Matlab Program for comparison of Fading Channel Simulator Output with Standard
Rayleigh Distribution
clear; fid = fopen(‘input.txt’,’r’); for I = 1:256 s = fgetl(fid); a1(I,1) = str2num(s); end fclose(fid); a2 = ray (0.7); a3 = ray (0.8); a4 = ray (0.9); for i =1:256 b(i,1) = sqrt((i-0.5)/256); end plot(b,a2,’r’,b,a3,’g’,b,a4,’b’,b,a1,’k’); function out= ray(sigma) for I=1:256 out(I,1)= uint16(1000000*(raylcdf((I-1)/256,sigma)-raylcdf((i/256),sigma))); end
49
50
REFERENCES
1. ‘Radio Propagation and Cellular Engineering Concepts’ by K. Feher.
2. ‘A Multipath Fading Simulator for Mobile Radio’ by Gaston A. Arredondo and Advert
H. Walker- IEEE Transactions on Vehicular Technology, Vol. VT-22, No. 4, Nov.
1973.
3. ‘A Computer Generated Multipath Fading Simulation for Mobile Radio’ by John I.
Smith- IEEE Transactions on Vehicular Technology, Vol. VT-24, No. 3, Aug. 1975.
4. ‘Mobile Radio Communications’ by R. Steele and Hanzo – IEEE Press.
5. http://www.engr.sjsu.edu/filt175_s01/sp2001/proj_sp01c/mai_trang/Filt_pass_last.htm
6. http://dbserv.maxim-ic.com/appnotes.cfm?appnote_number=700
7. http://www.tele-ip.com/fadesimulator.html
8
9
. T. Eyceoz, A. Duel-Hallen and H. Hallen, Deterministic Channel Modeling and Long
Range Prediction of Fast Fading Mobile Radio Channels, IEEE Communications
Letters, Vol. 2, No. 9, pp. 254-256, September 1998.
. A Systematic Approach to the Design and Analysis of Optimum DPSK Receivers for
Generalized Diversity Communications over Rayleigh Fading Channels, IEEE
Transactions On Communications, Vol. 47, No. 9, September 1999 1365, Mahesh K.
Varanasi, Senior Member, IEEE.