Efficiency Analysis of Drive Train Topologies Applied to Electric/Hybrid Vehicles

IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 3, MARCH 2012 1021

Efficiency Analysis of Drive Train TopologiesApplied to Electric/Hybrid Vehicles

Jorge O. Estima, Member, IEEE, and Antonio J. Marques Cardoso, Senior Member, IEEE

Abstract—One of the most important research topics in drivetrain topologies applied to electric/hybrid vehicles is the efficiencyanalysis of the power train components, including the global driveefficiency. In this paper, two basic traction electric drive systems ofelectric/hybrid vehicles are presented and evaluated, with a specialfocus on the efficiency analysis. The first topology comprises atraditional pulsewidth-modulation (PWM) battery-powered in-verter, whereas in the second topology, the battery is connectedto a bidirectional dc–dc converter, which supplies the inverter.Furthermore, a variable-voltage control technique applied to thissecond topology is presented, which allows for the improvementof the drive overall performance. Some simulation results arepresented, considering both topologies and a permanent-magnetsynchronous motor (PMSM). An even more detailed analysis isperformed through the experimental validation. Particular at-tention is given to the evaluation of the main drive componentsefficiency, including the global drive efficiency, presented in theform of efficiency maps. Other parameters such as motor voltagedistortion and power factor are also considered. In addition, thecomparison of the two topologies takes into account the driveoperation under the motoring and regenerative-braking modes.

Index Terms—Automotive applications, bidirectional powerflow, dc–dc power converters, electric vehicles, energy efficiency,machine vector control, permanent-magnet machines, power con-version, voltage control.

NOMENCLATURE

vd, vq dq-axes voltage components.id, iq dq-axes current components.icd, icq dq-axes iron losses current components.imd, imq dq-axes magnetizing-current components.Rs Stator winding resistance.ω Stator currents frequency.Ld, Lq dq-axes inductance components.ψPM Flux linkage due to the rotor magnets.ωr Rotor mechanical speed.

Manuscript received June 28, 2011; revised November 29, 2011; acceptedJanuary 29, 2012. Date of publication February 6, 2012; date of currentversion March 21, 2012. This work was supported in part by the PortugueseGovernment through the Foundation for Science and Technology under GrantSFRH/BD/40286/2007 and Grant PTDC/EEA-ELC/105282/2008. The reviewof this paper was coordinated by Dr. A. Davoudi.

J. O. Estima is with the Department of Electrical and Computer Engineering,University of Coimbra, 3030-290 Coimbra, Portugal, and also with the Institutode Telecomunicações, 3030-290 Coimbra, Portugal (e-mail: [email protected]).

A. J. Marques Cardoso is with the Department of Electromechanical En-gineering, University of Beira Interior, 6201-001 Covilhã, Portugal, and alsowith the Instituto de Telecomunicações, 3030-290 Coimbra, Portugal (e-mail:[email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TVT.2012.2186993

J Moment of inertia.Te Electromagnetic torque.D Damping coefficient.TL Load torque.θ Rotor electrical position.p Pole pairs number.PCu Stator winding copper power losses.PFe Stator iron power losses.η Efficiency.Pin Input power.Pout Output power.Plo Power losses.XRMS Waveform root means square (RMS) value.X1 Waveform fundamental component RMS value.

I. INTRODUCTION

CONSIDERING the high oil consumption rate of the trans-portation sector and due to the arising concerns about

global warming and energy resource constraints, governmentagencies and organizations are imposing more stringent regula-tions for fuel consumption and emissions.

With this increasing demand for environmentally friendlierand higher fuel economy vehicles, the automotive companiesare focused on the development of new technologies such ashybrid electric, electric, fuel cell, and plug-in vehicles [1].Although electric and fuel cell vehicles represent a possiblesolution for this problem, currently, they have some limita-tions that make them a less attractive option than the othervehicles [2].

For these automotive traction systems, the electric motoris one of the most important key components. The majorrequirements for these applications include high torque andpower density, high starting torque, high efficiency over widetorque and speed ranges, a very wide speed range, includingconstant torque and constant power regions, high intermit-tent overload capability, and reasonable cost [3]–[7]. In thiscontext, permanent-magnet synchronous motors (PMSMs) arebecoming much more attractive. These machines have inherentfeatures such as high power density and high efficiency, whichmake them very suitable for these applications.

With regard to the traction electric drive system, thefollowing two basic configurations are used [8]–[10]: 1) a tra-ditional pulsewidth-modulation (PWM) battery-poweredinverter [see Fig. 1(a)] and 2) a configuration in whichthe battery is connected to a bidirectional dc–dc converter,typically a boost converter, which supplies the inverter [seeFig. 1(b)]. The use of a bidirectional dc–dc converter to

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1022 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 3, MARCH 2012

Fig. 1. Basic traction electric drive topologies for electric/hybrid vehicles.(a) Topology T1. (b) Topology T2.

connect the battery and the inverter has several advantagesover the configuration where the battery directly powers theinverter. Beyond the possibility of minimizing the stress of theinverter with an extra dc stage, it is possible to improvethe motor output by increasing and controlling the invertersupply voltage (system voltage) without increasing the batterycost and size due to the requirement of building up a highercell number, simultaneously keeping the same motor size [11].This topology also permits manufacturers to separately designthe system voltage and battery, allowing for flexible systemdesigns for vehicles with different output characteristics.System voltage design can be based on the motor output andwithstanding voltage of parts, and for battery voltage design,the number of cells can be altered to match the required batteryoutput and capacity [12], [13]. However, this configurationhas some disadvantages such as power losses in the dc–dcconverter and the increase of system complexity and cost.This configuration also raises the concern about the powercomponents of physical dimensions. Despite the batterysize reduction, the additional inductor and power switches,which may require the optimization and size reduction of thepower electronics components, must be taken into account.The reliability issues due to the introduction of an additionalcomponent must also be taken into account.

One important aspect in the selection of these topologiesis the system cost versus the cost of use relation. The secondtopology requires a smaller battery, and therefore, its cost canbe reduced. Nevertheless, the cost with regard to the extrapower components and more complex design issues can com-pensate for the cheaper battery. Considering the cost of use,beyond the drive system efficiency, it strongly depends on themaintenance aspects. As a consequence, important issues suchas battery lifetime, its corresponding cost and the reliabilitylevels that are associated with the topologies complexity mustbe taken into account for an accurate analysis.

Other configurations can be obtained by combining thesetopologies with different energy storage technologies. In [14]

and [15], a high-voltage battery pack directly supplies theinverter, supported by an ultracapacitor bank, connected tothe battery bus through bidirectional dc–dc converters. Par-allel converters were considered, which allow for reducingthe inductor current ripple and improving the global effi-ciency. The same power sources can also be connected to-gether to a dc–dc converter, which supplies the high-voltagebus [16].

The combination of batteries and fuel cells can also beused for the drive train of hybrid/electric vehicles. In [17],the integration of these power sources with several topologiesof isolated and nonisolated dc–dc converters was addressed.This subject, as well as other different configurations, was alsoinvestigated in [18].

The advantages of fuel cells can also be combined with thegreat dynamic response and power density of ultracapacitorsfor these electric traction systems [19]–[21]. The presentedresults allow for concluding that, with an appropriate powermanagement control technique, this hybrid source can provideany load demand. Furthermore, by using bidirectional dc–dcconverters, it is possible to achieve better control over the fuelcell voltage during transients. Considering the combination ofthese hybrid sources and using several dc–dc converters, anenergy management strategy based on flatness and fuzzy logiccontrol techniques was presented in [22]. As a main advantage,the power management is carried out with a single generalcontrol algorithm in different operating modes.

Due to the great importance of bidirectional dc–dc convertersin advanced transportation systems, a novel capacitor-switchedregenerative snubber for dc–dc boost converters was proposedin [23]. The resulting topology allows for reducing the switch-ing losses so that high-switching-frequency operation can beachieved. As a consequence, higher efficiency values can beobtained, and the overall converter mass can significantlybe reduced.

The literature review shows that, despite all the existing workdone by several authors with regard to the study of electric drivesystems for electric/hybrid vehicles, there is a lack of researchwith regard to the detailed efficiency analysis of different powertrain configurations for these specific applications. Accordinglyand considering the work reported in [24], this paper providesan in-depth analysis of two basic electric drive topologiesbased on a PMSM (see Fig. 1), with a special focus on theefficiency analysis. Furthermore, considering topology T2 [seeFig. 1(b)], an improved variable-voltage control technique forthe dc–dc converter is described and implemented, allowingfor dynamically adjusting the system voltage, thus contributingto achieving higher efficiency values and improving the driveoverall performance. Some simulation results are presentedusing a PMSM dynamic model that takes the iron losses intoaccount. An even more detailed analysis is performed throughthe experimental validation, where efficiency maps are obtainedfor each drive component, including the efficiency maps ofboth topologies’ global efficiency. In addition, motor voltagesdistortion and power factor results are evaluated. Finally, acomparison of the two topologies is provided for the driveoperation under the motoring and regenerative-braking modestrough a drive-cycle analysis.

ESTIMA AND MARQUES CARDOSO: EFFICIENCY OF DRIVE TRAIN APPLIED TO ELECTRIC/HYBRID VEHICLES 1023

Fig. 2. dq-axes equivalent circuits for the PMSM model, taking iron lossesinto account. (a) d-axis. (b) q-axis.

II. PERMANENT-MAGNET SYNCHRONOUS MOTOR

DYNAMIC MODEL WITH IRON LOSSES

Typical PMSM mathematical models found in the literaturedo not take iron losses into account. Therefore, to obtain a moreaccurate modeling, particularly for the iron losses, a dedicatedparameter that is aimed at accounting for the iron losses inthe stator core, particularly the eddy current losses, has beenconsidered. These losses are modeled by a resistor Rc that isinserted in parallel with the magnetizing branch so that thepower losses depend on the air-gap flux linkage [25]–[30].Thus, the dq-axes currents (id, iq) are divided into the iron losscurrents (icd, icq) and the magnetizing currents (imd, imq), asshown in Fig. 2.

Considering this condition and assuming that the saturation isneglected, the electromotive force is sinusoidal, and a cagelessrotor, the state equations of the dynamic model of the PMSM inthe synchronous reference frame, also taking into account theiron losses, are given by

dimd

dt=

1Ld

(vd − Rsid + ωLqimq) (1)

dimq

dt=

1Lq

(vq − Rsiq − ωLdimd − ωψPM ) (2)

dωr

dt=

1J

(Te − Dωr − TL) (3)

dθ

dt= ω = ωrp (4)

where

id =1

Rc

(Ld

dimd

dt− wLqimq + Rcimd

)(5)

iq =1

Rc

(Lq

dimq

dt+ ωLdimd + ωψPM + Rcimq

)(6)

icd = id − imd; icq = iq − imq (7)

Te =32p [ψPM imq + (Ld − Lq)imdimq] . (8)

Fig. 3. Required system voltage with variable-voltage control.

By referring (1), (2), (5), and (6) to a steady-state condition,it is possible to define a mathematical expression for the powerlosses in the stator windings. These losses can be calculatedaccording to the following expression:

PCu =32Rs

(i2d + i2q

). (9)

Similarly, the power losses that are caused by the fundamen-tal component of the total flux linkage in the iron stack can becalculated as

PFe =32Rc

(i2cd + i2cq

)

=32

ω2

Rc

[(Lqimq)2 + (ψPM + Ldimd)2

]. (10)

Although not considered in this paper, hysteresis losses canalso be taken into account. These losses are proportional to themachine phase current frequency. Therefore, to include theminto the machine model, the iron loss resistance Rc is usuallytreated as a function of ω.

III. VARIABLE-VOLTAGE CONTROL

Considering topology T2 and by adjusting the inverter supplyvoltage through a variable-voltage control strategy applied tothe bidirectional dc–dc boost converter, the system minimumlosses can be achieved according to the motor operating condi-tions. When the motor operates below its base speed, the systemvoltage does not need to be equal to the rated inverter supplyvoltage. Therefore, it can be concluded that, to minimize theoverall system losses, the system voltage must be proportionalto the PMSM back electromotive force (EMF). Taking intoaccount that the back EMF that is generated by a PMSM istypically proportional to its mechanical speed, the requiredsystem voltage can then be obtained according to the curveshown in Fig. 3.

When the motor speed is low, the inverter supply voltage willbe the lowest voltage of the boost converter, which is imposedby the battery. Above the motor base speed, in the constantpower zone, the dc–dc converter is controlled to supply theinverter with the system rated voltage. For the intermediaterange, the system voltage dynamically changes in accordancewith the machine mechanical speed.


Fig. 4. Block diagram of the control strategy for topology T2.

The overall performance can be even more improved if thesystem voltage also takes into account the PMSM load level.These two fundamental goals can simultaneously be consideredif the system voltage is controlled according to the modula-tion index m of the inverter space-vector PWM (SV-PWM)technique. Because it directly depends on the amplitude of thereference voltage vector and the inverter supplying voltage Vdc,the dc–dc converter can be controlled to impose a system volt-age that maintains the inverter modulation index at a specificreference value, as shown in Fig. 4. By doing so and assuminga large value for the reference modulation index, it is possible todynamically adjust and optimize the system voltage accordingto the motor mechanical operating conditions. Therefore, theoverall drive performance can greatly be improved by reducingthe voltage and currents distortion values and increasing theefficiency levels.

IV. SIMULATION RESULTS

The modeling and simulation of the PMSM drive systemsshown in Fig. 1 was carried out using the MATLAB/Simulinkenvironment, in association with the Power System Blockset.This software toolbox provides a great number of severalpower electronics mathematical models that were used for thesimulation. The PMSM model described in Section III wasimplemented in Simulink, as well as the battery model. Thislast approach is based on the work published in [31].

For both topologies, a rotor-field-oriented control strategythat employs proportional–integral PI current controllers andthe SV-PWM technique was applied to the inverter to controlthe PMSM mechanical speed. The PMSM parameters are sum-marized in Table VI, which is shown in the Appendix.

The switching frequency of the SV-PWM was chosen to beequal to 6 kHz. With regard to the topology in Fig. 1(a), theinverter is directly supplied by the battery with a rated voltageof 384 V. Considering the other topology, the system voltage iscontrolled by the bidirectional dc–dc converter within the rangeof 168–384 V (the former value corresponds to the batteryrated voltage). Assuming that the system rated voltage is 384 V,the battery voltage for topology T2 can be selected according

to several criteria. Beyond the size limitation, one importantaspect is the desired global drive maximum efficiency operatingpoint. Therefore, by choosing a battery voltage near the systemrated voltage, allows for achieving maximum efficiency at high-speed values. On the other hand, a lower battery voltage allowsfor achieving high efficiency at low-speed values.

For both topologies, the same insulated gate bipolar transis-tor (IGBT) inverter and PMSM were considered. The controlsystem that was implemented is described in Fig. 4. A valueof 9 kHz was chosen for the switching frequency of the PWMtechnique. The modulation index control loop is controlledwithin the linear range between 0 and 1 (overmodulation is notconsidered). Therefore and also taking into account that lowerharmonic distortion is obtained for a high modulation index[32], a reference value of 0.93 was assumed.

Two distinct operating conditions were considered: 1) whenthe machine operates as a motor (the motoring mode) and2) when it operates as a generator (regenerative braking). Forboth cases, several results are presented, with the aim of estab-lishing a performance comparison of both topologies.

To analyze all the considered cases, several performanceparameters are calculated. The efficiency values of the maindrive components, i.e., the dc–dc converter, the inverter, andthe machine, as well as the global drive efficiency values,are presented. All the efficiency values and power losses arecalculated through the analysis of the drive power flow, usingthe input and output power of each component as

η =Pout

Pin× 100% (11)

Plo = Pin − Pout. (12)

With regard to the PMSM, beyond the analysis of its powerfactor, the harmonic distortion values of its supplying voltagesare also investigated. To take into account the dc component andall the subharmonics, the distortion of the waveforms is moreproperly evaluated by calculating the total waveform distortion(TWD), which is defined as

TWD =

√X2

RMS − X21

X1× 100%. (13)

Finally, for all the considered operating conditions, a con-stant torque equivalent to 50% of the PMSM rated torque isassumed, together with a reference speed of 600 r/min. Consid-ering these speed and load values, the dc–dc converter variable-voltage control adjusts the system voltage to a reference valueof approximately 177 V.

Tables I and II present the efficiency results of the drive sys-tem main components for topologies T1 and T2 under the mo-toring and regenerative-braking operating modes, respectively.These results clearly demonstrate that higher global efficiencyresults can be achieved by topology T2. Furthermore, the resultsalso show that the inverter is the device that is mostly affected,because its efficiency considerably varies when comparing bothtopologies. Considering topology T1, the inverter is suppliedby its rated voltage, which corresponds to the battery voltage.This case means that, for the considered operating conditions,the modulation index is relatively low, and as a consequence,


TABLE IEFFICIENCY RESULTS (IN PERCENTAGE) UNDER THE MOTORING

OPERATION FOR 50% OF THE PMSM RATED TORQUE

TABLE IIEFFICIENCY RESULTS (IN PERCENTAGE) UNDER REGENERATIVE THE

BRAKING OPERATION FOR 50% OF THE PMSM RATED TORQUE

TABLE IIITWD AND POWER FACTOR RESULTS FOR 600 r/min AND 50% OF THE

PMSM RATED TORQUE

the inverter switching losses are larger compared to the lossesobtained for topology T2.

With regard to the voltage distortion values, the results inTable III show that, for both the motoring and regenerative-braking modes, with topology T2, the TWD values can ap-proximately be reduced by one-half. This also leads to thereduction of the motor phase currents distortion, as shown inFig. 5. These lower TWD values are justified by the fact thatthe system voltage imposed by the dc–dc converter is controlledby taking into account the chosen reference modulation index.This also leads to the improvement of the machine power factor.By reducing the voltage TWD, the corresponding root meanssquare (RMS) values will also decrease, as well as the PMSMapparent power, subsequently increasing the power factor.

V. EXPERIMENTAL RESULTS

The experimental setup comprises a battery pack, two IGBTSemikron power modules SKiiP 132GD120-3DU (one modulefor the inverter and the other module for the dc–dc converter),a dSPACE DS1103 digital controller, and a Yaskawa PMSMthat is coupled to a four-quadrant servomotor test system (seeFig. 6).

Fig. 5. Time-domain waveforms of the PMSM phase currents for 600 r/minand 25% of the motor rated torque. (a) Topology T1. (b) Topology T2.

Fig. 6. General view of the experimental setup.

The PMSM parameters can be found in the Appendix. Forthe dc–dc converter, an inductor of 10 mH was used, togetherwith an output filter capacitor of 1100 μF. The battery packcomprises several 12-V 12-Ah lead acid batteries that areconnected in series (32 and 14 batteries for topologies T1 andT2, respectively). Due to the limited number of dedicated PWMoutputs available at the controller, the modulation strategieswere implemented by software. The switching frequencies forthe inverter SV-PWM and the dc–dc converter PWM techniqueswere therefore chosen to be of 6 and 9 kHz, respectively.

The real-time interface board library for the DS1103 con-troller is designed as a common MATLAB/Simulink Blocksetthat provides blocks to implement the input/output (I/O) capa-bilities in Simulink models. The dSPACE ControlDesk soft-ware provides functions for real-time control and monitoring,allowing for simultaneously capturing data files that can beplotted using the MATLAB (see Fig. 7).

The control system was implemented for the DS1103 boardusing a sampling time of 25 μs. The voltage and current signals,as well as all the IGBTs gate commands, are connected to the


Fig. 7. General view of the real-time interface and measuring equipment.

Fig. 8. Block diagram with the main components of the experimental setup.

dSPACE controller trough interface and isolation boards (seeFig. 8). Two Yokogawa WT3000 digital power analyzers wereconnected in series in the power circuit to obtain the requireddata such as power flow, power factor, efficiency, and RMSvalues. The rotor position is obtained by an incremental encoderwith 1024 pulses per revolution, with the PMSM mechanicalspeed obtained by filtering the derivative of the rotor position.

For topology T1, the inverter is directly supplied by a systemrated voltage of 384 V. On the other side, considering thesecond topology, the drive system is supplied by a batterypack with a rated voltage of 168 V. The system voltage isdynamically adjusted by the dc–dc converter within the range of168–384 V. For the modulation index control, a reference valueof 0.93 is again assumed. The same inverter power module wasalso used for both topologies.

Experimental results are presented, considering the driveoperation under the motoring and regenerative-braking modes.To compare both topologies, particular attention is given to theefficiency analysis of the drive main components, including theglobal efficiency values. The PMSM voltages TWD and powerfactor results are also presented.

A. Motoring Operation

Fig. 9 presents an acceleration test for 50% of the PMSMrated load torque, showing the behavior of the variable-voltage

Fig. 9. Time-domain waveforms of the PMSM mechanical speed and systemvoltage for a change of the reference value from 500 r/min to 800 r/min.

Fig. 10. Inverter efficiency map for topology T1.

control for topology T2. At t = 1 s, a change of the referencespeed from 500 r/min to 800 r/min is introduced, with anacceleration rate of 100 r/s. The system voltage is kept constantand equal to 168 V when the motor operates below 550 r/min.Under these conditions, the modulation index is lower than theimposed reference value, and the dc–dc converter will feedthe inverter with the lowest voltage value, corresponding to thebattery voltage. When the machine accelerates above 550 r/min,the modulation index becomes equal to the reference value,and the variable-voltage control is enabled. Then, the systemvoltage proportionally increases to the PMSM speed until thenew reference speed is reached.

The results in Figs. 10–15 present the efficiency maps of theinverter and the PMSM, including the global drive efficiencyvalues for topologies T1 and T2.

Comparing the inverter efficiency maps for both topologiesin Fig. 10 and in Fig. 11, it is clearly shown that, withtopology T2, it is possible to obtain significantly higherefficiency values for the inverter. Moreover, for the operatingpoints above 600 r/min, the inverter efficiency is relatively highand constant compared to topology T1. This case is justifiedby the fact that the drive operates in the variable-voltage


Fig. 11. Inverter efficiency map for topology T2.

Fig. 12. PMSM efficiency map for topology T1.

Fig. 13. PMSM efficiency map for topology T2.

control zone, and consequently, the inverter losses, particularlythe switching losses, are minimized. Below this speed value, theinverter efficiency starts to more rapidly decrease, because thesystem voltage is imposed by the battery, and the modulation

Fig. 14. Global drive efficiency map for topology T1.

Fig. 15. Global drive efficiency map for topology T2.

index control is no longer possible. Nevertheless, the efficiencyvalues in this zone are still significantly higher for topology T2.

With regard to the PMSM, Figs. 12 and 13 present itsefficiency maps for the two considered topologies T1 andT2, respectively. For high speed and load levels, the PMSMpresents similar efficiency values for both topologies, becausethe system voltage is practically the same. However, the mostimportant difference is observed for load levels below 50%of the motor rated torque, where the PMSM efficiency fortopology T2 becomes higher than for topology T1. Again, dueto the variable-voltage control, the global machine efficiencycan be improved, particularly for low speed and torque values.By adjusting the inverter that supplies voltage through thecontrol of its modulation index, the PMSM voltage RMS valuesare minimized, reducing the machine losses, i.e., the iron lossesin the stator stack.

Figs. 14 and 15 present the efficiency maps of the globaldrive efficiency for topologies T1 and T2, respectively.Comparing these results, it can be concluded that, despite thehigher efficiency values of the inverter and the motor for topol-ogy T2, due to the dc–dc converter losses, the global efficiencyis negatively affected when the PMSM operates at high speed


Fig. 16. PMSM phase-to-phase voltage TWD values for topology T1.

Fig. 17. PMSM phase-to-phase voltage TWD values for topology T2.

and load levels. Under these conditions, the modulation indexfor topology T1 is naturally high. Therefore and also takingalso into account that there are losses only in the inverter andin the machine, the overall efficiency values will be higherthan for topology T2. However, for low speed values or, inparticular, for load levels below 50% of the motor rated torque,the global drive efficiency for topology T2 becomes muchhigher than for topology T1. Under these operating conditions,the modulation index control allows for decreasing the systemvoltage, resulting in lower losses in all drive components andhigher efficiency values.

Based on these tests, the maximum global drive efficiencywas achieved for topology T2 at 600 r/min and 3 Nm, with avalue of 90.2%. Considering topology T1, a value of 89.9%was achieved at 1200 r/min and 4 Nm.

The PMSM that supplies voltage distortion can significantlybe improved for topology T2 using the variable-voltage controltechnique. This approach is clearly shown in Figs. 16 and 17,where the TWD values of the motor phase-to-phase voltagesfor topologies T1 and T2 are, respectively, presented. Whenthe system voltage is directly imposed by the battery (topologyT1), the voltage distortion values are relatively high for low-speed operation, because the inverter modulation index is low.As the mechanical operation speed increases, the modulationindex also increases, and as a result, the TWD values decrease.It is also noticed that, for a given speed, the distortion values

Fig. 18. Time-domain waveforms of the PMSM phase currents for 600 r/minand 25% of the motor rated torque. (a) Topology T1. (b) Topology T2.

are lower for higher load levels due to the small increasingof the modulation index. With regard to topology T2, it canbe concluded that the overall distortion values are significantlylower than the values obtained for topology T1, in whichsome cases reduced by more than 50%. Note that, for thevariable-voltage control zone, approximately between 600 and1200 r/min, all the TWD values are relatively constant and low.As a result of the inverter modulation index control, the systemvoltage is constantly adjusted to maintain the modulation indexin its reference value. Thus, the voltage distortion is kept atrelatively low values, independent of the PMSM operatingconditions.

The voltage distortion also affects the PMSM phase currents,particularly for low speed and torque values, as shown inFig. 18. Therefore, the distortion of the motor phase currentsfor topology T1 [see Fig. 18(a)] will be larger than for topologyT2 [see Fig. 18(b)].

One important issue that is related to the distortion of thePMSM that supplies voltages and currents is the impact onmachine efficiency and its lifetime. Hysteresis and eddy currentlosses are part of the iron losses that are produced in themachine core due to the alternating magnetic field. Hysteresislosses are proportional to the frequency, whereas eddy currentlosses vary with its square value. Therefore, high-frequencyvoltage components produce additional losses in the PMSMcore, which, in turn, increase the operating temperature of thecore itself and also of the surrounding windings.

As shown, the application of a nonsinusoidal voltage to themotor results in harmonic current circulation in its windings.As a consequence, due to the skin effect, these losses willcontribute even more to the decrease of machine efficiency.These additional power losses also negatively affect the PMSMtemperature, this way contributing to the reduction of itslifetime.

Another important issue that is related to the harmonicdistortion is the noise generation. This aspect is particularlyimportant in electric/hybrid vehicles applications, because itis required to keep the noise levels to the minimum to ensurepassenger comfort. Accordingly, by reducing the PMSM that


Fig. 19. Power factor results for topologies T1 and T2 for 50% of the PMSMrated torque under the motoring operation.

TABLE IVEFFICIENCY RESULTS (IN PERCENTAGE) UNDER THE REGENERATIVE


supplies voltage TWD values with topology T2, it is possible todecrease the machine-generated noise.

Finally, Fig. 19 presents the PMSM power factor resultsfor several mechanical speed values and for a load torqueequivalent to 50% of the motor rated value. It is clearly shownthat higher power factor values can be obtained for topologyT2, because the variable-voltage control allows for reducing thevoltage RMS values and, therefore, the apparent power.

B. Regenerative-Braking Operation

Table IV presents the efficiency of the main drive compo-nents for topologies T1 and T2, including the global driveefficiency, for a load level that is equivalent to 50% of thePMSM rated torque under the regenerative-braking operation.

As expected, comparing both configurations, higher effi-ciency values can be obtained for topology T2, particularly forlow operating speeds. Under these conditions, it is shown thatthe global drive efficiency can significantly be improved fortopology T2, despite the power losses in the dc–dc converter.Similar to the motoring-mode operation, high-efficiency valuescan be obtained for the drive main devices, in particular forthe inverter. For low mechanical speeds, the variable-voltagecontrol imposes a lower system voltage to maintain the definedmodulation index. As a consequence, the inverter losses, i.e.,the switching losses, are reduced. This control strategy also al-lows for decreasing the motor voltage RMS values, which leadto the subsequent decreasing of the machine losses, particularlythe iron losses. However, for higher PMSM speed values, theglobal efficiency for topology T2 is lower than for topology T1.Although the inverter and PMSM efficiency values are higherfor topology T2, under these conditions, the dc–dc converter

TABLE VTWD AND POWER FACTOR RESULTS UNDER THE REGENERATIVE


Fig. 20. Considered drive cycle.

efficiency is negatively affected, contributing to the decrease ofthe overall drive efficiency.

Table V presents the PMSM power factor and phase-to-phasevoltage TWD values for both topologies and for a load levelthat is equivalent to 50% of the machine rated torque. Again, itcan be verified that, with the variable-voltage control applied totopology T2, it is possible to obtain lower distortion values forthe motor voltages and, therefore, for its phase currents, as wellas higher power factor values.

C. Drive-Cycle Analysis

To evaluate the drive dynamic behavior, for both topologies,under the motor and regenerative-braking modes, the analysisof a whole driving cycle is also presented.

Fig. 20 presents the drive cycle used for this purpose. Thisprofile is similar to the ECE 15 urban drive cycle, also in-tegrated in the first part of the New European Drive Cycle(NEDC) [33], [34].

Based on this drive cycle, Fig. 21 presents the time-domainwaveforms of the power drawn from the battery for both topolo-gies T1 and T2. During the acceleration periods, the powerthat is required by the drive system proportionally increasesto the machine speed and reaches larger values than for thesteady-state operation. In the steady state, the PMSM operateswith a constant mechanical speed, and the electromagnetictorque is less significant. As a consequence, the power that isdrawn from the battery is constant and relatively low. Duringthe deceleration periods, a braking torque is developed by themachine, resulting in a negative power that charges the battery.Comparing the two considered drive topologies (see Fig. 21),it becomes clear that topology T1 requires more battery outputpower than topology T2.


Fig. 21. Time-domain waveforms of the battery output power for topologiesT1 and T2 and for the considered drive cycle.

Fig. 22. Time-domain waveforms of the drive energy consumption for topolo-gies T1 and T2 and for the considered drive cycle.

Finally, considering the drive energy consumption resultsshown in Fig. 22, the energy increases when the system operatesunder the motoring mode and decreases due to the kineticenergy recovery under regenerative braking. However, the mostimportant conclusion is that, as expected, topology T2 is themore efficient topology for the considered drive cycle, consum-ing about 5.62 Wh, as opposed to the 5.98 Wh consumed bytopology T1. If a drive cycle that corresponds to an extraurbanprofile is considered, the PMSM will operate at higher speeds.Under these conditions and also taking into account the globaldrive efficiency maps presented in Figs. 14 and 15, topology T1will perform better than topology T2.

VI. CONCLUSION

A detailed efficiency analysis of two basic drive train topolo-gies based on a PMSM and applied to electric/hybrid vehicleshas been presented in this paper. The first topology comprises atraditional PWM battery-powered inverter, whereas in the sec-ond topology, the battery is connected to a bidirectional dc–dc

TABLE VIPARAMETERS OF THE PMSM USED

converter, which supplies the inverter. In addition, consideringthis second topology, a variable-voltage control technique thatis applied to the dc–dc converter was presented, with the aim ofoptimizing the global drive performance.

The presented results, in the form of efficiency maps, providea very good perspective about the torque/speed combinationsat which a specific propulsion motor drive or device is mostefficient. This case allows for better design of these advancedvehicular systems that make use of electric traction motors forpropulsion and also allow for choosing and optimizing the rightkind of technology to improve the system as a whole.

With regard to the results obtained, it can be concludedthat, with the variable-voltage control applied to the secondtopology, the inverter efficiency can significantly be improvedcompared to the first topology. The PMSM efficiency can alsobe enhanced, particularly for low load values. With regardto the global drive efficiency, it is concluded that, for theconsidered operating conditions, higher efficiency values canbe obtained for the second topology at low speeds and partialload. However, the inclusion of a dc–dc converter brings somedisadvantages such as increased complexity, cost, and size.Comparing both topologies, it can also be concluded that themaximum efficiency value for the first topology is achieved athigh speeds, whereas for the second topology, it is reached atlow speeds.

The presented voltage distortion and power factor resultsalso allow for demonstrating that better performance can beobtained for the second topology with the variable-voltage con-trol technique. The distortion results obtained are particularlyimportant, because high acoustic noise levels are typically as-sociated with high distortion values. Hence, this topology alsoallows for reducing the overall acoustic noise that is generatedby the traction motor.

Finally, considering all these factors and taking into accountthat, for an urban driving cycle, a traction machine most fre-quently operates at light loads and low speeds, the electric drivetrain system should be designed to operate at maximum effi-ciency and minimum acoustic noise in this region. Therefore,the presented variable-voltage control technique applied to thesecond topology allows for fulfilling these requirements.


APPENDIX

Table VI contains the parameters of the PMSM used in thispaper.

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Jorge O. Estima (S’08–M’10) was born in Aveiro,Portugal, in 1984. He received the Diploma degree inelectrical engineering in 2007 from the University ofCoimbra, Coimbra, Portugal, where he is currentlyworking toward the Ph.D. degree in the Departmentof Electrical and Computer Engineering.

He is also currently with the Instituto de Teleco-municações, Coimbra. His research interests includecondition monitoring and diagnostics of power elec-tronics and ac motor drives, fault-tolerant variable-speed ac drives, traction motor drives applied to

electric/hybrid vehicles, and wind energy conversion systems.

Antonio J. Marques Cardoso (S’89–A’95–SM’99)was born in Coimbra, Portugal, in 1962. He receivedthe Diploma degree in electrical engineering and theDr. Eng. and Habilitation degrees from the Univer-sity of Coimbra, Coimbra, in 1985, 1995, and 2008,respectively.

From 1985 to 2011, he was with the Universityof Coimbra, where he was the Director of the Lab-oratory of Electrical Machines. Since 2011, he hasbeen with the University of Beira Interior (UBI),Covilhã, Portugal, where he is a Full Professor with

the Department of Electromechanical Engineering. He is also currently with theInstituto de Telecomunicações, Coimbra. His teaching interests cover electricalrotating machines, transformers, and the maintenance of electromechatronicsystems. His research interests include the condition monitoring and diag-nostics of electrical machines and drives. He is the author of the book FaultDiagnosis in Three-Phase Induction Motors (Coimbra, Portugal: CoimbraEditora, 1991; in Portuguese) and about 300 papers that were published intechnical journals and conference proceedings.

Efficiency Analysis of Drive Train Topologies Applied to Electric/Hybrid Vehicles

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