13

Chapter-2

RECONFIGURABLE MICROSTRIP ANTENNAS

2.1 INTRODUCTION

Reconfigurability, when used in the context of antennas, is the capacity to

change an individual radiator’s fundamental operating characteristics through

electrical, mechanical, or other means. Thus, under this definition, the traditional

phasing of signals between elements in an array to achieve beam forming and

beam steering does not make the antenna “reconfigurable” because the antenna’s

basic operating characteristics remain unchanged in this case. Ideally,

reconfigurable antennas should be able to alter their operating frequencies,

impedance bandwidths, polarizations, and radiation patterns independently to

accommodate changing operating requirements. However, the development of

these antennas poses significant challenges to both antenna and system designers.

These challenges lie not only in obtaining the desired levels of antenna

functionality but also in integrating this functionality into complete systems to

arrive at efficient and cost-effective solutions. As in many cases of technology

development, most of the system cost will come not from the antenna but the

surrounding technologies that enable reconfigurability [45].

2.2 NECESSITY OF RECONFIGURABILITY

Let us consider two general application areas, single-element scenarios and

array scenarios, In single-element scenarios an antenna used in portable wireless

devices, such as a cellular telephone, a personal digital assistant, or a laptop

computer. Single antennas typically used in these devices are monopole or

microstrip antenna based and may or may not have multiple-frequency

capabilities. Some packages may use two or three antennas for diversity reception

on small devices to increase the probability of receiving a usable signal, but

14

usually only one of the antennas is used for transmission. The transmission from

the portable device to a base station or other access point is the weakest part of the

bidirectional communication link because of the power, size, and cost restrictions

imposed by portability. Moreover, the portable device is often used in

unpredictable and/or harsh electromagnetic conditions, resulting in antenna

performance that is certainly less than optimal. Antenna reconfigurability in such a

situation could provide numerous advantages. For instance, the ability to tune the

antenna’s operating frequency could be utilized to change operating bands, filter

out interfering signals, or tune the antenna to account for a new environment. If

the antenna’s radiation pattern could be changed, it could be redirected toward the

access point and use less power for transmission, resulting in a significant savings

in battery power. The antennas are mostly used in array configuration, feed

structures with power dividers/combiners and phase shifters. For instance, current

planar phased array radar technology is typically limited in both scan angle and

frequency bandwidth as a result of the limitations of the individual array elements

and the restrictions on antenna element spacing. This restriction comes from

mutual coupling effect on one hand, appearance on grating lobe on other hand.

Many of these established applications assume that the antenna element pattern is

fixed, all of the elements are identical, and the elements lie on a periodic grid. The

addition of reconfigurability to antenna arrays can provide additional degrees of

freedom that may result in wider instantaneous frequency bandwidths, more

extensive scan volumes, and radiation patterns with control on side lobe

distributions [46].

There are several antenna structures that are suitable for implementation of

reconfigurable antennas, Among them microstrip patch antennas are very

attractive structures for various types of reconfigurable antennas, all such antennas

are usually equipped with switches that are controlled by DC bias signals. Upon

toggling the switch between on and off states, the antenna can be reconfigured

[47]. The following section describes the design procedure of microstrip patch

antenna types presented and different feed types used in this dissertation.

15

2.3 MICROSTRIP PATCH ANTENNA DESIGN

A microstrip antenna in its simplest configuration consists of a radiating patch on

one side of a dielectric substrate, which has a ground plane on the other side. The

patch conductors usually made of copper or gold can be virtually assumed to be of

any shape. However, conventional shapes are normally used to simplify analysis

and performance prediction. The radiating elements and the feed lines are usually

photo etched on the dielectric substrate.

The radiating patch may be square, rectangular, circular, ring, elliptical or any

other configuration. Square, rectangular and circular shapes are the most common

because of ease of analysis and fabrication. Some of the advantages of the

microstrip antennas compared to conventional microwave antennas are

Low weight, low volume,

Low fabrication cost,

Easy mass production,

Linear and circular polarization are possible with simple feed,

Easily integrated with MIC,

Feed lines and matching networks can be fabricated simultaneously with

antenna structures.

Patch antennas find a variety of applications starting from military to commercial,

because of their ease of design and fabrication. Patch arrays are extensively used

in phased array radar applications and in applications requiring high directivity

and narrow beam width.

One of the first tasks in the printed antenna design is the selection of a suitable

substrate material. The major electrical parameters to be considered are the

relative dielectric constant and the loss tangent. A higher dielectric constant results

in smaller patch antenna but generally reduces bandwidth and results in tighter

fabrication tolerances. A high loss tangent reduces the antenna efficiency and

increases feed losses. As a rule of thumb, select a substrate with lowest possible

dielectric constant. Substrate thickness is chosen as large as possible to maximize

16

bandwidth and efficiency but not so large to allow surface-wave excitation. For a

maximum operating frequency of ‘f’, the substrate height ‘h’ should satisfy

h ≤ . √ (2.1)

Where ‘c’ is the speed of light and ‘εr’ is the relative dielectric constant of the

substrate material.

2.3.1 Design Procedure of the Microstrip Rectangular Patch Antenna

The rectangular patch antenna is simplest form of patch antenna as shown in

Fig.2.1 and usually designed to operate near the resonance. The length ‘L’ of the

patch radiator is then selected such that it satisfies the condition of resonance. It is

usually chosen close to λ/2 such that the input impedance of the patch is pure real

at the desired frequency. Since the two ends of the patch are open, an open-end

correction is usually taken into account while calculating the physical length ‘L’ of

the patch.

Figure 2.1 Geometry of rectangular microstrip patch antenna

The patch width ‘W’ generally lies between 0.5 to 2 times ‘L’. The width

can be used to vary the input impedance of the patch. As we have to ultimately

match the patch input impedance to 50-ohms, we can control input impedance to

some extent by changing the patch width ‘W’. If ‘W’ is chosen very small, the

antenna radiation efficiency will be reduced. So there is tradeoff between the input

17

impedance and radiation efficiency. Once W and L are selected, we can calculate

the input impedance of the patch and then use an impedance transformer to match

this impedance to the 50-ohm feed line [48]. The design of a rectangular

microstrip patch antenna begins with (a) choice of a substrate (b) selecting feed

mechanism, (c) determining patch width w and (d) selecting the feed location. The

microstrip rectangular structure as shown in Fig.2.1has been used to implement a

reconfigurable rectangular patch antenna and triple-polarization diamond shape

antenna in this report.

In the following steps rectangular patch antenna design procedure is given after

choosing the its essential parameters such as frequency of operation (fr ),dielectric

constant of the substrate (ε ), height of dielectric substrate (h).

1) Calculation of Width (W):

For an efficient radiator, a practical width that leads to good radiation

efficiencies is given as

W

(2.2)

With the substituting the values of c= 3x108

m/s, fr and ε by by using the

the above formula ‘W’ can be calculated.

2) Calculation of Effective dielectric constant ( ε ): The effective dielectric constant due to the air dielectric boundary is given

by

ε ε ε

1 12 /

(2.3)

With the substituting the values ε ,h and W effective dielectric constant

ε can be calculated using the above formula.

3) Calculation of effective length (Leff):

18

By choosing the substrate, the width and length of the patch can be

estimated. An initial approximation for the length can be made for a half

wave microstrip antenna radiated by the formula:

L !"" #

$ (2.4)

With the substituting the values of c= 3x108 m/s, fr and ε by using the

the above formula ‘Leff’ can be calculated.

4) Calculation of the length extension(∆L):

Due to the fringing fields along the radiating edges of the antenna, there is a

Line extension associated with the patch, which is given by the formula

∆L 0.412h)#!"".$*+.,-

#!""../$*+./0 (2.5)

With the values from h, w and ε by using the the above formula ‘∆L’

can be calculated.

5) Calculation of the length of the patch (L):

The length of the patch can now be calculated as

L = Leff - 2∆L (2.6)

With the values from Leff and ∆L by using the above formula ‘L’ can be

calculated.

6) Calculation of the ground plane dimensions:

The transmission line model is applicable to infinite ground planes only.

However, for practical considerations, it is essential to have a finite ground

plane. It has been shown by [48] that similar results for finite and infinite

ground plane can be obtained if the size of the ground plane is greater than

the patch dimensions by approximately six times the substrate thickness (h)

19

all around the periphery. Hence,the ground plane dimensions would be

given as:

Lg = 12h + L, Wg = 12h + W (2.7)

7) The feed point determination:

All feeding excitation port is to be matched to 50Ω. In order to have a

matching of the impedance the connecter has to be placed at some distance

from the edge which has a match of 50Ω. Normally, there is a trial and

error method that has been adopted to check the minimum value of the

Return loss. The feed point initially kept in the center of X-axis and varies

in the Y-direction to get the optimum return loss.

2.3.2 Design Procedure of the Microstrip Ring Patch Antenna

Ring microstrip antennas of various shapes, such as circular, rectangular, square

and triangular have been studied as alternatives to standard rectangular and

circular disks. These antennas are geometrically and electrically an intermediate

configuration between a printed loop and a patch. The microstrip ring structure as

shown in Fig.2.2 has been used to implement a wheel antenna in this report.The

details of the design of this antenna are given next.

Figure 2.2 Geometry of ring microstrip patch antenna

20

Like many forms of microstrip patches, the annular ring has received considerable

attention. When operated in its fundamental (TM11) mode, this printed antenna is

smaller than its rectangular or circular counterparts. The annular ring may also be

somewhat broadband in nature when operated near the TM12 resonance. It has

been shown that the structure is a good resonator (with very little radiation) for

TM1m modes (m odd), and a good radiator for TM1m modes (m even) [48].

The circular ring has been analyzed extensively using the cavity model because

the cavity model analysis is found to be simple. This can be accomplished using a

procedure similar to that for the rectangular patch but now using cylindrical

coordinates [49].

Figure 2.3 Geometry of circular ring in cylindrical coordinates

The cavity model of the ring is obtained by replacing its peripheries with magnetic

walls. Because there is no variation of the fields along the z direction for thin

substrates, the modes are designated as TMnm modes. With no excitation current,

the wave equation for electrical field can be written as

#1 2$ 0 (2.8)

Where

2 23 45/6 (2.9)

21

The general solution for the wave Eq.2.8 in cylindrical coordinates is given as

78 7)9:#2;$<:=#2>$ ? 9:= #2;$<:#2>$0@ABCD (2.10)

and

EF GHIF

JKLJM (2.11)

EM GHI JKLJF (2.12)

Where Jn(.) and <n(.) are the Bessel functions of the first and second kind,

and of order n, respectively. The other field components are zero inside the cavity.

The surface current on the lower surface of ring metallization is given by

9PQ ?R T EUUP ?DVEF ;WEM (2.13)

or

9M G:KXHIF )9:#2;$<:=#2>$ ? 9:#2>$<:#2;$0BYCCD (2.14)

9F G:KXHI )9:= #2;$<:=#2>$ ? 9:#2>$<:=#2;$0@ABCD (2.15)

The radial component of the surface current must vanish along the edges at ; = >

and ;= Z to satisfy the magnetic wall boundary conditions. This gives

9F#; Z$ EM#; Z$ 0 (2.16)

Application of this boundary condition leads tothe well- known characteristic

equation for the resonant modes:

9:= #2Z$<:=#2>$ ? 9:= #2>$<:=#2Z$ 0 (2.17)

For the given values of >, b, εr and n, the frequency is varied and the roots of the

characteristic equation for the resonant modes are determined. These roots are

denoted by 2nm for the resonant ]Mnm modes and form _nm such that

_nm 2nm.a (2.18)

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The integer C denotes the azimuthal variation as per Cosnϕ, while the integer a represents the ath

zero of the characteristic equation and denotes the variation of

fields across the width of the ring.

If C = b/a then Eq.2.17 can be written as

9:= #b_:c$<:=#_:c$ ? 9:= #_:c$<:=#b_:c$ 0 (2.19)

Using zeroth-order approximation,The resonant frequency is obtained as,

d:c efg hij√kl (2.20)

where

@= velocity of light in free space

> = Inner radii of ring

4m= dielectric constant of substrate

In Eq.2.20, the effect of the fringing fields has not been considered. Thus the

frequency calculated by this formula is lower than the measured value. The

accuracy can be improved by using effective dielectric constant (εre).

d:c efg hij kln (2.21)

To determine the value of 4re, the ring resonator is modeled as a microstrip line

bent in a circular shape. The effect of the curvature on the resonant frequency is

expected to be small provided the radius of curvature is large compared with the

width of the strip conductor. The effective dielectric constant can be determined as

p5q r sltuv slwuv xy/ (2.22)

Where

z = b-a

Z = Outer radii of the ring

ℎ=thickness of dielectric

23

The modified values of the inner and outer radii of the ring can be determined

using parallel plate waveguide model of a microstrip line and are given by

ae = a –(We-W)/2 (2.23)

be = b + (We-W)/2 (2.24)

Where z|is the effective width of the ring and can be given by,

W#f$ W #!#$$)~ ""

v0 (2.25)

W#0$ (2.26)

f (2.27)

Where μis the permeability and’ z’is the quasi-static characteristic impedance of

microstrip line of width W.

A pair of empirical formulas for the modified radii of the ring is

ae = a - - (2.28)

be = b + - (2.29)

The above model gives reasonably accurate results as long as z| is less than the

mean diameter of the ring, i.e. (> + Z).

An approximate value of _C1= kC1acan be obtained using the Eq. 2.31. This

expression gives reasonably accurate value of kn1 for C ≤ 5 and

)#$#$0< 0.35 (2.30)

_C1 = kC1a = #$ (2.31)

Based on the cavity model formulation described above a design procedure is

outlined which leads to practical designs of circular ring microstrip antennas for

the dominant TMz110 mode. The procedure assumes that the specified information

includes the dielectric constant of the substrate (ℰr ), the resonant frequency (fr )

and the height of the substrate h.

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2.3.3 Design Procedure of the Microstrip Meander line Patch Antenna [4]

In microstrip, two types of transmission line antennas are possible, the

travelling wave antenna where the end of the microstrip line is terminated in a

resistive match and the standing wave antenna where the microstrip line is

terminated by an open or short circuit. Standing wave antennas radiate a broadside

beam while travelling-wave antennas may be designed to radiate a beam in any

direction between broad side and end fire. The meander line antenna is a standing

wave antenna realized as a periodic structure with broadside beam radiation.

The lowest resonance of the meander line antenna occurs when the length of the

Meander line antenna is approximately half of the operating wavelength [50]. The

antenna element also resonates when the total length of the antenna is in multiples

of the wavelength. The studies by Jalil[51] and Hanyang[50] show that the

resonant frequency of the antenna is inversely proportional to the number of

meander line sections within a constant physical length .

The microstrip meander line structure as shown in Fig.2.4b has been used to

implement a reconfigurable multiband meander line antenna in this report. The

details of the design of this antenna are given next.

The meander line element consists of a meandering microstrip line formed by a

series of sets of right angled bends. The fundamental element in this case is

formed by four right angled bends as shown in Fig.2.4 (a). The right angled bends

are chamfered to reduce the right angled discontinuity susceptance for impedance

matching.

(a)

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The radiation from a meander line antenna mainly occurs from the discontinuities

(bends) of the structure. The radiation mechanism of a travelling wave meander

line (rampart line) antenna array has been reported by P.S.Hall [52-53]. The

radiations from a meander line antenna originate from the right angle bends and

that can be represented by a small magnetic current element located diagonally.

This gives rise to radiated field polarizations from each bend as shown in

Fig.2.4(c), where the vector ‘E’ denotes the Electric field (E-field) direction.

Figure 2.4 (a) Fundamental meander line with 4 bends

(b) Geometry of the Meander antenna element

(c) Radiated field polarizations from fundamental meander line antenna

E E

E E

s

d

s

l

(c)

Y

X

(b)

L

d

s

Feed point

26

The beamwidth and bandwidth are both controlled by the meander line

parameters viz. spacing between the short segments(s), spacing between the long

segments(d) and the width of the meander line(w) as defined in Fig.2.4(b). The

antenna will resonate when the total length of the meander line equals one or more

than one guided wavelength. In the case of meander line antenna, the resonant

length of the patch is more than its total physical length (The equivalent length

when all the short and long segments added together). This is due to the effects of

the bends and the open ended lines [53]. The element has been designed to give

two resonances where they are peaking close to each other in order to get a

wideband performance at the desired frequency band. The lower and upper

resonances in this case occur when the total length (Lt) of the antenna becomes

multiples of the fundamental resonant length. The fundamental element will

resonate when the total length is equal to λg/2 or its multiples, where λg is the

wavelength in the dielectric material. Here, the introduction of the multiple

meander sections gives a bandwidth enhancement by shifting the higher resonance

away from the lower resonance. The radiations from the bends add up to produce

the desired polarization which in turn depends on the dimensions of the meander

line antenna. The spacing between two bends are very critical, where if the bends

are too close to each other, then cross coupling will be more, which affects the

polarization purity of the resultant radiation pattern. In other case, the spacing is

limited due to the available array grid space and also the polarization of the

radiated field will vary with the spacing between the bends and the spacing

between the microstrip lines [54-55].

The radiation from a meander line antenna is diagonally polarized, when

one cell of the meander line antenna is considered as shown in Fig.2.4(c). If we

assume that the bends are all identical and matched then the total radiated field

from one period is given by [52].

27

7φ#θ$ b@ABθ |G#t$v βGv sin Qβ sin Q β?

?|#G$#Q$βG sin β sin β ?

(2.32)

7θ#θ$ b|G#t$v βGv sin Qβ cos Q β ?

|#G$#Q$βG sin β cos β ?

(2.33)

?2 sin θ (2.34)

Where

θ = Elevation angle in Cartesian coordinates

φ = Azimuth angle in Cartesian coordinates

β = 2π/λg, the microstrip wave number

k0 = 2π/λ0, the free space wave number

C = Radiated field strength

Here, the |Gω time dependence is suppressed

2.3.4 Design Procedure of the Microstrip Square Spiral Patch Antenna

Spiral antennas belong to the class of frequency independent antennas which

operate over a wide range of frequencies. Polarization, radiation pattern and

impedance of such antennas remain unchanged over large bandwidth [56]. Such

antennas are inherently circularly polarized with low gain. Array of spiral antennas

can be used to increase the gain. Spiral antennas are reduced size antennas with its

windings making it an extremely small structure [57-59]. It consists of one or

multiple arms as well. Lossy cavities are usually placed at the back to eliminate

back lobes because a unidirectional pattern is usually preferred in such antennas.

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Spiral antennas are classified into different types; archimedean spiral, square spiral

and star spiral etc. Archimedean spiral is the most popular configuration.

These antennas operate in 3W way; travelling wave, fast wave and leaky wave.

The travelling wave, formed on spiral arms, allows for broadband performance,

fast wave due to mutual coupling phenomenon occurring between arms of spiral

and leaky wave leaks the energy during propagation through the spiral arms to

produce radiation. Ring theory (band theory) explains the working principle of

spiral antenna. The theory states that spiral antenna radiates from a region called

active region [60] where the circumference of spiral equals wavelength.

Different design parameters are to be considered while designing a square spiral

antenna as shown in Fig. 2.5 include spacing between the turns 's', width of arm

'w', inner radius 'r1' and outer radius 'r2’. The inner radius is measured from centre

of the spiral to centre of the first turn while the outer radius is measured from

centre of the spiral to centre of the outermost turn.

Figure 2.5 Geometry of single arm square microstrip patch antenna

Other than these design parameters, spiral antennas have lowest and highest

operating frequencies which can be calculated from the following relations where

'c' corresponds to speed of light:

f c 2πr¡ (2.35)

29

f¢£ c 2πr¡ (2.36)

In an rθ coordinate system, spiral grows along r-axis and θ-axis simultaneously.

All spirals satisfy following equation where 'a' corresponds to growth factor and 'b'

corresponds to multiplication factor:

r = a + bθ (2.37)

Different designs of spiral antenna can be obtained by varying number of turns it

contains, the spacing between its turns and the width of its arm.

2.4 FEED MECHANISMS OF MICROSTRIP ANTENNAS

Microstrip antennas have radiating elements on one side of a dielectric substrate,

and thus early microstrip antennas were fed either by a microstrip line or a coaxial

probe through the ground plane. Since then, a number of new feeding techniques

have been developed. Prominent among these are coaxial feed, microstrip

(coplanar) feed, proximity-coupled microstrip feed, aperture coupled microstrip

feed, and coplanar waveguide feed. Selection of feeding technique is governed by

a number of factors. The most important consideration is the efficient transfer of

power between the radiating structure and the feed structure, that is, impedance

matching between the two. Another consideration is the suitability of the feed for

array applications.

2.4.1 Coaxial Feed/Probe Coupling

Coupling of power through a probe is one of the basic mechanisms for the transfer

of microwave power. The probe can be an inner conductor of a coaxial line in the

case of a coaxial line feeding or it can be used to transfer power from a triplate

line (strip line) to a microstrip antenna through a slot in the common ground plane.

A typical microstrip antenna using a coaxial connector is shown in the Fig.2.6.

The coaxial connector is attached to the back side of the PCB, and the center

conductor after passing through the substrate is soldered to the patch metallization.

30

Figure 2.6 Coaxial probe feeding of a microstrip antenna

The location of the feed point is determined for the given mode so that the best

impedance match is achieved. Excitation of the patch occurs principally through

the coupling of the feed current Jz to the Ez field of the patch mode [61]. The

coupling constant can be obtained as:

Coupling © ª EJdv © cos#πx/L$¯ (2.38)

Where L is the resonant length of the patch, and x0 is the offset of the feed point

from the patch edge. Eq.2.38 shows that coupling is maximum for a feed located

at a radiating edge of the patch(x0 = 0 or L).

31

Figure 2.7 Equivalent circuit of a microstrip antenna fed by a coaxial probe

Figure 2.8 Modified equivalent circuit of a microstrip antenna fed by a coaxial

Probe

The coaxial feed, using Huygen’s principle, can be modeled by a cylindrical band

of electric current flowing on the center conductor from the bottom to the top

along with an annular ribbon of magnetic current in the ground plane. An

idealization that simplifies the computation is to replace the electric current

cylinder by a uniform line current ribbon. In order to determine the probe

impedance for a microstrip antenna, the canonical problem of a parallel plate

waveguide fed by a coaxial line has been analyzed using the integral equation

formulation [62]. The input impedance of this geometry shown in Fig.2.6 has been

determined.

32

In an approximate analysis, an excitation field corresponding to TEM field

distribution in the annular ring about the probe gives fairly accurate result [17].

The resulting impedance can be modeled by an equivalent circuit, as shown in

Fig.2.7 the modified equivalent circuit is shown in Fig.2.8. A number of other

approaches for the analysis of a probe fed microstrip antenna are described in the

literature [63-66].

Coaxial feed has the advantage of simplicity of design through the

positioning of the feed point to adjust the input impedance level. But it has several

limitations. First, coaxial feeding of an array requires a large number of solder

joints, which makes fabrication difficult and compromises reliability. Second, for

increased bandwidth of a patch antenna, a thicker substrate is used and therefore

requires a longer probe. This gives rise to an increase in spurious radiation from

the probe, increased surface wave power and increased feed inductance. However,

the feed inductance can be compensated for by a series capacitor. One approach to

introduce a capacitor in series with the probe is to etch out an annular slot in the

patch metallization concentric with the probe. However .the electrical field in the

annular region will introduce cross-polar components in the radiation pattern. The

antenna impedance can be transformed to a desired value by introducing

electromagnetic coupling between the patch and the probe. The coupling region is

increased by terminating the probe in a disk, and positioning the disk below the

patch. The input impedance depends on the disk size, spacing of the disk from the

patch, and the probe position.

2.4.2 Microstrip Edge Feed

Excitation of the microstrip antenna by a microstrip line on the same substrate is

the natural choice because the patch can be considered an extension of the

microstrip line, and these both can be fabricated simultaneously. The coupling

between the microstrip line and the patch is in the form of the edge or butt-in

coupling as shown in the Fig.2.9.

33

Figure 2.9 Antenna with microstrip edge feed

The edge-coupled microstrip feed can be modeled by means of the step-in-width /

impedance junction. The equivalent circuit for this is shown in Fig.2.10. A design

based on edge-coupled feed is available in [67], and an analysis of the edge

coupled feed based on the FDTD approach is described by Vandenbosch G.A.E

and A.R.Van de Capelle [68].

Figure 2.10 Equivalent circuit of edge feeding

The edge-coupled feed described above suffers from a limitation of impedance

mismatch because the input impedance of the patch at its radiating edge is very

high compared to the 50Ω impedance of the feed line. Therefore, an external

impedance matching circuit is used between the patch edge and the 50Ω

34

microstrip line. The impedance matching circuit, besides giving rise to spurious

radiation, cannot be accommodated in arrays, because of the non-availability of

physical space on the substrate. The microstrip line blocks radiation from the

portion of the patch with which it is in contact resulting in reduced radiation. This

is a serious limitation of this feed at millimeter wave frequencies where the patch

width can be comparable to the width of the microstrip line.

2.4.3 Microstrip Inset Feed

This type of feed overcomes the some of the short comings of microstrip edge

feed. Here, the microstrip line is inset into the patch, a diagram of an inset fed

patch is shown in Fig.2.11, where Y represents the inset length. The approximate

equivalent circuit for the inset microstrip feed is shown in Fig.2.12.The purpose of

the inset cut in the patch is to match the impedance of the feed line to the patch

without the need for any additional matching element. This is achieved by

properly controlling the inset position. Hence this is an easy feeding scheme, since

it provides ease of fabrication and simplicity in modeling as well as impedance

matching.A parametric study of rectangular patch with various inset lengths Y has

been reported [69].

Figure 2.11 Antenna with microstrip inset feed

35

Figure 2.12 Equivalent circuit of inset feeding

The input impedance for an inset fed patch is given by the simplified expression

[71].

Zin (Y) = Cos2(i°± $Zin(0) (2.39)

Here Zin(0) is the input impedance if the patch was fed at the end. Hence by

feeding the patch antenna as shown Fig.2.11, the input impedance can be

decreased. This method can thus be used to tune the input impedance to the

desired value. The feed position is selected such that the input impedance of the

antenna is 50Ω. The resulting feed point is about the same as that used for coaxial

feed.

2.5 RECONFIGURABLE ANTENNA DESIGN METHODOLOGIES

This section discusses the different methodologies that are used to design

reconfigurable antennas. It describes each method and illustrates the strengths and

weaknesses associated with each. Classical non-reconfigurable design methods

have dominated antenna engineering for the majority of antenna design history. In

order to make the transformation from fixed element operation to reconfigurable

antenna design requires a suitable conversion in design methodology. There are

three broad methodologies that have been identified for achieving reconfigurable

36

antenna designs and operation namely antenna geometry morphing, matching

network morphing and smart geometry reconfiguration. Antenna geometry

morphing represents the most structurally complicated of the methods. It is

implemented through a large array of switchable sub-elements which are

combined to form the desired radiating structure. Feed geometry morphing is the

simplest of the methods and modifies only the feed structure or impedance

matching network of the antenna while the radiating structure remains constant.

The smart geometry reconfiguration method lies between the other two in its

structural implementation complexity. It modifies only critical parameters of the

antenna radiating structure to achieve the desired range of reconfigurable control

[72].

2.5.1 Antenna Geometry Morphing

The antenna geometry morphing method achieves reconfigurable operation by

switching a large array of interconnected sub-elements. The sub-elements are

connected together via RF switches and are typically less than ² size

(conductors). Because the sub-elements are much less than a wavelength in size,

they do not form efficient radiating elements individually. However, switching

together multiple adjacent sub-elements results in an aggregate structure that

forms the desired radiator. This sub-element arraying allows considerable

flexibility in forming the radiator. The geometry of the aggregate radiating

structure can take a wide variety of forms depending on the desired application.

The reconfigurable antennas designed via this method are often referred to in the

literature as distributed radiators because the total radiating structure is distributed

over many smaller structures. Fig.2.13 illustrates the concept of the antenna

geometry morphing method. The example is a reconfigurable microstrip patch

antenna consisting of a large grid of switched microstrip sub-patches that are

available on the dielectric substrate. These sub-patches do not represent individual

microstrip patch antennas themselves but act as actively reconfigurable conducting

structures. The detailed blow-up in Fig.2.13 shows a single functional unit cell for

37

the composite antenna. These unit cells illustrate the concept of the sub-patch

conducting structure. Each unit cell consists of a small conducting patch of metal

and four RF micro switches. The switches provide the RF conduction path to the

nearest neighboring unit cell. The composite antenna is then constructed by

activating the necessary switches to form then antenna. In this example, the

structure is first configured to form a conventional rectangular microstrip patch

antenna. Next, several of the sub-patches along the length of the microstrip feed

are switched to the off state. This moves the effective feed point for the patch

antenna closer to the center of the patch and alters the input impedance of the

patch. Finally, the unit cells are configured to form a bow-tie patch antenna which

has different radiation characteristics than the rectangular patch.

Figure 2.13 Reconfigurable Antenna structure [72]

The antenna geometry morphing method has the obvious advantage of providing a

large amount of antenna reconfigurability. The array of sub-elements provides a

large level of flexibility in composing the aggregate antenna. Because of the

flexibility in configuring the antenna, a wide range of control over many antenna

characteristics is offered by employing this method. Thus, a single reconfigurable

platform could be used for a large number of applications. System operation over

38

multiple frequency bands, with variable radiation pattern characteristics and

selective polarization is possible with a single reconfigurable platform. Likewise,

the layout of the sub-element array pattern is not limited to two dimensional planar

microstrip geometries. Surface conformal and three dimensional geometries also

represent viable configurations. The highly flexible nature of total geometry

morphing dictates the primary difficulty with implementing this method. The

extreme complexity involving numerous individual components to necessary to

realize the geometries is inherent in this method. A large number of sub-elements,

switches and control lines are required to implement the reconfigurable geometry.

This leads directly to geometry and component management issues. A large

number of active components also means there are a large number of points of

failure. Recent advancements in RF switches have been the driving force behind

much the reconfigurable antenna designs. As with any mechanical switch, RF

switches are susceptible to reliability issues due to mechanical fatigue. Thus, any

structure which depends explicitly on the reliable operation of these switches will

be subject to performance degradation in the case of switch failure.

2.5.2 Feed Geometry Morphing Technique

The feed geometry morphing method represents the simplest of the three

techniques for achieving reconfigurable antenna operation. In this method, the

actual radiating structure remains constant and only the feed or impedance

matching section of the antenna is reconfigured. Like the antenna geometry

morphing method, this method is often employed with microstrip geometries

because of the relative ease in placing RF switches on planar structures. In the

case of microstrip feed lines, there are typically 10 or more sub elements in the

transverse direction across the width of the microstrip line for adequate parameter

control. They are on the order of ² in length along the longitudinal direction. Fig.

2.14 illustrates one implementation of the matching network morphing method. In

this example a microstrip patch antenna is edge fed by a reconfigurable microstrip

39

line. The reconfigurable microstrip line consists of a small array of switchable

microstrip sub-elements. Each of these sub-elements may be switched on or off by

activating one of the miniature RF switches that form the interconnections

between the sub-elements and compose the overall microstrip structure. The width

and length of the feed line is altered to change the impedance of the microstrip.

The grey boxes represent inactive sub-elements and the a boxes represent active

sub-elements. The top left configuration in Fig. 2.14 shows the microstrip patch

antenna and the available microstrip feed lattice. The top right sub-element

arrangement shows the feed configured as a narrow microstrip line having a

characteristic impedance. The patch antenna operates in a radiation mode that is

specified by this feed configuration. The bottom two arrangements in Fig.2.14

show the microstrip feed line configured in two other possible formations. These

variations in feed impedance then excite different radiation modes in the

microstrip patch antenna.

The matching network morphing technique carries the distinct advantage of being

extremely simple to implement in practice. The only component of the antenna

that is changed is the feed network and thus the complexity of the design is

minimized. As a result, the number of physical switching components is kept to a

minimum and switch reliability becomes less of an issue. Conversely, this method

exhibits the disadvantage of limited antenna reconfigurability. The antenna

operation is varied only through changes in matching. Consideration is not given

to other critical radiation characteristics. Because the principal radiation mode is

altered by the impedance, the electrical performance characteristics are likely to

change as well.

40

Figure 2.14 Feed geometry morphing technique [72]

2.5.3 Smart Geometry Reconfiguration Method

The third method of reconfigurable antenna design is smart geometry

reconfiguration. Falling between antenna geometry morphing and the feed

geometry morphing method in both the amount of achievable parameter control

and system complexity, this method modifies only critical parameters of the

antenna radiating structure to achieve the desired reconfigurable performance. It

can be implemented with considerably fewer control elements than the antenna

geometry morphing method and thus has the advantage of reduced design

complexity. However, with a thorough understanding of the underlying antenna

design and careful design consideration, it can yield a high level of

reconfigurability and antenna parameter control. The primary disadvantage of this

method is that the underlying physics of the particular antenna must be known in

order to take advantage of minor geometry modifications to achieve the

reconfigurable goal. Additionally, the amount of reconfigurability is ultimately

limited by the electrical characteristics of the antenna geometry. In this

41

dissertation antenna geometry method and smart geometry reconfiguration are

used to design and implement various reconfigurable antennas.

2.6 CLASSIFICATION OF RECONFIGURABLE ANTENNAS AND

RECONFIGURATION TECHNIQUES [1]

There are various techniques are there to implement reconfigurable antennas, as

indicated in Fig.2.15.Antennas based on electronic switching components to

redirect their surface currents are called electrically reconfigurable. Antennas that

rely on photoconductive switching elements are called optically reconfigurable

antennas. Physically reconfigurable antennas can be achieved by altering the

structure of the antenna. Finally, reconfigurable antennas can be implemented

through the use of smart materials such as ferrites and liquid crystals. In this

dissertation electrical reconfiguration technique is used to design and implement

various reconfigurable antennas.

Figure 2.15 various techniques to achieve reconfigurable antennas

When designing reconfigurable antennas, we must address which reconfigurable

property (e.g., frequency, radiation pattern, polarization or combination of these)

42

needs to be modified. Based on that Reconfigurable antennas can be classified into

four different categories

1. Frequency reconfigurable Antennas

2. Radiation Pattern reconfigurable Antennas

3. Polarization reconfigurable Antennas

4. Compound Reconfigurable Antennas

2.6.1 Frequency Reconfigurable Antennas

Frequency reconfigurable antenna has the reconfiguration of the resonant

frequency by the change of the structure, while the radiation patterns and

polarization remain unchanged. Frequency-reconfigurable antennas (also called

tunable antennas) can be classified into two categories: continuous and switched.

Continuous frequency-tunable antennas allow for smooth transitions within

or between operating bands without jumps.

Switched tunable antennas, on the other hand, use some kind of switching

mechanism to operate at distinct and/or separated frequency bands.

Both kinds of antennas in general share a common theory of operation and

reconfiguration, the main differences are in the extent of the effective length

changes that enable operation over different frequency bands and the devices used

to achieve these changes.

2.6.1.1 Principle behind frequency reconfiguration

Many common antennas such as linear antennas, loop antennas, slot

antennas, and microstrip antennas, are usually operated in resonance. In these

cases, the effective electrical length of the antenna largely determines the

operating frequency, its associated bandwidth (typically no more than about 10%

and usually around 1% to 3% for a single resonance), and the current distribution

on the antenna that dictates its radiation pattern. For instance, for a traditional

43

linear dipole antenna, the first resonance occurs at a frequency where the antenna

is approximately a half wavelength long, and the resulting current distribution

results in an omnidirectional radiation pattern centered on and normal to the

antenna axis. In this case, if one wants the antenna to operate at a higher

frequency, the antenna can simply be shortened to the correct length

corresponding to a half wavelength at the new frequency. The new radiation

pattern will have largely the same characteristics as the first because the current

distribution is the same relative to a wavelength. The same principle holds true for

loops, slots, and microstrip antennas as well.

2.6.1.2 Frequency Reconfiguration Mechanisms

A number of mechanisms can be used to change the effective length of resonant

antennas, although some of these are more effective than others in maintaining the

radiating characteristics of the original configuration. The effective length of the

antenna, and hence its operating frequency, can be changed by adding or removing

part of the antenna length through electronic, mechanical, or other means. There

are different kinds of switching technology FETs and PIN diode in frequency

tunable monopole and dipole antennas for various frequency bands. Although

these are just some of the many examples of switched frequency reconfigurability

with a range of different antenna types and geometries, they all share the common

approach of discrete changes in effective length to achieve their goals.

The use of variable reactive loading has much in common with the switched

reconfigurability discussed above the only real difference between the two is that,

in this case, the change in effective antenna length is achieved with devices like

varactor diodes or mechanisms that can take on a continuous range of values

(typically capacitance) that allows smooth rather than discrete changes in the

antenna’s operating frequency band.

2.6.2 Polarization Reconfigurable Antennas

Antenna polarization reconfiguration can help provide immunity to interfering

signals in varying environments as well as provide an additional degree of

44

freedom to improve link quality as a form of switched antenna diversity. They can

also be used in active read/write tracking/tagging applications.

2.6.2.1 Principle behind Polarization reconfiguration

The direction of current flow on the antenna translates directly into the

polarization of the electric field in the far field of the antenna. To achieve

polarization reconfigurability, the antenna structure and/or feed configuration have

to change in ways that alter the way current flows on the antenna without

significant changes in impedance or frequency characteristics. Polarization

reconfigurations can take place between different kinds of linear polarization,

between right- and left-handed circular polarizations, or between linear, circular

and elliptical polarizations.

2.6.2.2 Polarization Reconfiguration Mechanisms

The mechanisms to achieve polarization reconfigurability are largely the same as

those described for frequency reconfigurability earlier, although their

implementations are necessarily different. By inserting a switch (FET or PIN

diode) into the antenna geometry, by carefully controlling the current direction we

will be able to achieve different polarizations. Polarization reconfigurability is also

achieved by including switches in the feed excitation slots rather than the surface

of the patch.

2.6.3 Radiation Pattern Reconfigurable Antennas

Radiation patterns reconfigurable antenna has the reconfiguration of the radiation

patterns by the change of the structure, while the resonant frequency and

polarization remain unchanged.

2.6.3.1 Principle behind Radiation pattern reconfiguration

The arrangement of currents, either electric or magnetic, on an antenna structure

directly determines the spatial distribution of radiation from the structure. This

relationship between the source currents and the resulting radiation should makes

pattern reconfigurability. To develop antennas with specific reconfigurable

45

radiation patterns, a designer must determine what kinds of source current

distributions, including both magnitude and phase information, are necessary.

Once a topology for the current distribution is determined, a baseline antenna

design can be selected and then altered to achieve the desired source current

distribution. This design process is very much similar to that of array synthesis.

The remaining task is to either arrange the design so that the frequency

characteristics are largely unchanged or to compensate for changes in impedance

with tunable matching circuits at the antenna terminals. In some cases, antenna

types (such as reflector antennas or parasitically coupled antennas) are selected so

that the input is more isolated from the reconfigured portion of the structure,

allowing the frequency characteristics to remain relatively unchanged while

radiation patterns are reconfigured.

2.6.3.2 Radiation Pattern Reconfiguration Mechanisms

One of the most effective and widespread methods to change radiation patterns

independently from frequency behavior is the use of electrically tuned or switched

parasitic elements. Parasitic element lengths are changed with FET or PIN diode

switches or varactors which, in turn, alter the magnitudes and phases of the

currents on the parasitic elements relative to the driven element. Tilts in the main

beam in one plane can then be switched or swept as the lengths of the parasitic

elements are changed. These methods possess several attractive qualities: isolation

of the driven element(s) from the tuned element(s) and a range of available

topologies and functionalities. Fundamentally, tuning of antenna radiation patterns

in this manner relies on the mutual coupling between closely spaced driven and

parasitic elements, resulting in effective array behavior from a single feed point.

Therefore, changes in radiation patterns are achieved through changes in the

coupling between the elements, which, in turn, change the effective source

currents on both the driven and parasitic elements. Relative leads or lags in the

induced current on the parasitic element(s) result in classical “reflector” or

“director” behavior that leads to steered or tilted beams. The effects of radiation

46

from DC bias lines with the driven element relatively isolated from the

reconfigured sections of the structure, the operating frequency and impedance

bandwidth are preserved. This antenna, in a similar manner to most parasitically

tuned antennas, can be analyzed theoretically using a combination of coupling and

array theory to explain pattern tilts.

2.6.4 Compound Reconfigurable Antennas

The ability of the antenna to reconfigure the selective operating frequency,

bandwidth, radiation pattern and polarization characteristics are termed as

compound reconfigurable antenna. The fundamental theory of operation behind

any compound reconfigurable antenna is no different from that of an ordinary

reconfigurable antenna, design and control in this case are obviously more

complex. To date, most single-element compound reconfigurable antennas have a

primary focus on one operating dimension and then additional functionality is

achieved by tuning of the existing structure. However, in the case of

reconfigurable apertures, the frequency and radiation characteristics are

determined together.

2.7 ANTENNA SWITCHING TECHNOLOGIES

This section gives a brief overview of the RF switches available for use in antenna

systems. It includes switches that have been used in both classical antenna systems

and more reconfigurable implementations. In particular it explores conventional

mechanical switches and solid state switches and makes recommendations of

candidates for use in reconfigurable antenna designs.

The fundamental role of a switch or relay is a device to make or break an electric

circuit. In static and quasi-static terminology, a switch operates simply as either a

conduction path or a break in the conduction path. However, switch operation in

an RF system will include additional electrical properties. Switch resistance,

capacitance and inductance along the RF signal path must be included in the

analysis of the system. In RF antenna systems, switch function typically entails

controlling and directing the flow of RF energy along a desired RF path.

47

Traditionally, this path may include any of the RF subsystems leading to the

antenna feed distribution network as well as the antenna feed and, in the case of

arrays, any power distribution network. The introduction of reconfigurable

antennas has also added the antenna itself to the list of places where switches are

utilized to control the direction and flow of RF current. Irrespective of the type of

switch used, there are several important characteristics that must be evaluated for

all RF switch applications and particularly reconfigurable antenna designs.

Similar to electrical switches, RF and microwave switches come in different

configurations providing the flexibility to create complex matrices and automated

test systems for many different applications. Below is a list of typical switch

configurations and usage:

• Single-pole-double-throw (SPDT or 1:2) switches route signals from one

input to two output paths.

• Single-pole-multiple-throw (SPnT) switches allow a single input to

multiple (three or more) output paths.

• Double-pole-double-throw (DPDT) switches can be used to switch

between two inputs and two outputs, as a drop-out switch, for signal

reversal, or to bypass a test component.

The following are the important parameters for proper choice of switch at

microwave frequencies

Frequency range/Bandwidth

The chosen switches should have better performance at operating frequency of the

antenna. The impedance bandwidth should be broader than the antenna bandwidth.

It should have less effect on the antenna performance. The parasitic due to

switching circuitry should be minimal.

Insertion loss

In addition to proper frequency selection, insertion loss is critical. Losses greater

than 1 or 2 dB will attenuate peak signal levels. A low insertion loss system can be

48

achieved by minimizing the number of connectors and through-paths, or by

selecting low insertion loss devices for system configuration.

Return loss

Return loss is caused by impedance mismatch between circuits. At microwave

frequencies, the material properties as well as the dimensions of a network

element play a significant role in determining the impedance match or mismatch

caused by the distributed effect. Switches with excellent return loss performance

ensure optimum power transfer through the switch and the entire network.

Isolation

Isolation is the degree of attenuation from an unwanted signal detected at the port

of interest. Isolation becomes more important at higher frequencies. High isolation

reduces the influence of signals from other channels, sustains the integrity of the

measured signal, and reduces system measurement uncertainties.

Switching speed

Switching speed is defined as the time needed to change the state of a switch port

(arm) from “ON’ to “OFF” or from “OFF” to “ON”.

Settling time

As switching time only specifies an end value of 90% of the settled/final value of

the RF signal, settling time is often highlighted in solid state switch performance

where the need for accuracy and precision is more critical. Settling time is

measured to a level closer to the final value. The widely used margin-to-final

value of settling time is 0.01 dB (99.77% of the final value) and 0.05 dB (98.86%

of the final value).

Power handling

Power handling defines the ability of a switch to handle power and is very

dependent on the design and materials used.

Termination

A 50-ohm load termination is critical in many applications, since each open

unused transmission line has the possibility to resonate. This is important when

designing a antenna for higher frequencies where switch isolation drops

49

considerably. When the switch is connected to an active device, the reflected

power of an unterminated path could possibly alter the characteristics of the

antenna.

Operating life

A long operating life reduces cost per cycle and budgetary constraints allowing

manufacturers to be more competitive.

Types of Switches

RF switches may be mechanical or semiconductor. A switch is an open circuit

when no actuation voltage is applied and a low-impedance path for the RF signal

when an actuation voltage is applied. The switch can be implemented in a series or

Shunt configuration. The Table 2.1 summarizes the performance comparison of

mechanical switches with the solid state switches.

Parameter Mechanical Switch Solid State Switch

Frequency range from [DC] from kHz

Insertion loss Low High

Return loss Good Good

Isolation Good excellent

Switching speed in ms in ns

Settling time < 15 ms < 1 us

Power handling High Low

Operating life 5 million cycles infinite

ESD immunity High Low

Sensitive to Vibration RF power overstress

TABLE 2.1 Performance comparison of Mechanical and Solid state switches

2.7.1 Mechanical Switch

Due to their large size conventional mechanical switches are not practical for

reconfigurable antenna applications. However, they have been used extensively in

high power RF applications. Mechanical switching is normally accomplished by

breaking the conducting path of a transmission line within the switch. The

50

electrical performance of mechanical switches are often much better than those of

solid state switches. Insertion losses of less than 0.1 dB and isolations of over 70

dB are common. Generally they are big in size they might not be compatible with

the printed circuit board. These are capable of handling high power levels.

Because we have moving parts in the switch system their switching speed is

limited by the resonant frequency of the mechanical parts, we will have less

switching speeds, this justifies the use of mechanical switch at lower frequency

and high power handling situations.

2.7.2 Semiconductor Switches

2.7.2.1 PIN diode switch

The PIN diode switch is a popular in microwave circuit applications due to its fast

Switching times and relatively high current handling capabilities. Conventional

electromechanical RF switches are inherently speed limited devices due to inertial

and contact potential effects. The PIN diode can operate at speeds orders of

magnitude faster than mechanical switches and can be placed in packaging

measuring a fraction the size of mechanical RF switches. The PIN diode along

with other solid state switches utilize a semiconductor junction as the RF control

element which accounts for the increase in switch speed and reduction in package

size. Switching speeds of less than 100 ns are typical. An important quality for RF

applications is the fact that it can behave as an almost pure resistance at RF

frequencies. This resistance may be varied over a range of approximately 1 to 10 k

by biasing with a dc or low frequency current. The bias current required for on

state operation is normally on the order of 10 mA, therefore, the power

consumption for overall system will be less when we use PIN diodes.

Structure of PIN diode consists of three regions namely p-doped, n-doped, in

between these layers Intrinsic (un-doped) layer is sandwiched. The structure of

PIN diode is shown Fig.2.16.

51

Fig 2.16 PIN diode cross section and its circuit equivalents

PIN diode shows a large difference between forward bias and reverse bias

states, it shows small impedance while in forward bias and shows large impedance

in reverse bias, this effect has been used for switching purpose. PIN diode

switching circuit almost like filter designing type, a band pass filter with pass band

and stop band. Figures 2.17, 2.18 and 2.19 show the typical PIN diode as SPST &

SPDT switch configurations for both series and shunt orientations.

(a) PIN diode switch circuit (b) Ideal equivalent RF circuit representation

Figure 2.17 Series SPST PIN diode RF Switch model

52

(b) PIN diode switch circuit (b) Ideal equivalent RF circuit representation

Figure 2.18 Shunt SPST PIN diode RF Switch model

(a) Series SPDT switch (b) Shunt SPDT switch

Figure 2.19 PIN diode as SPDT switch

2.7.2. 2 Field Effect Transistor (FET) Switches

FET is a semiconductor device where the output current is controlled with

help of applied field i.e voltage. Normally, PIN diode switch is used as the

predominant switch in antenna switching applications. But recent innovation of

new semiconductor materials like GaAs technology are proved to be more

attractive compared to the other semiconductor switches due to the following

53

reasons. First it provides very good isolation between the RF path and biasing

circuit. Secondly low power consumption as they consume almost zero DC power

while operating. Third important characteristics of the switch is switching speed,

FET switching time is only a few nanoseconds. FET switch is a three terminal

device, the bias voltage at the FET gate terminal provides the switch control.

Switching the gate voltage between zero and greater than the device pinch-off

voltage toggles the switch between it’s on and off states. Figures 2.20 and 2.21

show the typical FET as SPST & SPDT switch configurations for both series and

shunt orientations.

(a) FET switch circuit (b) Ideal equivalent RF circuit representation

Figure 2.20 FET switch configuration (series switch)

(a) switch circuit (b) Ideal equivalent RF circuit representation

Figure 2.21 FET switch configuration (shunt switch)

54

RF FET switches tend to operate over a broader bandwidth than PIN diode

switches but the associated insertions losses for FET switches are comparatively

higher than PIN diode implementations. FET switch tend to suffer from increased

insertion losses at frequencies over 1 GHz and reduced isolation in the off state.

Typically 1-2 dB of insertion loss and 20 to 25 dB of isolation in the off state is

seen at these higher frequencies. The Table 2.2 summarizes the performance

comparison of different solid state switches such as standard FET switches and

PIN diodes.

Parameter PIN FET

Insertion loss at 1 GHz (dB) 0.5-1.0 0.5-1.0

Isolation at 1 GHz (dB) 20-40 40

Switching speed (s) 650X10-9

10-9

-10-8

Bandwidth (MHz) 20-2000 -

Actuation voltage (V) 3-5 5-50

Bias current (µA) 104 <10

TABLE 2.2 Comparison of performance of FET and PIN switches [27]

By comparing overall performance characteristics of solid state switches such as

FET transistors and PIN diodes, it can be concluded that the PIN diode switches

can offer promising characteristics for reconfigurable antennas.

2.8 RECONFIGURABLE ANTENNA CAD SIMULATOR

The tasks of accurately modeling microwave circuits containing switches are

critical for predicting antenna behavior. In the past 15 years, EM field simulators

have become the most common type of tool for designing and simulating antennas

[73]. A wide variety of field solver techniques exist including moment

method(MoM), finite element method (FEM), finite difference time domain

(FDTD), finite integration technique (FIT) and transmission line modeling (TLM).

Though the specific details of each approach are different, all field solvers solve

Maxwell's equations to determine the antenna's electrical properties.

55

Fundamentally, field solvers take a microscopic approach to calculating the

antenna design problem and work by subdividing the problem space into smaller

pieces referred to as cells. Cell size is typically based on many conditions and is

highly dependent on the particular solver formulation used, but in all cases, the

guided wavelength is critical. Several other cell discretization parameters include

direction of field propagation, material properties, and geometrical discontinuities.

Consequently, the numerical size of the total field problem is directly related to the

resolution of the geometry discretization. Field solvers are an excellent choice for

antenna design because it is possible to include all electromagnetic effects from

first principles without having to include compensation terms which are often

required in circuit theory or other model approximations methods. It is often

impractical to model both the PIN device and the antenna in the same simulation.

As a result, the antenna is simulated separately in microwave mode and PIN

diodes are simulated in circuit mode, then, co-simulation will be carried out to

calculate the overall response. The modeling of the PIN diode should be more

accurate to get the optimum result.

All the reconfigurable antenna elements reported in this dissertation has been

simulated using Agilent Advanced System Design (ADS) software package. It has

been carried out in two stages. First, the antenna layout is simulated in ADS

Momentum window, then the simulated file has been used in the schematic

window, i.e co-simulation, for adding the effect of PIN diodes and other bias and

matching circuits, Post-processing has been carried out after S-Parameters

simulation to know the radiation characteristics. This software has the feature of

co-simulation where we can incorporate the effect of biasing circuits and PIN

diodes embedded in the antenna.

2.8.1 Agilent ADS Momentum Formulation [29]

Momentum is based on a numerical discretization technique called the Method of

Moments. This technique is used to solve Maxwell's electromagnetic equations

for planar structures embedded in a multilayered dielectric substrate. Momentum

56

has two modes of operation, microwave or full-wave mode and the RF or quasi-

static mode. The main difference between these two modes lies within the Green

functions formulations that are used. The full-wave mode uses full-wave Green

functions, these are general frequency dependent Green functions that fully

characterize the substrate without making any simplification to the Maxwell

equations. This results in L and C elements that are complex and frequency

dependent. The quasi-static mode uses frequency independent Green functions

resulting in L and C elements that are real and frequency independent. Because of

the approximation made in the quasi-static mode, the RF simulations run a lot

faster since the matrix L and C elements only have to be calculated for the first

frequency simulation point. The approximation also implies that the quasi-static

mode typically should be used for structures that are smaller than half the

wavelength. Both engine modes are using the so-called star-loop basis function,

ensuring a stable solution at all frequencies. Both engines also make use of a mesh

reduction algorithm which reduces the number of unknowns in the simulation by

generating a polygonal mesh. This mesh reduction algorithm can be turned on or

off, with a toggle switch.

The sources applied at the ports of the circuit yield the excitations in the

equivalent network model. The currents in the equivalent network are the

unknown amplitudes of the rooftop expansion functions. Solving the equivalent

network for a number of independent excitation states yields the unknown current

amplitudes. A port calibration process is used to calculate the S-parameter data of

the circuit from the current solution, when calibration is requested.

The following sections contain more information about:

• The method of moments technology

• The Momentum solution process

2.8.1.1 The Method of Moments Technology

The method of moments (MoM) technique is based upon the work of R.F.

Harrington, an electrical engineer who worked extensively on the method and

applied it to electromagnetic field problems, in the beginning of the 1960's. It is

57

based on older theory which uses weighted residuals and variational calculus. In

the method of moments, prior to the discretization, Maxwell's electromagnetic

equations are transformed into integral equations. These follow from the definition

of suitable electric and magnetic Green's functions in the multilayered substrate.

In Momentum, a mixed potential integral equation (MPIE) formulation is used.

This formulation expresses the electric and magnetic field as a combination of a

vector and a scalar potential. The unknowns are the electric and magnetic surface

currents flowing in the planar circuit.

Using notations from linear algebra, we can write the mixed potential integral

equation in very general form as a linear integral operator equation:

³´µ¶#m, m=$. 9#m$ 7#m$ (2.40)

Here, J(r) represents the unknown surface currents and E(r) the known excitation

of the problem. The Green's dyadic of the layered medium acts as the integral

kernel. The unknown surface currents are discretized by meshing the planar

metalization patterns and applying an expansion in a finite number of sub-

sectional basis functions B1(r), ...,BN(r):

J(r) ¸ ∑ ºG»G#m$¼Gr (2.41)

The standard basis functions used in planar EM simulators are the sub sectional

rooftop functions defined over the rectangular, triangular, and polygonal cells in

the mesh. Each rooftop is associated with one edge of the mesh and represents a

current with constant density flowing through that edge as shown in the Fig.2.22.

The unknown amplitudes Ij, j=1,..,N of the basis function expansion determine the

currents flowing through all edges of the mesh.

58

Figure 2.22 Discretization of the surface currents using rooftop basis

functions

The integral Eq.2.40 is discretized by inserting the rooftop expansion Eq. 2.41 of

the currents. By applying the Galerkin testing procedure, that is, by testing the

integral equation using test functions identical to the basis functions, the

continuous integral Eq.2.40 is transformed into a discrete matrix equation for

i=1,...,N

∑ ½¾,GºG ¿¾ ¼¾r or [or [Z] . [I] = [V] (2.42)

With

½¾.G ³ ´»¾#m$.³ ´=µ¶ÀÁ #m, m=$ . »G#m$ À (2.43)

¿¾ ³ ´»¾#m$. 7#m$À (2.44)

The left hand side matrix [Z] is called the interaction matrix, as each element in

this matrix describes the electromagnetic interaction between two rooftop basis

functions. The dimension N of [Z] is equal to the number of basis functions. The

right-hand side vector [V] represents the discretized contribution of the excitations

applied at the ports of the circuit.

The surface currents contribute to the electromagnetic field in the circuit by means

of the Green's dyadic of the layer stack. In the MPIE formulation, this Green's

59

dyadic is decomposed into a contribution from the vector potential A(r) and a

contribution from the scalar potential V(r):

µ¶#m, m=$ µÃ#m, m=$º ¶ ? GH Ä)µÅ#m, m=$Ä=0 (2.45)

The scalar potential originates from the dynamic surface charge distribution

derived from the surface currents and is related to the vector potential through the

Lorentz gauge. By substituting the Eq.2.45 for the Green's dyadic in the Eq.2.43

for the interaction matrix elements, yields the following form:

½¾,G Âƾ,G GHÇÈ,É (2.46)

With

ƾ,G ³ ´»¾#m$.³ ´=µÃ#m, m=$»G#m=$ ÀÁ

À (2.47)

ÇÈ,É ³ ´Ä. »¾#m$³ ´=µÅ#m, m=$Ä. »G#m=$

ÀÁ À (2.48)

This allows the interaction matrix equation to be given a physical interpretation by

constructing an equivalent network model shown in the Fig.2.23 In this network,

the nodes correspond to the cells in the mesh and hold the cell charges. Each cell

corresponds to a capacitor to the ground. All nodes are connected with branches

which carry the current flowing through the edges of the cells. Each branch has in

inductor representing the magnetic self coupling of the associated current basis

function. All capacitors and inductors in the network are complex, frequency

dependent and mutually coupled, as all basis functions interact electrically and

magnetically as shown Fig.2.24. The ground in this equivalent network

corresponds with the potential at the infinite metallization layers taken up in the

layer stack. In the absence of infinite metallization layers, the ground corresponds

with the sphere at infinity. The method of moments interaction matrix equation

follows from applying the Kirchoff voltage laws in the equivalent network. The

currents in the network follow from the solution of the matrix equation and

represent the amplitudes of the basis functions.

60

Figure 2.23 The equivalent circuit is built by replacing each cell in the

mesh with a capacitor to the ground reference and inductors to the

neighboring cells

Figure.2.24 Equivalent network representation of the discretized

MoM problem

61

2.8.1.2 The Momentum Solution Process

The following are the different steps and technologies enable the Momentum

solution process:

• Calculation of the substrate Green's functions

• Meshing of the planar signal layer patterns

• Loading and solving of the MoM interaction matrix equation

• Calibration and de-embedding of the S-parameters

• Reduced Order Modeling by Adaptive Frequency Sampling

2.8.2 PIN Diode Switch Modeling and Simulation in ADS

A considerable amount of effort and research has gone into accurately predicting

the electrical behavior of PIN diode switches. One of the most compelling reasons

for using PIN diode switches in antenna applications are that they have shown to

approximate, to a very good degree, ideal switches. This allows us to forego the

nearly impossible task of a complete field simulation of an entire antenna structure

including all the minute details of the PIN diode switches. Thus, reconfigurable

antenna modeling can be performed using a less complex approach with simplified

equivalent switch circuit in place of the actual PIN diode structures.

Figure 2.25 Comparison of ideal and non-ideal transmission line circuit model

Switches

62

Two options are examined in this dissertation to model reconfigurable antennas in

a simplified manner. The first approach taken is called the reactive switch model

and treats the RF switch as a non-ideal component. Electrically the switch is

modeled with lumped elements. The on-state consists of a small series resistance,

Rs, in the switch transmission line path. Rs is the switch contact resistance and is

normally depends on the actual switch to be used. The off-state is modeled as a

large series resistance, in parallel with a small capacitance. Incorporating the

reactive switch model can be bulky if large numbers of switches are to be used.

Placing a discrete component within the model requires the insertion of a port at

the physical switch location in the geometry definition. The task of adding the

switch properties now becomes solving a circuit based model. The switch

electrical properties are added to the S-parameter data and the final solution to the

problem is obtained during co-simulation after post-processing of the antenna is

completed. This model takes into account the well understood behavior of the

reactive properties of PIN diode switches.

Complete Model Reduced Models

Figure 2.26 Reconfigurable antenna modeling comparison

The second approach will model the PIN diode switches as ideal elements and will

only include the reactive characteristics intrinsic to the discontinuities caused by

the presence of the switch itself. Fig.2.25 shows the two models in terms of

equivalent transmission lines. Fig.2.26 summarizes the methods available for

simulating reconfigurable in the presence of RF switches. The complete model

Full

Electromagnetic

analysis of

antenna with

detailed RF

Switch models

Electromagnetic

analysis of antenna

structure with

circuit model

switches

Electromagnetic

analysis of

antenna structure

with ideal model

switches

Reactive Ideal

63

which performs a full electromagnetic analysis on both the antenna structure and

the RF switch structure is currently considered impractical due to the large

problem size required to model the switches. This leaves the reduced models

which represent the switches in a more manageable manner. The non-ideal model

treats the switches as reactive elements requires additional port structures in the

model. It also requires the additional step of post processing to include switch the

effects within the model and is practical for reconfigurable antennas with only a

small number of switches (less than 10). The ideal model is the only method

currently practical for antennas with a relatively large number (greater than 10) of

switches. There is no post processing needed to include the switch effects and

toggling between switch states is performed simply by adding or removing metal

patches.

2.9 SUMMARY

In this chapter detailed study on reconfigurable methodologies using

microstrip antennas and the design procedures of different microstrip antenna

geometries used in the thesis which form the basis for the reconfigurable antenna

and also modeling of different feed types used in this dissertation. The detailed

description of reconfigurable antenna technology including various methodologies

for achieving frequency, pattern and polarization reconfiguration along with the

principle behind reconfiguration and mechanisms to achieve them. Brief overview

of the RF switches available for use in reconfigurable antenna systems has been

discussed and formulation & technology of the CAD simulator used to design

various antennas in this dissertation.