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1368 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 3, MARCH 2014

Soft-Switching Bidirectional Isolated Full-BridgeConverter With Active and Passive Snubbers

Tsai-Fu Wu, Senior Member, IEEE, Jeng-Gung Yang, Chia-Ling Kuo, and Yung-Chun Wu

Abstract—A bidirectional isolated full-bridge dc–dc converterwith a conversion ratio around nine times, soft start-up, andsoft-switching features for battery charging/discharging is pro-posed in this paper. The converter is equipped with an activeflyback and two passive capacitor–diode snubbers, which canreduce voltage and current spikes and reduce voltage and cur-rent stresses, while it can achieve near zero-voltage-switching andzero-current-switching soft-switching features. In this paper, theoperational principle of the proposed converter is first described,and its analysis and design are then presented. A 1.5-kW prototypewith a low-side voltage of 48 V and a high-side voltage of 360 V hasbeen implemented, from which measured results have verified thediscussed features.

Index Terms—Bidirectional, snubbers, soft switching.

I. INTRODUCTION

IN RENEWABLE dc supply systems, batteries are usu-ally required to back up power for electronic equipment.

The voltage levels of the batteries are typically much lowerthan the dc-bus voltage. Bidirectional converters for charging/discharging the batteries are therefore required. In past studies,bridge-type bidirectional isolated converters have been widelyapplied to fuel cell and electric vehicle driving systems [1]–[6].For raising power level, a dual full-bridge configuration isusually adopted [7]–[18], and their low and high sides aretypically configured with boost- and buck-type topologies,respectively. However, component stress, switching loss, andelectromagnetic interference (EMI) noise are increased dueto diode-reverse-recovery current and MOSFET drain–sourcevoltage, resulting in low reliability. A more severe issue is dueto leakage inductance of the isolation transformer, which willresult in high-voltage spike during switching transition. A pos-sible solution is to pre-excite the leakage inductance to raise itscurrent level up to that of the current-fed inductor, which can re-duce their current difference and, in turn, reduce voltage spike.However, since the current level varies with load condition, it is

Manuscript received February 3, 2012; revised March 28, 2012; acceptedApril 29, 2012. Date of publication May 13, 2013; date of current versionAugust 23, 2013.

T.-F. Wu is with the Elegant Power Electronic Application Research Labo-ratory, Department of Electrical Engineering, National Tsing Hua University,Hsinchu 300, Taiwan (e-mail: [email protected]).

J.-G. Yang and C.-L. Kuo are with the Elegant Power Application ResearchCenter, Department of Electrical Engineering, National Chung Cheng Univer-sity, Minxiong 621, Taiwan (e-mail: [email protected]).

Y.-C. Wu is with the Innovative Design and Energy Application Laboratory,Department of Aeronautical Engineering, National Formosa University, Huwei632, Taiwan (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIE.2013.2262746

Fig. 1. Bidirectional isolated full-bridge dc–dc converter with an active clampsnubber.

Fig. 2. Bidirectional isolated full-bridge dc–dc converter with a flybacksnubber (type A).

hard to tune the switching timing to match these two currents.Thus, a passive or an active snubber circuit is still needed.

Passive and active clamp circuits were proposed to sup-press the voltage spike due to the current difference be-tween the current-fed inductor and leakage inductance currents[13], [14]. A conventional passive approach is employing aresistor–capacitor–diode snubber to clamp the voltage, and theenergy absorbed in the buffer capacitor is dissipated on theresistor, resulting in low efficiency. On the other hand, a simpleactive clamping circuit [13] was proposed, which is shown inFig. 1. However, the resonant current will flow through the mainswitches, increasing current stress significantly. An isolatedbidirectional converter with a flyback snubber was thereforeproposed [18], [19], which is shown in Fig. 2. The flybacksnubber can recycle the absorbed energy which is stored in theclamping capacitor CC , while without current flowing throughthe main switches. It can also clamp the voltage to a desiredvalue just slightly higher than the voltage across the low-side transformer. Since the snubber current does not circulatethrough the main switches, current stress can be reduced a lotunder heavy-load condition. Furthermore, the flyback snubbercan be controlled to precharge the high-side capacitor to avoidin-rush current during a start-up period. However, the low- andhigh-side switches are operated with hard switching turnoff,resulting in high-voltage spikes.

To solve the aforementioned problem, we first introduce twobuffer capacitors (Cb1 and Cb2) connected in parallel with theupper legs of the voltage-fed bridge, as shown in Fig. 3. With

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Fig. 3. Bidirectional isolated full-bridge dc–dc converter with a flybacksnubber and paralleled snubber capacitors Cb1 and Cb2 (type B).

Fig. 4. Proposed soft-switching bidirectional isolated full-bridge converterwith an active flyback and two passive capacitor–diode snubbers (type C).

these two buffer capacitors, the low- and high-side switchescan operate with near zero-voltage switching (ZVS) and zero-current switching (ZCS). However, when it is operated in step-down conversion, these capacitors will resonate with leakageinductance of the transformer, causing EMI noise and increas-ing switching loss. Thus, two passive capacitor–diode snubbersare proposed to supplement the active flyback snubber, asshown in Fig. 4. The proposed snubber configuration cannotonly reduce the voltage spike caused by the current differencebetween the leakage inductance and current-fed inductor cur-rents but also can relieve the drawbacks of high-current andhigh-voltage stresses imposed on the main switches at bothturn-on and turnoff transitions. Moreover, it can achieve nearZVS and ZCS for the switches on both sides of the transformer.

II. CONFIGURATION AND OPERATION

The proposed soft-switching bidirectional isolated full-bridge converter with an active flyback and two passivecapacitor–diode snubbers is shown in Fig. 4. It can be operatedwith two types of conversions: step-up conversion and step-down conversion. Fig. 4 consists of a current-fed switch bridge,an active flyback snubber at the low-voltage side, a voltage-fedswitch bridge, and a passive snubber pair at the high-voltageside. Inductor Lm performs output filtering when power flowsfrom the high-voltage side to the low-voltage side, which isdenoted as a step-down conversion. On the other hand, it worksin the step-up conversion. Moreover, snubber capacitor CC anddiode DC are used to absorb the current difference betweencurrent-fed inductor current iL and leakage inductance currentiP of isolation transformer TP during switching commutation.The flyback snubber is operated to transfer the energy storedin snubber capacitor CC to buffer capacitors Cb1 and Cb2,and voltage VC can drop to zero. Thus, the voltage stresses ofswitches M1 ∼ M4 can be limited to a lower level, achievingnear ZCS turnoff. The main merits of the proposed snubberinclude no spike current circulating through the switches andachieving soft-switching features. Note that high spike current

can result in charge migration, over current density, and extramagnetic force which will deteriorate in MOSFET carrier den-sity, channel width, and wire bonding and, in turn, increase itsconduction resistance.

In the step-up conversion, switches M1 ∼ M4 are controlled,and the body diodes of switches M5 ∼ M8 serve as a rec-tifier. In the step-down conversion, switches M5 ∼ M8 arecontrolled, and the body diodes of switches M1 ∼ M4 operateas a full-bridge rectifier. To simplify the steady-state analysis,several assumptions are made as follows.

1) All components are ideal except that the transformer isassociated with leakage inductance.

2) Inductor Lm is large enough to keep the current iLconstant over a switching period.

3) Snubber capacitor CC is much larger than the parasiticcapacitance of switches M1 ∼ M8.

A. Step-Up Conversion

In the step-up conversion, switches M1 ∼ M4 are oper-ated like a boost converter, where switch pairs (M1,M2) and(M3,M4) conduct to store energy in Lm. At the high-voltageside, body diodes D5 ∼ D8 of switches M5 ∼ M8 will conductto transfer power to CHV. When switch pairs (M1,M2) and(M3,M4) are switched to (M1,M4) or (M2,M3), currentdifference iC (= iL − iP ) will charge capacitor CC until iPrises up to iL, and capacitor voltage VC will be clamped toVHV · (NP /NS), achieving near ZCS turnoff for M2 or M4.In the meantime, high-side current iS has the priority flowingthrough one of the two passive capacitor–diode snubbers, andeither Cb1 or Cb2 will be fully discharged before diode D5

or D7 conducts. When switch pair (M1,M4) or (M2,M3) isswitched back to (M1,M2) and (M3,M4), switch M2 or M4

can have near ZCS turn-on feature due to leakage inductanceLll limiting the di/dt of high-side diode-reverse-recovery cur-rent. The flyback snubber operates simultaneously to dischargesnubber capacitor CC and transfer the stored energy to buffercapacitors Cb1 and Cb2. With the flyback snubber, the energyabsorbed in CC will not flow through switches M1 ∼ M4,which can reduce their current stresses dramatically when theleakage inductance of the isolation transformer is significant.

The key voltage and current waveforms of the converteroperated in the step-up conversion are shown in Fig. 5. Adetailed description of the converter operation over a half-switching cycle is presented as follows.

1) Mode 1 [t0 ≤ t < t1]: Before t0, all of the four switchesM1 ∼ M4 are turned on. Inductor Lm is charged by VLV.At t0, M1 and M4 remain conducting, while M2 and M3

are turned off. Then, clamping diode DC conducts, andsnubber capacitor CC is charged by the current differenceiC . In this mode, the flyback snubber still stays in the OFF

state. The equivalent circuit is shown in Fig. 6(a).2) Mode 2 [t1 ≤ t < t2]: In this mode, leakage inductance

current iP will start to track current iL, and buffer capac-itor Cb1 will start to release energy. At time t2, currentiP is equal to current iL, the voltage of switches M2

and M3 and capacitor CC will reach the maximum valuesimultaneously, and its equivalent circuit is shown in



Fig. 5. Key voltage and current waveforms of the proposed converter operatedin the step-up conversion.

Fig. 6(b). A near ZCS soft switching is therefore attainedduring t0 to t2.

3) Mode 3 [t2 ≤ t < t3]: Before t3, the energy stored inbuffer capacitor Cb1 is not fully discharged yet. Thus, thecapacitor will not stop discharging until Vb1 drops to zero.The equivalent circuit is shown in Fig. 6(c).

4) Mode 4 [t3 ≤ t < t4]: When the energy stored in Cb1 hasbeen completely released to the output at t3, diode D5

will conduct. The circuit operation over this time intervalis identical to a regular turnoff state of a conventionalcurrent-fed full-bridge converter. The equivalent circuitis shown in Fig. 6(d).

5) Mode 5 [t4 ≤ t < t5]: At t4, all of the four switchesM1 ∼ M4 are turned on again, and switch MS of theflyback snubber is turned on synchronously. Switches M2

and M3 achieve a ZCS turn-on soft-switching feature dueto Lll, and current iP drops to zero gradually. In theflyback snubber, the energy stored in capacitor CC willbe delivered to the magnetizing inductance of transformerTS . The equivalent circuit is shown in Fig. 6(e).

6) Mode 6 [t5 ≤ t < t6]: When switch MS is turned off att5, capacitor voltage VC drops to zero, and the energystored in the magnetizing inductance will be transferredto buffer capacitor Cb1. In this mode, the time intervalof driving signal Vgs(Ms) is slightly longer than thedischarging time of capacitor CC . The purpose is toensure that the energy stored in capacitor CC can be com-pletely released, creating a ZCS operational opportunityfor switch M2 or M4 at the next turnoff transition. Theequivalent circuit is shown in Fig. 6(f).

7) Mode 7 [t6 ≤ t < t7]: At t6, the energy stored in the mag-netizing inductance of transformer TS was completelytransferred to buffer capacitor Cb1, and the circuit opera-tion is identical to a regular turn-on state of a conventional

Fig. 6. Operation modes of the step-up conversion. (a) Mode 1. (b) Mode 2.(c) Mode 3. (d) Mode 4. (e) Mode 5. (f) Mode 6. (g) Mode 7.

current-fed converter. Its equivalent circuit is shown inFig. 6(g). The circuit operation stops at t7 and completesa half-switching cycle.



Fig. 7. Key voltage and current waveforms of the proposed converter operatedin the step-down conversion.

B. Step-Down Conversion

In the analysis, the leakage inductance of the transformerat the low-voltage side is reflected to the high-voltage side, inwhich equivalent inductance L∗

eq equals (Llh + Lll ·N2s /N

2p ).

In the step-down conversion, switches M5 ∼ M8 are oper-ated like a buck converter in which switch pairs (M5,M8) and(M6,M7) take turns conducting to transfer power from capac-itor CHV to battery BLV. For alleviating leakage inductanceeffect on voltage spike, switches M5 ∼ M8 are operated withphase-shift control, achieving ZVS turn-on features. Althoughthere is no need to absorb the current difference between iLand iP , capacitor CC can help clamp the voltage ringing due toL∗eq and the parasitic capacitance of M1 ∼ M4. With the two

passive capacitor–diode snubbers, switches M6 and M8 canachieve near ZCS turnoff.

The key voltage and current waveforms of the converteroperated in the step-down conversion are shown in Fig. 7. Adetailed description of its operation over a half-switching cycleis presented as follows.

1) Mode 1 [t0 ≤ t < t1]: In this mode, switches M5 and M8

are turned on, while M6 and M7 are in the OFF state. Thehigh-side voltage VHV is crossing the transformer, andit is, in fact, crossing the equivalent inductance L∗

eq anddrives current iS to rise with the slope of VHV/L

∗eq. With

the transformer current increasing toward the load-currentlevel at t1, the body diodes (D1 and D4) are conductingto transfer power, and the voltage across the transformerterminals on the low-voltage side changes immediatelyto reflect the voltage from the high-voltage side. Theequivalent circuit is shown in Fig. 8(a).

2) Mode 2 [t1 ≤ t < t2]: At t1, switch M8 remains conduct-ing, while M5 is turned off. The body diode of M6 thenstarts conducting the freewheeling leakage current. Thetransformer current iS reaches the load-current level at t1,and VAB rises to the reflected voltage (VHV ·NP /NS).Clamping diode DC starts conducting the resonant cur-rent of L∗

eq and the parasitic capacitance of M1 ∼ M4. At

Fig. 8. Operation modes of the step-down conversion. (a) Mode 1. (b) Mode 2.(c) Mode 3. (d) Mode 4. (e) Mode 5.

the same time, switch MS of the flyback snubber is turnedon and starts transferring the energy stored in capacitorCC to buffer capacitors Cb1 and Cb2. The process ends att2 when the resonance goes through a half resonant cycleand is blocked by clamping diode DC . With the flybacksnubber, the voltage of capacitor CC will be clamped toa desired level just slightly higher than the voltage ofVds(M4). The equivalent circuit is shown in Fig. 8(b).

3) Mode 3 [t2 ≤ t < t3]: At t2, the body diode of switch M6

is conducting, and switch M6 can be turned on with ZVS.The equivalent circuit is shown in Fig. 8(c).

4) Mode 4 [t3 ≤ t < t4]: At t3, switch M6 remains con-ducting, while M8 is turned off. Buffer capacitor Cb2

is discharging by the freewheeling current. When Cb2 isfully discharged, a near ZCS turnoff condition is thereforeattained, and the body diode of M7 then starts conductingthe freewheeling current. The equivalent circuit is shownin Fig. 8(d).



5) Mode 5 [t4 ≤ t < t5]: At t4, with the body diode ofswitch M7 conducting, M7 can be turned on with ZVS.Over this time interval, the active switches change tothe other pair of switches, and the voltage across thetransformer reverses its polarity. The circuit operationstops at t5 and completes a half-switching cycle. Theequivalent circuit is shown in Fig. 8(e).

III. DESIGN AND PRACTICAL CONSIDERATION

OF SNUBBERS

The purpose of using an active flyback snubber is to transferenergy from snubber capacitor CC to buffer capacitors Cb1 andCb2, which can attain a near ZCS soft-switching feature. Toreduce high-voltage spike occurring on switch MS , the flybacksnubber is operated in discontinuous conduction mode, andthe key components of the proposed snubber are designed asfollows.

A. Snubber Capacitor CC

For clamping the switch voltage at the low-voltage side,snubber capacitor CC needs to satisfy the following inequality:

CC ≥ Leq · (iL · ip)2V 2C

(1)

where Leq = Lll + Llh.N2p/N

2s .

B. Leakage Inductance

When the proposed converter is operated under step-downconversion, diode reverse recovery will cause high-currentspike during switching transition. The leakage inductance of thetransformer can not only limit diode-reverse-recovery currentbut also can achieve ZVS turn-on with a phase-shift operationmanner. The leakage inductance needs to satisfy

iS >

√2

Leqh

(4

3CMOSV 2

in +1

2CTRV 2

in

)(2)

where Leqh = Llh + Lll ·N2s /N

2p . However, large leakage in-

ductance will cause high current difference during the step-upconversion, and the flyback snubber needs to process a highpower level.

C. Flyback Snubber

During the interval of [t0, t2], a high transient voltage occursunder the step-up conversion due to the current difference be-tween the leakage inductance and current-fed inductor currents,which can be suppressed by DC and CC . The energy stored incapacitor CC is transferred to buffer capacitors Cb1 and Cb2 bythe flyback snubber. The power rating of the flyback snubbercan be expressed as

PFB = 0.5CC · V 2C · fS (3)

where fS is the switching frequency.

D. Magnetizing Inductance of Flyback Snubber

To ensure that snubber capacitor CC can be fully dischargedby the flyback snubber, the turn-on time of MS (TON) cannotbe longer than 1/4 resonant cycle of the magnetizing inductanceand CC . Thus, the magnetizing inductance of the flybacksnubber should satisfy the following inequality:

Lmf >4T 2

ON

π2CC(4)

where TON is the conduction time of switches M1 ∼ M4 in thestep-up conversion.

E. Buffer Capacitors Cb1 and Cb2

When the converter is operated in the step-down conversion,capacitors Cb1 and Cb2 can share current iS , and Vds(M6) andVds(M8) will rise up with a lower slope at switches M6 and M8

turnoff transition, reducing switching loss. If the power ratingbecomes higher, buffer capacitors Cb1 and Cb2 can be enlargedto achieve near ZCS turnoff. On the other hand, when the con-verter is operated in the step-up conversion, snubber capacitorCC will be fully discharged by the proposed operation scheme.When switches M1 ∼ M4 are switched to either (M1,M4) or(M2,M3) conducting, current difference iC (= iL − iP ) willfirst charge capacitor CC . Meanwhile, current iP can start torise up because of Cb1 and Cb2 holding voltage VHV, whichcan reduce duty loss. For Cb1 and Cb2 being charged up toVHV by the flyback snubber, they need to satisfy the followinginequality:

Cb1, Cb2 ≤ CC · V 2C

V 2HV

. (5)

F. Snubber Diodes Db1 and Db2

In the analysis, if capacitors Cb1 and Cb2 are connected inparallel with the upper legs of the voltage-fed bridge directly,they will resonate with the leakage inductance during switchingtransition in the step-down conversion, increasing switchingloss. Thus, two snubber diodes Db1 and Db2 are introducedto connect with Cb1 and Cb2 in series, respectively, and theringing current through the high-side switches can be blockedeffectively.

G. Soft Start-Up

High inrush current is a start-up problem with the step-upconversion. The initial high-side voltage VHV should not belower than VLV ·NS/NP to avoid inrush current. When VHV

is not high enough, the controller will drive the flyback snubberto precharge high-side capacitor CHV to achieve a soft start-upfeature.

IV. EXPERIMENTAL RESULTS

To verify the operational principle and performance of theproposed converter, three experimental prototypes of 1.5 kW,the converter shown in Fig. 2 (type A), the one shown in Fig. 3(type B), and the proposed, shown in Fig. 4 (type C), weredesigned and built. The low-side voltage is 42–54 V, and the



TABLE ISPECIFICATIONS OF PROTOTYPE

Fig. 9. Photograph of proposed converter (type C).

high-side voltage is 360 V. Type C was implemented with thespecifications listed in Table I, and a photograph of the 1.5-kWexperimental prototype of the proposed converter (type C) isshown in Fig. 9.

In the following discussion, type A will be first com-pared with the proposed converter to verify that a turnoffsoft-switching feature can be achieved in both step-up andstep-down conversions. The voltage and current waveformsmeasured from type B and the proposed one prove that twosnubber diodes Db1 and Db2 are needed to block the ringingcurrent. Then, the current waveforms iP measured from theconverter with an active clamp circuit [13] and the proposed oneshow that the energy stored in CC is recycled by the proposedsnubbers and its discharging current does not flow through themain switches.

Measured waveforms of low-side voltage VLV and current iPunder step-up conversion are shown in Fig. 10. For the voltagevariation of batteries, the proposed bidirectional converter cancover input voltage range from 42 to 54 V.

Measured voltage waveforms of snubber capacitor VC andVds(M4) from type A and the proposed one under step-upconversion are shown in Fig. 11. It can be observed fromFig. 11(a) that VC is regulated to a desired value just slightlyhigher than Vds(M4) in type A. Since the regulated voltage VC

is slightly higher than Vds(M4), the energy transferred by theflyback snubber is just a small amount. However, the parasitic

Fig. 10. Measured waveforms of voltage VLV and current iP from inputvoltages (a) 42, (b) 48, and (c) 54 V under step-up conversion.

Fig. 11. Measured voltage waveforms of VC and Vds(M4) from (a) type A,and (b) the proposed one of which VC is discharged completely in eachswitching cycle, under step-up conversion and with 1.5-kW power rating.

Fig. 12. Measured current iP waveforms from (a) type A and (b) the proposedone under step-up conversion and with 1.5-kW power rating.

capacitance of the switches and the stray inductance of the cir-cuit will result in high-voltage spike. In the proposed converter,capacitor CC is fully discharged, which could result in higherenergy transferred by the flyback snubber than type A. Thus,capacitor CC should be chosen with a smaller capacitance. Fortype A, the value of CC is 1 μF, while that of the proposedone is 0.1 μF. Referring to (2), the processed power PFB by theflyback snubber of the proposed one is 36 W, just around 2.4%of the maximum power level. Moreover, the proposed operatingscheme can achieve near soft-switching feature for switchesM1 ∼ M4 at turnoff transition, and the overshoot voltage canbe suppressed to a lower level, as shown in Fig. 11(b). Fig. 12shows the measured waveform of current iP from type A andthe proposed one. Although capacitor CC is fully discharged,the proposed passive snubber can hold voltage Vb1 or Vb2.



Fig. 13. Measured voltage Vds(M4) and current Ids(M4) waveforms at M4

turnoff transition from (a) type A and (b) the proposed one with 1.5 kW and (c)the proposed one with 500 W, under step-up conversion.

Referring to the following equations, diP /dt of the proposedone is higher than that of type A, which can reduce duty loss

dipdt

=VAB − VCD × NP

NS

Leq

=VCc − (VHV × NP

NS)

Leq(Type A) (6)

dipdt

=VAB − VCD × NP

NS

Leq

=VCc − (VHV − VCb1)× NP

NS

Leq(proposed). (7)

Fig. 13 shows the measured waveforms of voltage Vds(M4)

and current Ids(M4) from type A and the proposed one atM4 turnoff transition, in which Fig. 13(a) and (b) are witha load of 1.5 kW and Fig. 13(c) is with 500 W. It can beobserved that the voltage spike in type A is up to 197 V, dueto the parasitic capacitance of switches M1 ∼ M4 and strayinductance on the circuit. On the other hand, the proposed onecan not only achieve near ZCS turnoff soft-switching featurebut also can alleviate the voltage spike to 107 V, as shown inFig. 13(b). Moreover, Fig. 13(b) and (c) shows that the soft-switching feature can be achieved at both light- and heavy-loadconditions. Fig. 14 shows the measured waveforms of voltageVds(M8) and current Ids(M8) from type A and the proposed oneat M8 turnoff transition. It can be observed that, with Cb1 andCb2, Vds(M8) of the proposed converter rises up with a lowerslope and the switching loss of (Vds(M8) · Ids(M8)) becomeslower, achieving near ZCS turnoff soft-switching feature.

Fig. 15 shows the measured waveforms of voltages Vds(M5)

and Vds(M6) from type B and the proposed one under step-down conversion. It can be observed that, due to large buffercapacitors Cb1 and Cb2, voltage Vds of type B is ringing atswitching transition, as shown in Fig. 15(a). Fig. 16 shows themeasured waveforms of voltages Vds(M5) and Vb1 and currentIds(M5) from type B and the proposed one under step-down

Fig. 14. Measured voltage Vds(M8) and current Ids(M8) waveforms at M8

turnoff transition from (a) type A and (b) the proposed one under step-downconversion and with 1.5-kW power rating.

Fig. 15. Measured voltage Vds(M5) and voltage Vds(M6) waveforms from(a) type B and (b) the proposed one under step-up conversion and with 1.5-kWpower rating.

Fig. 16. Measured voltages Vds(M5) and Vb1 and current Ids(M5) wave-forms from (a) type B and (b) the proposed one under step-down conversionand with 1.5-kW power rating.

conversion. It can be observed from Fig. 16(a) that, because oflarge buffer capacitors Cb1 and Cb2, there exists high ringingcurrent in type B at switching transition, which will result inhigh EMI noise and switching loss. On the other hand, theringing current is much lower in the proposed converter, asshown in Fig. 16(b).

Fig. 17 shows the measured waveform of current iP understep-up conversion from the converter with an active clampcircuit [13] and the proposed one. It can be observed fromFig. 17(a) that the peak current of iP under the full-loadcondition is 48.1 A, which means the discharging current(48.1− 30 = 18.1 A) of CC flowing through the main switchesand increasing stress significantly. On the contrary, the energystored in CC is recycled by the proposed passive and activesnubbers, which can reduce the current stress.

Fig. 18 shows the plots of conversion efficiency versus powerlevel of type A and the proposed one operated in the step-upconversion. The processed power PFB by the flyback snubber inthe proposed converter is 36 W, while PFB is 52.5 W in type A.The conversion efficiency of the proposed converter under thefull-load condition is about 91.5%. However, since capacitor



Fig. 17. Measured current iP waveforms from (a) the converter with an activeclamp circuit [13] and (b) the proposed one under step-up conversion and with1.5-kW power rating.

Fig. 18. Plots of conversion efficiency of type A and the proposed oneoperated in the step-up conversion.

Fig. 19. Plots of conversion efficiency of type A and the proposed oneoperated in the step-down conversion.

TABLE IIKEY COMPONENT LOSS ESTIMATION UNDER 1.5-kW POWER RATING

CC is fully discharged by the proposed operation scheme,power loss of the flyback snubber results in low efficiency underlight-load condition, and the flyback snubber is controlled likethat in type A when the load is lower than 500 W. Notethat, under light-load condition, there is no voltage spike sincecapacitor CC can absorb the current difference between theleakage inductance and the inductor currents. Fig. 19 showsthe plots of conversion efficiency versus power level of type Aand the proposed one operated in the step-down conversion. Itcan be observed that the conversion efficiency of the proposedconverter is close to that of type A since both of them areoperated in similar manner. The key component loss estimationis summarized in Table II.

V. CONCLUSION

This paper has presented a soft-switching bidirectional iso-lated full-bridge converter, which allows input voltage variationfrom 42 to 54 V, for battery charging/discharging applications.The proposed converter can reduce the voltage spike caused bythe current difference between leakage inductance and current-fed inductor currents, the current spike due to diode reverserecovery, and the current and voltage stresses, while it canachieve near ZVS and ZCS soft-switching features. The passivesnubber can hold voltage Vb1 or Vb2 and improve the slew rateof diP /dt, which can reduce duty loss. However, near ZVSturn-on transition cannot be achieved under light-load conditionin step-down conversion. Experimental results measured fromthe three types of 1.5-kW isolated bidirectional full-bridgedc–dc converters have verified that the proposed converter(type C) can yield the performance of lower voltage and currentspikes, higher efficiency, and less ringing. It is suitable for high-power applications with galvanic isolation.

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Tsai-Fu Wu (S’88–M’91–SM’98) received the B.S.degree in electronic engineering from NationalChiao Tung University, Hsinchu, Taiwan, in 1983,the M.S. degree in electrical and computer engi-neering from Ohio University, Athens, OH, USA, in1988, and the Ph.D. degree in electrical engineeringand computer science from the University of Illinois,Chicago, IL, USA, in 1992.

From 1985 to 1986, he was a System Engi-neer with SAMPO, Inc., Kuei-Shan Hsiang, Taiwan,where he was involved in developing and designing

graphic terminals. From 1988 to 1992, he was a Teaching and ResearchAssistant with the Department of Electrical Engineering and Computer Science,University of Illinois, Chicago, IL, USA. From 1993 to 2012, he was withthe Department of Electrical Engineering, National Chung Cheng University,Minxiong, Taiwan. He is currently a Professor with the Department of ElectricalEngineering, National Tsing Hua University, Hsinchu. His current researchinterests include developing and modeling of power converters; design of elec-tronic dimming ballasts for fluorescent lamps, metal halide lamps and plasmadisplay panels; design of solar-array supplied inverters for grid connection; anddesign of pulsed-electrical-field generators for transdermal drug delivery andfood pasteurization.

Dr. Wu is a Senior Member of the Chinese Institute of Engineers. He was therecipient of three Best Paper Awards from the Taipei Power Electronics Asso-ciation in 2003–2005. In 2006, he was named as an Outstanding Researcher bythe National Science Council, Taiwan.

Jeng-Gung Yang was born in Taiwan in 1985. Hereceived the B.S. and M.S. degrees in electricalengineering from National Chung Cheng University,Minxiong, Taiwan, in 2009 and 2011, respectively,where he is currently working toward the Ph.D.degree in the Department of Electrical Engineering.

His current research interests include the designand development of soft-switching power converters.

Chia-Ling Kuo was born in Taiwan in 1985. Shereceived the B.S. and M.S. degrees in electricalengineering from National Chung Cheng University,Minxiong, Taiwan, in 2008 and 2010, respectively,where she is currently working toward the Ph.D.degree in the Department of Electrical Engineering.

Her research interests include the design andimplementation of bidirectional inverters for dc-distribution applications.

Yung-Chun Wu was born in Kaohsiung, Taiwan,in 1971. He received the Ph.D. degree in electricalengineering from National Chung Cheng University,Minxiong, Taiwan, in 2000.

From 2000 to 2001, he was an Assistant Pro-fessor with the Department of Electrical Engineer-ing, Chienkuo Institute of Technology, Changhua,Taiwan. From 2001 to 2007, he was an AssistantProfessor with the Department of Electrical En-gineering, I-Shou University, Kaohsiung, Taiwan.Since 2007, he has been with the Department of

Aeronautical Engineering, National Formosa University, Huwei, Taiwan, wherehe is currently an Assistant Professor. His research interests include developingand designing converter topologies, electronic ballasts, and high-efficiencyconverters.


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