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Receiver DesignTutorial
James B. Offner(Author)
Harris Corporation
Government Communications Systems Division
2400 Palm Bay Rd
Palm Bay, Florida 32905
AbstractNumerous interrelated trade-offs are undertaken for
any receiver or receive chain design, which must be jointly
optimized for the intended operational environment. Some of
the requirements resulting from this environment are: noise
figure (NF), input 3rd order intercept point (IP3), input 1dB
compression point (P1dB), dynamic range, input desensitization
level, non-damage input power, out-of-band (OOB)
interference rejection, gain and output power. This paper
focuses on these requirements and the subtleties associated
with achieving them.
Keywords-receiver; receive chain; frequency plan; spur
analysis; spur; cascade analysis; Out-of-band; interference;
noise source; AGC; ALC; Monte-Carlo; intermodulation
I. INTRODUCTION
Receive (Rx) chain design is grouped into six key areaslisted below, which are then expanded and treated morefully.
1. Frequency planning / spurious (spur) analysis2. Cascade analysis of device Gains, NFs, IPs, P1dBs
and damage levels3. Non-standard noise sources other than from cascade
(wideband amplifier, image, LO, and reciprocal
mixing)4. Out-of-band interference rejection5. Automatic Gain or level Control (AGC, ALC)6. Statistical parameter variation, gain alignment and
compensationMany of these topics are interrelated, where optimization
of one area often negatively impacts one or more of the
others. An optimized receiver design globally optimizes all
six key design areas with equally weighted margins across
all parameters.
II. FREQUENCY PLANNING /SPURANALYSIS
One of the first things to do for any Rx chain design is to
develop a frequency plan and perform a spur analysis (SA)on that plan. The frequency planning process determines theRF, IF, and LO frequency ranges. Usually the RF is givenfor a receiver, but sub-bands of RF may be more palatable todeal with by using a switched filter bank. Without the aid ofa good frequency planning tool, this can be a timeconsuming, bring me a rock exercise, where the optimumplan may not be discovered or worse, the complexity, costand performance are subpar. All Rx chains are susceptible toin-band and OOB spurious responses, which must be
managed to provide robust performance in a receiverintended for hostile signal and interference environments.
Filter quantity and complexity, as well as gain and phaselinearity, are directly impacted by the frequency plan. The
number of filters in a RF switched filter bank for a widebandreceiver can be influenced by the IF selection, and an IFswitched filter bank (selectable IFs after the first conversionstage) can reduce the RF filter quantity. Also, multipleconversion stages may reduce the total filter count and areoften necessary to meet spur requirements for widefrequency ranges and large ratios of RF to final IF.
A frequency planning tool facilitates finding an IF that isspur free or exhibits the lowest spur levels possible, giventhe input signal and interferer levels, selected mixer spurresponses, and required bandwidths (BW) or tuning range ateach mixer port. More than one IF may be usable for a givenconversion stage and the best one will optimize all of theabove parameters as a group. Once a frequency plan is
chosen, it can be further refined by modeling actual RF andIF filter responses and performing a SA given the RF, LOand IF frequency ranges developed using the frequencyplanning tool. The SA takes into account the filters rejectionof OOB input levels, which can significantly improve theresulting output spur levels that are caused from these OOBinput frequencies. The process of frequency planning andevaluating the plan via SA can be iterative, where thefrequency plan may need updating based on SA results.
Down converters (DC) are either non-inverting (NIDC)using a low-side LO (1x-1), or inverting (IDC) using a high-side LO (-1x1), where 1x1 represents the MxN mixingproduct of M x Fin plus N x FLO at the IF output. An IDCoften times yields better spur performance but at the price of
a higher frequency LO and greater LO phase noise. Primaryspurs to manage for either DC are those with M=-N, whichincludes the image response. The image response is removedby filtering or using an image reject mixer or both.
Spur management consists of four primary controls:
Signal powerat the mixer input: determined by gaindistribution and required NF
OOB power at the mixer input: determined byfiltering and influenced by frequency plan
LO power: higher levels (within limits of chosenmixer) raises the mixers input IP, and hence, lowersspur levels, however, it does not increase P1dBsignificantly [1].
Mixer type: class I, II and III (+7 dBm, +17 dBm and+27 dBm nominal LO drive levels) for low, mediumand high mixer input IP
The measured power in a given spur will vary as
(P[MxN])dB = (PRF)dB x |M|, (1)
where PRF is the change in input RF power of the signal
producing the spur [2] [3]. At higher input powers (positive
978-1-61284-080-2/11/$26.00 2011 IEEE
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PRF) the spur level increases. The ratio of spur power to
input RF power varies as
(R[MxN])dB = (PRF)dB x (|M|1). (2)
Additionally, [2] states that for doubly balanced diode
mixers, the LO drive level can also be taken into account to
predict changes in spur levels from measured values as
(R[MxN])dB = (|M|1) [PRF - PLO], (3)
where PLO is the change in LO power. Spur levels decrease
for lower input powers (negative PRF) and higher LO
power (positive PLO).
Single vs. dual or triple conversion depends on the BW
required at each mixer port: a tracking filter or switched
filter bank may allow a dual conversion, where triple would
otherwise be necessary. The choice of an IF frequency may
be dictated by component capability (i.e., performance and
cost). A trade-off must be made between continuous tuning
with increased intermod (IM) levels for tracking filters vs.fixed-tuned filter quantity in a filter bank when dealing with
high power inputs.
Some final notes on frequency conversion:
Harmonics of the LO should be filtered, otherwise aSA for each LO harmonic as the actual LOfrequency should be performed with the desiredinput and output frequency ranges (this type of SAshould also be done when using sub-harmonicmixers).
Assess the MxN mixer spurs resulting in a dual orhigher conversion from LO#1 leakage through thefirst conversion stage and mixing with LO#2 in the
second mixer (a 1x1 CW product could exist,which requires significant IF filtering if notdiscovered during the frequency planning stage).
For multi-stage conversions, assess spurs that areOOB at the 1st stage IF but higher than the in-bandrequirement. These spurs may become in-band at thefinal output IF. For example: two relatively poorspurs, a -1x2 produced in the 1st conversion stageand a -1x3 in the 2nd stage, can combine to becomein-band at the final IF output.
When de-hopping a spread waveform, better spurperformance is usually obtained by de-hopping withthe highest frequency LO (smallest percentage BW)
III. CASCADE ANALYSIS.
When beginning a new receiver/Rx chain design, a
rough cascade analysis is usually done first, which is
followed by the other key design areas, the exact order
dictated by overall requirements. Once a top level gain/loss
(G/L) budget has been done, with the resulting gain
distribution satisfying the basic in-band NF, IP,
compression (P1dB), and non-damage requirements, then
approximate levels will be known at the mixers. With these
levels in hand, a frequency plan/spur analysis can be
performed or updated with more accurately predicted spur
levels. As the design progresses, more detail is added to the
cascade analysis, such as actual part values, gain variation
due to device tolerance and gain versus frequency and
temperature (see the final section below for an expanded
discussion on parameter variation, alignment and
compensation). Space limitations prohibit deriving and
detailing the equations used to obtain overall performance
values for a string of cascaded devices which make up a
subsystem. However, most of these equations are readily
available in the literature and on web sites [4].
Some commercially available analysis programs
estimate the overall P1dB of a subsystem by approximating
the Pout vs Pin compression curve using individual device
PSAT and P1dB values. P1dB of a subsystem is not a fixed or
typical number of dBs below its IP3 as is often used in
approximations for an individual device, the cascading
mechanism and equations being different for the two
parameters. Input IP3 degrades two to three times faster
than does P1dB (in dBm) with additional devices of equal
contribution. The relative softness or hardness of thecompression curve depends on = PSAT P1dB of the
device. s greater than 3 dB yield soft curves and are
indicative of low power solid state devices and high power
traveling wave tubes (TWT). High power solid state PAs
(SSPA), linearized TWTs, and mixers have s on the order
of 1 dB, representing a hard curve. A piece-wise linear
curve is approached as tends to 0 dB , where device gain is
constant below an input power of PSATGain (small signal)
and output power is constant above. This is not realistic for
any device and care must be exercised when using a
program which models compression using the method. If
the value is set arbitrarily too low or a default value near 0
dB is used, the resulting prediction for P 1dB will be too highand possibly not discovered until test.
Power levels at which damage occurs throughout the Rx
chain should be determined using the compression
characteristics of each device and not their linear gains. For
specified high level, non-damage inputs an input limiter
may be necessary to protect the front end. However, even
though the front end is protected, its saturated output level
may not protect downstream components, and a lower
power downstream limiter may also be required to protect
components from high saturation levels of preceding stages.
When assessing damage levels throughout the Rx chain,
keep in mind that the PSAT and P1dB values used in the
analysis for the basic G/L distribution are worst-case (lowerbound) values and a higher bound set of values is needed for
the non-damage assessment. When a device is guaranteed to
provide minimum values, it will by definition exceed those
values most of the time. As a result, when using the min
values, maintain at least 2 or 3 dB of damage margin.
Receivers typically work over a large range of input
powers and often receive simultaneous in-band signals,
some of which are at the very bottom of the power range
while at the same time others exist at the upper end. This
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scenario stresses the receivers sensitivity and power
handling capability at the same time; high power can cause
suppression of small signals, reducing S/N and removing
AM, and it also creates IMs, which may fall on a weak
desired signal. This is the near-far problem of receiving
weak distant signals in the face of strong, local in-band
interferers. The interferer may only be in-band at RF or
through the first IF, but the receiver must remain linear
wherever the interferer exists. SFDR3 (3rd order spur free
dynamic range) is a figure of merit that gives the difference
(in dB) between a threshold level PT and the CW carrier
power that causes a single 3rd order IM to equal the noise
power in a given BW. This parameter is defined by the
receiver NF, IP3 and noise power BW as
SFDR3 = (IIP3 NFBW + 174)POffset, (4)
where IIP3 is the input IP3 in dBm, both NF and BW are in
dB, and POffset is the difference in dB between PT and the
noise power in BW [5]. This equation is usually shown with
POffset = 0 dB. However, it could be set to 10 dB (e.g.) toaccount for a minimum signal-to-noise (S/N) necessary in
BW, raising the minimum useful signal level and reducing
the dynamic range. Note that the IM power in BW will
reduce the minimum S/N to 7 dB in this example.
There are several dynamic range definitions, and it is not
always clear which one is invoked when a specification
simply states that the receiver dynamic range must exceed a
certain value. Some of these definitions are:
SFDR3 (as defined above) SFDR2: 2nd order SFDR, the difference between a
threshold level PT and the CW carrier level thatcauses a sum or difference frequency to equal the
noise power in a given BW. Many times PT is setequal to the noise power in BW.
P1dB PNOISE: distance between the input P1dB pointand the noise power in a given BW referred to thereceivers input
PDESENSITIZATION PNOISE: difference between theinput power which causes a specified amount ofdegradation (desensitization) and the noise power ina given BW referred to the receivers input
PCAR_MAXPNOISE: difference between the maximuminput carrier power for some specified degradationand the noise power in a given BW referred to thereceivers input
PCAR_MAX PCAR_MIN: difference between themaximum and minimum input carrier powers forspecified degradation(s)
Instantaneous SFDR3same as SFDR3 but with allvariable gain amps and attenuators fixed to respondto strong and weak signals with the same gains.
IV. NOISE POWER.
Several additional noise sources should be considered
beyond those typically assessed in a generic cascade
analysis. While noise floor and noise power within the noise
bandwidth (NBW) are important parameters to determine,
maintaining their values within acceptable limits does not
necessarily guarantee adequate system performance without
also evaluating and controlling these additional sources
discussed below.
A. Total Noise PowerTotal noise power from amplifiers over their individual
NBWs, can be much greater than the single fixed NBW
used for system evaluation. Crystal filters commonly
employed at lower frequency IFs can have 3 dB BWs which
are a tiny fraction of the IF amplifier(s) NBW. For example,
consider a 10 KHz crystal filter used in a 21.4 MHz IF strip
with amplifiers having significant gain out to 300 MHz. The
noise power over 300 MHz is 45 dB greater than that over
10 KHz and could easily compress or saturate the output of
an IF strip with relatively high gain and low P 1dB. Even if
the total (average) noise power is below P1dB, the 3 noise
peaks for additive white Gaussian noise (AWGN) are 9.5
dB higher given N0 = 2 [6]:
Pn (3) = [3(N0)]2 BW (5)
= 9 N0 BW, (6)
where N0 is noise density (W/Hz) and BW is in Hz. These
peaks can get clipped, making it no longer AWGN. It is
important to realize that noise saturation can occur even
though the signal is far below P1dB.
B.Image NoiseImage noise, without proper filtering prior to each mixer,
can increase the standard cascaded NF by up to 3 dB for
each conversion stage. To sufficiently lessen the impact of
image noise, an image reject filter must be placed close tothe mixers input. It is not sufficient to provide image
rejection to signals only (i.e., input filter followed by gain
prior to the mixer). The wideband response of the amplifiers
will fold over their noise generated at the image frequency
into the IF and can impact the system, even though the
image signals and noise prior to the filter have been
adequately suppressed.
C. Wideband, Unfiltered LO NoiseLO noise leaks through the mixer to the IF (LO-IF
isolation) and adds to the cascaded noise floor. The LO
chains output noise density at the input to the mixer LO
port must be assessed. High gains in the LO chain followedonly by a low pass filter (LPF) to remove harmonics
guarantee additional noise will be added to the IF, degrading
the NF predicted by the cascade analysis. A bandpass filter
(BPF) or highpass filter (HPF)
D.Reciprocal mixingReciprocal mixing occurs as a result of the transfer of
the LOs phase noise to each of the receive signals (in dBc)
via the convolution process of the mixer. Degradation to a
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weak signal can occur when high and low power in-band
signals are present that are closely spaced in frequency. The
added phase noise from the LO on the high power signal
may cover up the nearby lower power signal, significantly
reducing its S/N. For extremely large dynamic ranges
(PCAR_MAX PCAR_MIN), the AWGN of the LO may also
reduce the S/N of weak signals, independent of frequency
separation. The equivalent C/kT of the weak receive signal
in the presence of a large tone from reciprocal mixing is [7]:
( ) {(
) *( ) +
} (7)
.The (C/kT)LO term is either the LO carrier to noise density,
when assessing LO AWGN impacts, or it is the single side
band phase noise at a given frequency offset in dBc. Each
term above is in dB and must be converted to watts to do the
calculation and then back to dB again (10 log X) to give the
equivalent C/kT result in dB-Hz.
V. OUT-OF-BAND INTERFERENCE.
There are several mechanisms by which strong OOB
interferers at the input can degrade the performance of the
Rx chain and are similar in nature to how large in-band
signals degrade performance. The mitigation approach for
most of these involves filtering and higher intercept and
compression point devices. Since most microwave filters
have a re-entrant passband (PB) at 2Fo or 3Fo, an additional
low pass filter (LPF) should also be used to ensure signals
in one of these unprotected PBs dont degrade performance.
The filters ultimate rejection, which is limited by leakage
around it and isolation of its individual elements, is anotherreason to use additional filters, especially with very high
power OOB interferers. The magnitude of an OOB
interferer can be 200 V/m in accordance with MIL-STD-
461/464, incident power levels from some radar systems can
be orders of magnitude greater, and local TV stations can
have EIRPs up to 5 MW in the UHF frequency range.
A. Spurious ResponsesSpurs in-band to the IF output can also result from high
power OOB interferers (includes image spurious response).
The power at some OOB frequencies may be many 10 s of
dBs above the strongest desired in-band signals at the mixer,
which can produce very high spur levels relative to the
weaker desired signals. For example, consider a -3x2 spurwhere the OOB interferer power is filtered to not exceed the
mixers P1dB point of 0 dBm at its input (60 dB above a
desired -60 dBm signal). The mixer data sheet shows the
-3x2 spur level to be -50 dBc for an input of -10 dBm. For
the interferer level of 0 dBm, the spur relative to the
interferers power increases by 20 dB to -30 dBc in
accordance with (IAW) (2) (i.e., P x (|M|-1) = [0 dBm
(-10 dBm)] x (3-1)). The desired signal is 60 dB below the
OOB interferer, so the spur level with respect to the desired
carrier rises 60 dB to +30 dBc. Obviously, keeping the OOB
power just below the mixers P1dB point is not sufficient.
Additional input filtering is needed to drop the spur power
to at least 30 dB below the minimum desired signal level.
While this requires a 60 dB spur level reduction, it only
requires 20 dB of additional filter rejection. Recall the
earlier spur level discussion: absolute spur power at IF
(dBm) resulting from an OOB interferer varies IAW (1) as
P x |M|. Here we see that for a 3xN spur, an OOB power
change of only 20 dB yields the 60 dB spur reduction.
B. CompressionCompression of the front end and down-stream IF
components can occur until sufficient rejection is provided.
Compression from a large signal produces small signal
suppression of weak signals resulting in receiver
desensitization. The effective gain of a compressed stage is
reduced, degrading its ability to keep weak signals above
the noise floor of succeeding stages. For QAM signals, the
distance between the inner and outer points of theconstellation become compressed, causing degradation.
A weak signal experiences reduced gain from two
factors when passing through a stage that is driven into
compression by a large signal: device gain compression and
small signal suppression. Gain compression is determined
from the typical Pout vs. Pin curve as measured with the large
signal causing the compression. Small signal suppression is
an additional amount of up to 6 dB that only happens to
weak signals [8]. Gain seen by the weak (suppressed) signal
is given below where compression is < 4 dB:
GSUPPRESSED_SIGNAL GSS (small signal)GC
(compression) (small signal suppression). (8)
For a device at or driven beyond saturation (i.e., > 4 dB
compression), the suppressed signal gain is:
GSUPPRESSED_SIGNAL GSS10 dBP, (9)
where P is the amount the interferer power is above PSAT at
the input, and the 10 dB is comprised of 4 dB (compression
at saturation) and 6 dB (max small signal suppression). The
total gain reduction, GR, of a stage is:
GR 0 to 4 dB (compression)
+ 0 to 6 dB (suppression) +
P, (10)
where P only applies for a device at or beyond saturation.
A rule of thumb to ensure the gain seen by a weak signal is
not degraded more than 1 dB is to keep large signals 2 to 3
dB below input P1dB.
C. 3rdOrder IntermodsIMs (in-band) that result from OOB carriers can
dominate over those created solely from in-band carriers.
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The IM level depends on the IP3 of each stage and the
carrier levels throughout the chain, which depends not only
on the in-band G/L of each stage but also filter rejection. As
a result it is a little more involved to calculate the IM ratio
(IMR), the IM level compared to the desired carrier, than for
a standard in-band cascade. The individual IM generating
carrier levels can have different values at the Rx chain input
and typically experience different rejection amounts. Two-
tone and three-tone IM levels generated in any stage are
calculated as described below, with carrier levels defined at
the same location as IP3 (input or output).
Two-tone IMs are those generated from two carriers with
power in Watts defined as follows [9, 10]:
2
3
2
2
1
)2( 21 IP
PPP
FFIM
and
2
3
2
21
)2( 12IP
PPP
FFIM
(11)
where P1 and P2 are the power in Watts at frequencies F1
and F2, respectively, and IP3 is the 3rd order intercept point
in watts of the device creating the IMs.For powers in dBm, the equations become:
321)2( 2221 IPPPP FFIM (12)
321)2( 2212 IPPPP FFIM (13)
The 2-tone IMs are 6 dB lower in power than an IM
resulting from three carriers (3-tone IM), when the same
individual carrier power is used in both cases. The power in
a 3-tone IM is defined below [9, 10]:
23
321
)(
4
321IP
PPPP
FFFIM
W (14)
3321)( 26321 IPPPPP FFFIM dBm (15)
Management of these IMs is accomplished by filter
rejection, device IP3 and gain, and judicious gain
distribution throughout the Rx chain, especially through the
first mixer stage. For very large input levels additional
filters may be necessary to improve the ultimate rejection of
the front-end filter(s).
D.Harmonics2nd harmonic generation within the IF strip is a spurious
mechanism that is often overlooked. RF input frequenciesthat are offset from the tuned RF frequency by the IF
become in-band at the IF in two ways:
Mixer 2x-2 (or -2x2) spur from RF input and 2nd harmonic of IF/2
Both produce CW tone spurs for BPSK modulation (i.e.,
0/180 becomes 0/360).
The 2nd harmonic is 6 dB below 2nd order IMs (i.e.,
F1+F2, F2-F1) and experiences a 2:1 reduction (dBm) with
lower interferer levels [11].
Pspur= Pint(IP2Pint)6 dB
= 2PintIP26 dB (16)
A HPF or BPF is used in the IF strip to mitigate for less than
octave instantaneous IF BWs, but high IP2H (2nd harmonic
IP) is the only mitigation approach for larger BWs.A similar effect occurs for 3rd harmonic generation
within the IF strip, where RF input frequencies that are
offset from the tuned RF frequency by 2/3 the IF become in-
band at the IF in two ways:
Mixer 3x-3 (or -3x3) spur from RF input and 3rd harmonic of IF/3 (i.e., IF2/3 IF)
In this case the 3rd harmonic is 9.5 dB below 3rd order two-
tone IMs and experience a 3:1 reduction (dBm) with lower
interferer levels [10, 11].
Pspur= Pint2(IP3Pint)9.5 dB
= 3Pint2IP39.5 dB (17)
These spur generating frequencies are more easily filtered at
RF and in the IF strip than the IF offsets for 2 nd harmonic
generation. Higher IP3H devices also help to mitigate.
VI. AGC/ALC.
Automatic Gain or Level Control is employed in many
receivers as a means to increase dynamic range and hold
input power to the demodulator constant. There are two
main types of AGC by location:
Post-demod: reacts to the demodulated signal level(coherent AGC) + noise and any interference, primarily within the data filter BW. This type
responds to the desired signal and is relativelyinsensitive to undesired signals and interference (dueto the narrow data filter BW). The demod must belocked to the signal for a meaningful output to exist,and the AGC gain is typically at maximum prior tolock (no signal present).
Pre-demod: reacts to Signals + Noise + Interferers +Distortion products (spurs/IMs) in the wider IF BW prior to the AGCs detector. This type is used tomitigate front end compression, desensitization, orsmall signal suppression at the expense of NFdegradation. The benefits of its use are a trade-offbased on the EMI/RFI environment.
Either type can be an analog (continuously variable
attenuation/gain) or digital (step attenuator) implementation.Any AGC can be captured by or AGC on undesired
power. Strong OOB signals can easily exceed the weakest
desired signal power at the AGC detector due to insufficient
ultimate rejection of filters. An additional narrowband filter
may be necessary to prevent capture by undesired signals.
Total noise power can also capture a pre-demod AGC.
The IF BW can be many times larger than the occupied BW
of the desired signal, which affects the accuracy of the
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AGC. As the desired signal level approaches the total power
in the noise at the AGC detector, the gain control of the
AGC stops responding to the signal and maintains the S+N
constant, which is dominated by the total noise power. As a
result the signal continues to drop and eventually falls below
the operational range of the demod, even though adequate
S/N in the NBW may be present. This effect limits the
useful range of pre-demod AGCs from the maximum signal
level down to a level above the total noise power, which
typically is adequate to prevent compression.
Another example of AGC noise capture comes from the
earlier noise power discussion. Here a high gain IF strip has
multiple gain stages with a narrowband XTAL filter after
the first stage to limit IM generation. An AGC detector at
the end of this IF strip can easily be captured by noise with
filter-to-amp NBW ratios of 40 dB to 50 dB.
One final note concerns AGC control (or variable gain)
elements. Some voltage variable attenuators have the
undesirable characteristic that their input IP3 and P1dB do
not remain constant as their attenuation is varied. Worse,
some can exhibit both increasing and decreasing values (i.e.,their behavior is not monotonic). This makes it difficult to
predict overall receiver performance with one gain setting.
Input power levels must be swept (especially for devices
which are not monotonic) to exercise the AGC control
element while checking for compression and IMs. PIN
diode attenuators typically do not share this behavior.
VII. DEVICE PARAMETERVARIATIONS,ALIGNMENT, AND
GAIN COMPENSATION.
Parameters should be evaluated over statistical
tolerances, and frequency and environmental variations via
Monte-Carlo (MC) simulations with gain alignment and
compensation applied for each trial of random gain settings.Gain tolerance build up should be removed, and gain
compensation applied, often throughout the Rx chain for
best performance.
MC analysis is necessary to even find the worst-case (or
3) performance condition for anything other than a simple
Rx chain which does not use gain alignment, compensation,
or AGC. When an analysis is done using nominal values,
actual performance can be several dB (e.g., 5 dB) worse
than predicted, yielding production problems. On the other
hand, designing with absolute worst-case values
(simultaneously) will produce units that exceed
requirements by as much as 8 dB (depending on the
parameter and gain distribution), which leads to an over-designed unit and higher production costs. MC analysis can
make the difference between being able to produce a unit at
a reasonable cost versus a no bid for a very difficult set of
requirements.
Once a MC analysis has been completed, the Rx chain
should be evaluated at the condition which yielded the 3
performance for any parameter(s) that are non-compliant.
Often the offending part revealed is not the same as that
shown under nominal or even absolute simultaneous worst-
case condition (i.e., all high or low gains). Then optimize
the gain distribution and/or other parameters, as appropriate
(e.g., increase IP of the actual dominate component), and re-
run the analysis.
VIII. CONCLUSION.
Six key Rx chain design areas (spurious, cascaded
elements, non-standard noise, OOB interference, AGC, and
MC analysis) have been presented with several subtleties of
each discussed. The approach has been somewhat a design
check-list of topics to address for receiver design. The
importance of MC analysis to achieve a realizable design
that is cost effective cannot be understated.
Author. James B. Offner is an RF
Systems Engineer with Harris
Corporation in Melbourne, FL with
over 30 years of experience, working
on large multidiscipline programs
from mission requirements
determination and system analysisthrough HW/SW implementation. He has contributed to the
development of mobile tactical and large fixed strategic
satcom terminals, and recently was the RF System Architect
/Analyst for the Armys strategic MET terminal
development. He has performed analysis for development of
Navy shipboard terminals, operating with heavy EMI/cosite
interference and has developed analysis programs in use at
Harris for RF subsystem design. He was the Chief Systems
Engineer for the development of a vehicular, on-the-move
communication system, via aircraft relay, using multiple
phased array antennas. Jim earned his B.S.E.E. from
Michigan State University in 1977.
REFERENCES
[1] Daniel Cheadle, Selecting Mixers for Best InteremodPerformnace, 1993 Watkins-Johnson Co. Catalog Article
[2] William F. Egan, Practical RF System Design, Wiley-Interscience, 2003, pp. 171-180.
[3] Stephen A. Mass, Microwave Mixers, Artec House, 1996,pp. 151-154.
[4] William F. Egan, Practical RF System Design, Wiley-Interscience, 2003, pp. 49-53 and 91-122.
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[6] John G. Proakis, Digital Communications, McGraw-Hill,1983, pp. 93.
[7] Jim Offner, Internal Harris document, 2011.
[8] Robert M. Gagliardi, Satellite Communications, VanNostrand Reinhold, 1984, pp. 201-203.
[9] Stephen A. Mass, Microwave Mixers, Artec House, 1996,pp. 154-158.
[10] Hinrich Heynisch, Useful Design Criteria Predict TWTIntermod, MICROWAVES, March 1980
[11] Keneth A. Simons, The Decibel Relationships BetweenAmplifier Distortion Products, Proceedings of the IEEE,VOL. 58, NO. 7, July 1970