University of Calgary

PRISM: University of Calgary's Digital Repository

Graduate Studies The Vault: Electronic Theses and Dissertations

2014-09-29

Reconfigurable RF Devices Using Pneumatic Control

of Solid Dielectric Slugs

Wu, Billy

Wu, B. (2014). Reconfigurable RF Devices Using Pneumatic Control of Solid Dielectric Slugs

(Unpublished doctoral thesis). University of Calgary, Calgary, AB. doi:10.11575/PRISM/27560

http://hdl.handle.net/11023/1840

doctoral thesis

University of Calgary graduate students retain copyright ownership and moral rights for their

thesis. You may use this material in any way that is permitted by the Copyright Act or through

licensing that has been assigned to the document. For uses that are not allowable under

copyright legislation or licensing, you are required to seek permission.

Downloaded from PRISM: https://prism.ucalgary.ca

UNIVERSITY OF CALGARY

Reconfigurable RF Devices Using Pneumatic Control of Solid Dielectric Slugs

by

Billy Wu

A THESIS

SUBMITTED TO THE FACULTY OF GRADUATE STUDIES

IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE

DEGREE OF DOCTOR OF PHILOSOPHY

GRADUATE PROGRAM IN ELECTRICAL AND COMPUTER ENGINEERING

CALGARY, ALBERTA

September, 2014

c© Billy Wu 2014

Abstract

Many antennas reported with switchable polarization were designed using PIN diodes

because of their reliability and commercial availability. However, in realizing more

elaborate antennas, the biasing structures for these DC-controlled switching compo-

nents become more complex, which leads to design limitations, losses, and undesired

radiation. Other reconfigurable technologies, such as MEMS and liquid crystals, could

be chosen for a particular application depending on various trade-offs, for example

efficiency and ease of implementation. A recently developed scheme of using pneumat-

ically controlled solid dielectric slugs offers an alternative to the existing technologies.

These copper-backed slugs are confined within a dielectric channel adjacent to the

conducting layer. A slug is either directly above or situated away from a slit in the

conducting layer to realize its two switching states. Its translational movement is in-

duced by a pneumatic pressure gradient inside the channel. The technology is low-loss

and immune to RF distortion, and eliminates the need of DC biasing structures.

This technique was incorporated in the design of a reconfigurable ring slot an-

tenna capable of three polarization states (LP/LHCP/RHCP) at the 2.4-GHz ISM

band. The multilayered structure consisted of low-loss dielectric laminate layers that

were laser-machined to produce the precise profile and smooth edges required for

repeatable movement of the slugs. The measured axial ratio (AR) bandwidth is

5%, which compares favourably to similar antennas. Parametric analyses were per-

formed to accomplish the challenging task of realizing a design in which, the AR and

impedance bandwidths of all three states shared a common frequency band centered

ii

around 2.4 GHz. An overall three-state bandwidth of 2.4% was achieved, which is

the highest amongst similar antennas operating in the same frequency range. The

measured antenna performance — radiation patterns, gain values, AR, and |S11| —

in all three polarization states was consistent with the simulated performance. The

pneumatic actuation of four slugs inside a channel to realize the three different states

was successfully implemented, demonstrating that pneumatic slug control technology

is suitable for inclusion in reconfigurable antennas.

iii

Acknowledgements

The journey to this point of my academic life has been a wild yet very enjoyable

ride, and it was only possible because of the help, support, and mentoring from

many people. First and foremost, I would like to thank my supervisor Dr. Michal

Okoniewski. It has been a pleasure to work with Michal for all these years. His

wisdom, advice, and positive attitude have in no small way influenced my work ethic,

perception, and approach towards research and many other aspects of academia. He

has provided me with the best environment to learn, to perform research, and to grow

as a person. His tremendous support for me to pursue endeavors beyond research

— volunteering overseas, teaching, being involved in students’ associations, living a

second life in the mountains, to name a few — is rare to find in any supervisor.

I would also like to thank Dr. Chris Hayden for his help and unlimited support

in fabrication in this project. His experience and knowledge has proven invaluable,

and his accommodation and patience for my many requests and questions has been

greatly appreciated. His attention to detail and the high standard he upkeeps as

AMIF’s Operations Manager cannot go unmentioned in the success of this project.

His commitment to reading this thesis from cover to cover deserves special recognition.

I am also indebted to Dr. Mike Potter for his knowledge and support during my

pursuit of MSc and PhD degrees. His words of wisdom and timely encouragement

have helped me through a number of roadblocks I have encountered. His pragmatic

approach as a researcher and an educator has influenced my own approach in many

ways. Dr. Elise Fear and Dr. Bill Rosehart have offered assistance and advice on

iv

countless occasions. Working with them on research and on teaching has been an

enriching experience. Ideas for troubleshooting in the cleanroom are never in short

supply with AMIF’s Facility Manager Dr. Colin Dalton. This project would not have

started without his devotion to establishing AMIF over the years. His dedication

to a number of campus associations has been an inspiration. I am also indebted to

electrical engineering technicians Chris Simon and John Shelley for their technical

support. Chris’s positive outlook and enthusiasm towards education and mentoring

has resonated with me during many thought-provoking conversations.

Gratitude is owed to past AEG graduates: Adrian, Rudi, Greg, Imed, Yen, Kelly,

and Sean, who have been great mentors and role models for me. Many colleagues have

become great friends: Jeremie, Thomas, Andrew, Andy, Trevor, Robbie, Charlotte,

Andrea, John, Marcel N., Qiao, and Lincoln.

All my teachers in the past have played a role in shaping who I am today. Special

thanks go to Mr. Norm Sigalet and Mr. Dave Karbashewski for being two of the most

inspiring teachers I have ever had. I would not have entered electrical engineering

without their superb teaching in electro-technologies and physics during my high

school days.

This doctoral degree could not have been completed without the generous support

in funding from the Natural Sciences and Engineering Research Council of Canada

(NSERC), the Alberta Innovates - Technology Futures (AITF), and the Canadian

Microelectronics Corporation (CMC).

Last but not least, I would like to thank Marcel Seguin, who deserves a special

mention, for not cutting the rope when he had so many opportunities to do so.

v

To my family.

Your support means everything.

God bless you.

vi

Table of Contents

Abstract . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ii

Acknowledgements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . iv

Table of Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . vii

List of Tables . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . x

List of Figures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . xi

List of Abbreviations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . xvi

1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11.1 Project Goals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21.2 Thesis Outline . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

2 Background . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52.1 Circularly Polarized Antennas . . . . . . . . . . . . . . . . . . . . . . 5

2.1.1 Circular Polarization . . . . . . . . . . . . . . . . . . . . . . . 52.1.2 Figures of Merit . . . . . . . . . . . . . . . . . . . . . . . . . . 72.1.3 Printed Antennas with Switchable Polarization . . . . . . . . 10

2.2 Means of Reconfiguration . . . . . . . . . . . . . . . . . . . . . . . . 142.2.1 DC-Biased Switches . . . . . . . . . . . . . . . . . . . . . . . 152.2.2 Liquid Metals . . . . . . . . . . . . . . . . . . . . . . . . . . . 162.2.3 Permittivity Manipulation Techniques . . . . . . . . . . . . . 172.2.4 Flexible Materials . . . . . . . . . . . . . . . . . . . . . . . . . 20

2.3 Selecting a Circularly Polarized Antenna . . . . . . . . . . . . . . . . 232.4 Realizing Circular Polarization with Ring Slot Antenna . . . . . . . . 25

3 Pneumatically Controlled Switching Mechanism . . . . . . . . . . . 333.1 Concept . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 343.2 Process and Material Selection for Channel Fabrication . . . . . . . . 363.3 Processing SU8 Photoresist . . . . . . . . . . . . . . . . . . . . . . . 383.4 Channel Configurations . . . . . . . . . . . . . . . . . . . . . . . . . . 43

3.4.1 Simple Channels . . . . . . . . . . . . . . . . . . . . . . . . . 443.4.2 More Complex Configurations . . . . . . . . . . . . . . . . . . 46

4 Design and Fabrication of a Capacitive Switch . . . . . . . . . . . . 52

vii

4.1 Structure and Fabrication Process . . . . . . . . . . . . . . . . . . . . 534.2 Simulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 564.3 Fabrication and RF Measurement . . . . . . . . . . . . . . . . . . . . 584.4 Pressure Measurement . . . . . . . . . . . . . . . . . . . . . . . . . . 674.5 Discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68

5 Ring Slot Antenna with Switchable Polarization . . . . . . . . . . . 715.1 Changing Polarization Using Pneumatic Control . . . . . . . . . . . . 72

5.1.1 Linear Polarization . . . . . . . . . . . . . . . . . . . . . . . . 725.1.2 Right-Hand and Left-Hand Circular Polarizations . . . . . . . 73

5.2 Design using glass wafers and photoresist . . . . . . . . . . . . . . . . 765.2.1 Dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . 775.2.2 Parametric Analysis . . . . . . . . . . . . . . . . . . . . . . . 815.2.3 Simulated Performance . . . . . . . . . . . . . . . . . . . . . . 905.2.4 Sensitivity to Fabrication Tolerances . . . . . . . . . . . . . . 1035.2.5 Modeling for Measurement of Fabricated Antenna . . . . . . . 109

5.3 Design Using Laminate Boards . . . . . . . . . . . . . . . . . . . . . 1125.3.1 Structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1135.3.2 Simulated Performance . . . . . . . . . . . . . . . . . . . . . . 117

5.4 Design Advantages and Disadvantages . . . . . . . . . . . . . . . . . 127

6 Fabrication and Measurement of Ring Slot Antenna . . . . . . . . 1296.1 Fabrication . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 129

6.1.1 Glass Wafers as Substrate . . . . . . . . . . . . . . . . . . . . 1306.1.2 Laminate Boards as Substrate . . . . . . . . . . . . . . . . . . 1336.1.3 Pneumatic Control and Pressure Measurement . . . . . . . . . 136

6.2 Antenna Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . 1396.2.1 Antenna Patterns . . . . . . . . . . . . . . . . . . . . . . . . . 1426.2.2 Return Loss (−|S11|) . . . . . . . . . . . . . . . . . . . . . . . 1456.2.3 Axial Ratio (AR) . . . . . . . . . . . . . . . . . . . . . . . . . 1476.2.4 Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1486.2.5 Performance Summary . . . . . . . . . . . . . . . . . . . . . . 150

7 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1537.1 Contributions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1557.2 Future Work . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 156

7.2.1 Further Characterization of Switching Mechanism . . . . . . . 1567.2.2 Slug Position Control . . . . . . . . . . . . . . . . . . . . . . . 1567.2.3 A Complete System . . . . . . . . . . . . . . . . . . . . . . . . 1577.2.4 Miniaturization . . . . . . . . . . . . . . . . . . . . . . . . . . 158

viii

7.2.5 Other Antenna Designs . . . . . . . . . . . . . . . . . . . . . . 160

Bibliography . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 163

A Antenna Measurements in Anechoic Chamber . . . . . . . . . . . . 171A.1 Linear Polarization Measurements . . . . . . . . . . . . . . . . . . . . 172A.2 Circular Polarization Measurements . . . . . . . . . . . . . . . . . . . 174

A.2.1 Derivation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 175A.2.2 Procedure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 178

B Miniature Pumps . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 181

C Photolithography Process of SU8-2100 . . . . . . . . . . . . . . . . 183C.1 Initial Preparation . . . . . . . . . . . . . . . . . . . . . . . . . . . . 183C.2 Substrate Pretreat . . . . . . . . . . . . . . . . . . . . . . . . . . . . 185C.3 Coat . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 186C.4 Soft Bake . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 187C.5 Expose . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 187C.6 Post Exposure Bake (PEB) . . . . . . . . . . . . . . . . . . . . . . . 188C.7 Develop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 188

D Antenna Radiation Efficiency Measurements . . . . . . . . . . . . . 189

ix

List of Tables

2.1 Merits and drawbacks of various types of circularly polarized antennas. 242.2 Performance of notable antennas in literature with switchable polar-

ization in the 2.4 GHz range. . . . . . . . . . . . . . . . . . . . . . . . 32

4.1 Summary of de-embedded measured results. . . . . . . . . . . . . . . 69

5.1 Layer thickness of the ring slot antenna with glass substrate. . . . . . 775.2 Dimensions of the ring slot antenna with glass substrate. . . . . . . . 815.3 Simulated performance of the ring slot antenna using glass substrate. 995.4 Layer thickness of the ring slot antenna with laminate substrate. . . . 1165.5 Dimensions of the ring slot antenna with laminate substrate. . . . . . 1175.6 Simulated performance (realistic model) of the ring slot antenna using

laminate substrate. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124

6.1 Measured and simulated performance of the ring slot antenna. . . . . 1506.2 Comparison of antenna performance with similar reconfigurable anten-

nas in literature. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 152

B.1 Specifications of some commercially available miniature pumps. . . . 182

x

List of Figures

1.1 Thesis overview. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

2.1 Illustration of circular polarization. . . . . . . . . . . . . . . . . . . . 72.2 Examples of printed antennas. . . . . . . . . . . . . . . . . . . . . . . 122.3 Various designs of printed antennas with switchable polarization. . . . 142.4 Examples of using liquid metals in RF devices. . . . . . . . . . . . . . 172.5 Reconfigurable reflectarray design using COSMIX. . . . . . . . . . . . 202.6 Switch design based on pneumatically actuated membranes. . . . . . 212.7 Flexible patch antenna. . . . . . . . . . . . . . . . . . . . . . . . . . . 222.8 Ring slot antenna with switchable polarization using PIN diodes. . . 262.9 Field distribution on the ring slot antenna. . . . . . . . . . . . . . . . 272.10 Comparison of perturbation on the ring slot current for different posi-

tion of the rectangular opening. . . . . . . . . . . . . . . . . . . . . . 282.11 Illustration of phase lag in the radiated field due to the lowering of

resonant frequency of an antenna. . . . . . . . . . . . . . . . . . . . . 292.12 Ring slot antenna operating in LHCP and RHCP states. . . . . . . . 30

3.1 Pneumatic control of the slug position and the resulting capacitancevariation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

3.2 Computer screen-shot of the photomask design used to build the testchannels. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

3.3 Adhesion issue in the development of the SU8 process. . . . . . . . . 423.4 SU8 resist successfully deposited and developed. . . . . . . . . . . . . 433.5 Testing of the switching mechanism with simple channels. . . . . . . . 453.6 A meandering channel. . . . . . . . . . . . . . . . . . . . . . . . . . . 473.7 A device with multiple chambers in each channel path. . . . . . . . . 483.8 Prototype demonstrating the concept of channel blocking. . . . . . . 493.9 Concept of a nine-state device and its fabricated channels. . . . . . . 50

4.1 Exploded view of the switch showing various layers involved. . . . . . 544.2 Side profile illustrating two states of the switch. . . . . . . . . . . . . 554.3 Custom fabrication process. . . . . . . . . . . . . . . . . . . . . . . . 564.4 Simulated insertion loss of the closed switches. . . . . . . . . . . . . . 574.5 Simulated return loss of the closed switches and isolation of the open

switches. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 584.6 Cracked glass cover from mechanical drilling. . . . . . . . . . . . . . . 59

xi

4.7 Copper adhesion issue on the patterned glass wafer. . . . . . . . . . . 60

4.8 Fabricated switch in closed and open positions. . . . . . . . . . . . . 61

4.9 Experimental measurement setup of the switch. . . . . . . . . . . . . 62

4.10 Repeatability of the switch. . . . . . . . . . . . . . . . . . . . . . . . 62

4.11 Concept of TRL calibration. . . . . . . . . . . . . . . . . . . . . . . . 63

4.12 Custom TRL calibration standards. . . . . . . . . . . . . . . . . . . . 64

4.13 Measured performance of the open switch. . . . . . . . . . . . . . . . 65

4.14 Measured performance of the closed switch. . . . . . . . . . . . . . . 66

4.15 Copper-coated glass slugs. . . . . . . . . . . . . . . . . . . . . . . . . 66

4.16 Setup of pressure measurement. . . . . . . . . . . . . . . . . . . . . . 68

5.1 Structure of the switchable polarization ring slot antenna. . . . . . . 73

5.2 Direction of air flow and the resulting positions of slugs for generatinglinear polarization. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74

5.3 Direction of air flow and the resulting positions of slugs for generatingRHCP. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75

5.4 Direction of air flow and the resulting positions of slugs for generatingLHCP. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76

5.5 Dimensions of the ring slot with glass substrate. . . . . . . . . . . . . 78

5.6 Dimensions of microstrip-to-slotline transition with glass substrate. . 79

5.7 Simulation model of the reconfigurable ring slot antenna with glasssubstrate in HFSS. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80

5.8 Minimum AR achieved for various slug widths, slug lengths, and airgap separations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84

5.9 Frequency of minimum AR for various slug widths, slug lengths, andair gap separations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85

5.10 AR bandwidth obtained for various slug widths, slug lengths, and airgap separations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86

5.11 Minimum AR achieved for various ring outer radii, slug widths, andair gap separations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87

5.12 Frequency of minimum AR for various ring outer radii, slug widths,and air gap separations. . . . . . . . . . . . . . . . . . . . . . . . . . 88

5.13 AR bandwidth obtained for various ring outer radii, slug widths, andair gap separations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89

5.14 |S11| of the simulated ring slot antenna (glass substrate) under differentpolarization states. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 90

5.15 S11 (on the Smith Chart) of the simulated ring slot antenna (glasssubstrate) from 2 to 2.8 GHz under different polarization states. . . . 91

5.16 Broadside AR of the simulated ring slot antenna (glass substrate). . . 92

xii

5.17 AR of the simulated ring slot antenna (glass substrate) in the twoprincipal planes at 2.38 GHz. . . . . . . . . . . . . . . . . . . . . . . 93

5.18 Broadside gain of the simulated ring slot antenna (glass substrate). . 94

5.19 Simulated antenna pattern for φ = 0 and φ = 90 planes when oper-ating in the RHCP state at 2.38 GHz (glass substrate). . . . . . . . . 95

5.20 Simulated antenna pattern for φ = 0 and φ = 90 planes when oper-ating in the LHCP state at 2.38 GHz (glass substrate). . . . . . . . . 96

5.21 Simulated antenna pattern for φ = 0 and φ = 90 planes when oper-ating in the LP state at 2.38 GHz (glass substrate). . . . . . . . . . . 98

5.22 Ground current at different phases of a cycle in the LP state (glasssubstrate). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100

5.23 Ground current at different phases of a cycle in the RHCP state (glasssubstrate). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 101

5.24 Ground current at different phases of a cycle in the LHCP state (glasssubstrate). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102

5.25 Parameters considered in the sensitivity analysis. . . . . . . . . . . . 104

5.26 Performance variation due to changing hair (glass substrate). . . . . . 105

5.27 Simulated antenna pattern for different air gap separations hair (glasssubstrate). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106

5.28 Performance variation due to misalignment between the ground waferand the channel/slugs (glass substrate). . . . . . . . . . . . . . . . . . 107

5.29 Performance variation due to misalignment between the ground waferand the microstrip wafer (glass substrate). . . . . . . . . . . . . . . . 108

5.30 Simulation setup for antenna measurement (glass substrate). . . . . . 110

5.31 Comparison of various additional features in the simulation model onthe AR performance (glass substrate). . . . . . . . . . . . . . . . . . 111

5.32 Comparison of various additional features in the simulation model onthe |S11| performance (glass substrate). . . . . . . . . . . . . . . . . . 112

5.33 Various layers composing the ring slot antenna, with laminate substrate.114

5.34 Magnified plan view of the ring slot antenna with laminate substrateand channel. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 115

5.35 Simulation setup for antenna measurement (laminate substrate). . . . 118

5.36 |S11| of the simulated antenna (laminate substrate). . . . . . . . . . . 119

5.37 Broadside AR of the simulated antenna (laminate substrate). . . . . . 120

5.38 Broadside gain of the simulated antenna (laminate substrate). . . . . 121

5.39 Comparison of various additional features in the simulation model ongain performance (laminate substrate) for the RHCP state. . . . . . . 122

5.40 Simulated antenna pattern for φ = 0 and φ = 90 planes when oper-ating in the RHCP state at 2.4 GHz (laminate substrate). . . . . . . 123

xiii

5.41 Ground current at different phases of a cycle in the RHCP state (lam-inate substrate). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 125

5.42 Comparison of impedance and AR bandwidths (laminate substrate,realistic model). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 126

6.1 Fabricated channel layer on glass wafer. . . . . . . . . . . . . . . . . . 1306.2 Delamination of copper after etching. . . . . . . . . . . . . . . . . . . 1326.3 Fabricated antenna with laminate substrate (ground plane side). . . . 1336.4 Fabricated antenna with laminate substrate (microstrip side). . . . . 1346.5 View under the microscope of a slug resting in the channel. . . . . . . 1356.6 Positions of slugs for different polarization states. . . . . . . . . . . . 1356.7 Setup for the pneumatic control test of the antenna and the actuation

sequence of air inflow for each polarization state. . . . . . . . . . . . . 1376.8 Two-step sequence of air input and the slug movement for realizing

each polarization state. . . . . . . . . . . . . . . . . . . . . . . . . . . 1386.9 Conceptual diagram of the antenna measurement setup inside the ane-

choic chamber. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1406.10 Standard gain horn and the ring slot antenna (AUT). . . . . . . . . . 1406.11 Setup of the ring slot antenna on the turntable for scanning φ = 0

(left) and φ = 90 (right) planes. . . . . . . . . . . . . . . . . . . . . 1416.12 Comparison between measured and simulated antenna patterns when

operating in the LP state at 2.4 GHz. . . . . . . . . . . . . . . . . . . 1426.13 Comparison between measured and simulated antenna patterns when

operating in the RHCP state at 2.4 GHz. . . . . . . . . . . . . . . . . 1436.14 Comparison between measured and simulated antenna patterns when

operating in the LHCP state at 2.4 GHz. . . . . . . . . . . . . . . . . 1446.15 Comparison between measured and simulated |S11| of the ring slot

antenna under different polarization states. . . . . . . . . . . . . . . . 1466.16 Comparison between measured |S11| with and without pneumatic ac-

tuation setup. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1476.17 Comparison between measured and simulated broadside AR of the ring

slot antenna under different polarization states. . . . . . . . . . . . . 1486.18 Comparison between measured and simulated broadside gain of the

ring slot antenna under different polarization states. . . . . . . . . . . 1496.19 Comparison of measured impedance and AR bandwidths. . . . . . . . 151

7.1 Introducing pressure gradient in the orthogonal direction. . . . . . . . 1577.2 Exploded view of the frequency-switchable aperture-coupled antenna

with a pneumatically controlled swinging arm. . . . . . . . . . . . . . 1617.3 Operation of the frequency-switchable antenna. . . . . . . . . . . . . 162

xiv

A.1 Generalization of two orthogonal linear components. . . . . . . . . . . 176A.2 Linearly polarized antenna measurements required for obtaining CP

patterns of the AUT in φ = 0 and φ = 90 planes. . . . . . . . . . . 180

C.1 SU8-2100 process. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 183

D.1 Wheeler Caps. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 190D.2 S11 of the AUT inside the small Wheeler Cap, shown on the Smith Chart.194D.3 |S11| of the AUT inside the Wheeler Caps. . . . . . . . . . . . . . . . 195D.4 Determining the reflection circle from the measured S11 on the Smith

Chart, with the AUT inside the small Wheeler Cap. . . . . . . . . . . 196D.5 Determining the reflection circle from the measured S11 on the Smith

Chart, with the AUT inside the large Wheeler Cap. . . . . . . . . . . 197

xv

List of Abbreviations

Abbreviations

AMIF Advanced Micro/nanosystems integration FacilityAR axial ratioAUT antenna under testBST barium strontium titanateBW bandwidthCMOS complementary metal-oxide-semiconductorCOSMIX coaxial stub microfluidic impedance transformerCP circular polarization / circularly polarizeddB decibelsDUT device under testFET field-effect transistorGNSS global navigation satellite systemGPS global positioning systemHMDS hexamethyldisilazaneISM industrial, scientific, and medical equipmentLHCP left-hand circular polarization / left-hand circularly polarizedLP linear polarization / linearly polarizedMEMS micro-electromechanical systemsPCB printed circuit boardPDMS polydimethylsiloxaneRF radio frequencyRHCP right-hand circular polarization / right-hand circularly polarizedSOI silicon-on-insulatorTRL Thru-Reflect-LineUV ultravioletVNA vector network analyzer

xvi

1

Chapter 1

Introduction

As technologies such as the global positioning system (GPS) — an example of a global

navigation satellite system (GNSS) — permeate modern society, circularly polarized

antennas, which are a major building block in GPS, continue to garner attention from

antenna researchers and designers. Most antennas are either linearly polarized or

circularly polarized by design. Antenna polarization, which is one of the fundamental

properties of any antenna, is generally fixed. The ability to dynamically change

the antenna polarization between linear and circular represents an active category of

research in reconfigurable antenna designs. An antenna possessing such an ability is a

reconfigurable antenna with switchable polarization. Multi-functionality and space-

saving are two of the main driving forces behind the interest in reconfigurable antenna

designs. The idea of having one reconfigurable antenna that provides the functions of

multiple fixed antennas is attractive in a number of telecommunications applications,

such as polarization diversity systems and frequency reuse schemes [1]–[3].

Specific devices or structures need to be incorporated in an antenna to realize

reconfiguration. Some are simple two-terminal circuit components that can be sol-

dered onto the conducting layers of the antenna directly, such as PIN diodes1 and

micro-electromechanical-system (MEMS) switches; while others are embedded in the

1“PIN” represents the three semiconductor regions of the device: p-type, intrinsic, and n-type.

2

substrate layer of the antenna, as in the case of liquid crystals. Some are mature

technologies that are commercially available, while some are gaining popularity as

continual research demonstrates that their advantages outweigh their shortcomings

for a variety of applications. The choice for a particular end-use application is decided

based on the prioritization of various trade-offs: efficiency, size, ease of fabrication,

cost, design complexity, power handling, response time, and repeatability. By ex-

ploring alternative mechanisms, such as the technology introduced in this thesis, an

antenna designer has more options when designing a reconfigurable antenna.

1.1 Project Goals

Developing a novel scheme of reconfiguration for practical RF devices such as an

antenna with switchable polarization is the focus of this project.

In accomplishing this objective, some specific goals are:

• Perform a literature review of existing schemes for reconfiguration with the aim

to evaluate the advantages and disadvantages of each technology;

• Select a topology for the antenna with switchable polarization that is suitable

for implementing the novel scheme;

• Develop a fabrication process that can reliably produce prototypes of RF devices

implementing the novel scheme;

• Build an RF switch, which is the most basic RF component, using the novel

scheme to confirm the viability of the scheme and to make modifications to the

fabrication process that are deemed necessary;

• Design and simulate the antenna to understand its behaviour and performance

prior to its fabrication;

3

• Fabricate the antenna and test the pneumatic actuation for realizing the three

different polarization states; and

• Measure the performance of the fabricated antenna and compare the results

with those obtained from simulations.

1.2 Thesis Outline

As mentioned in Section 1.1, the main objective in this thesis is the exploration of a

novel switching mechanism in the design of an antenna with switchable polarization.

The steps taken and the contributions made towards fulfilling this objective are orga-

nized and documented in following chapters. The relations between the topic of each

chapter are illustrated in Fig. 1.1.

• Chapter 2 provides the necessary background information and literature review

for evaluating the advantages and disadvantages of existing switching mecha-

nisms and selecting an appropriate circularly polarized antenna to implement

the novel mechanism developed in this project.

• Chapter 3 introduces the concept of using pneumatic control to manipulate

the positions of solid slugs, the fabrication procedure, and the results of some

fabricated prototypes to demonstrate the mechanical feasibility in switching.

• Chapter 4 documents the investigation of the RF behaviour of the switching

mechanism via the fabrication of an RF switch. The measured RF performance

is compared with the simulated results. The merits and drawbacks of the mech-

anism are discussed.

• Chapter 5 describes the design of a ring slot antenna implementing the mecha-

nism to achieve switchable polarization. The simulated results are analyzed for

4

Novel

Switching

Mechanism

Reconfigurable

Antenna

Selecting a

circularly

polarized

antenna

Chapter 2

Evaluating

pros & cons

of existing

mechanisms

Chapter 2 Developing

pneumatically-

controlled

mechanism

Chapter 3

Pneumatically-

controlled

capacitive

switch

Chapter 4

Designing

reconfigurable

ring slot

antenna

Chapter 5

Fabricating

reconfigurable

ring slot

antenna

Chapter 6

Literature Review

Thesis Contribution

Thesis Objective

Figure 1.1: Thesis overview.

two designs, each using a different fabrication procedure and set of materials.

The two methods of fabrication are compared in terms of practicality, cost, and

ease of manufacturing.

• Chapter 6 compares the measured results of the fabricated antenna against

the simulated results to illustrate the viability of the design procedure and the

chosen fabrication method. Mechanical testing of the switching mechanism in

the antenna is performed. Minor defects and deviations from the expected

behaviour and performance are addressed.

• Chapter 7 summarizes the technical and scholarly contributions of this project,

and offers some avenues for furthering this technology and related research ideas

that could be pursued in the future.

5

Chapter 2

Background

In this chapter, the motivation behind designing a reconfigurable circularly polarized

(CP) antenna with switchable sense of polarization is provided. A literature review

of some of the existing designs on CP antennas facilitates the selection of an antenna

type suitable for implementing the novel switching mechanism proposed in this thesis.

Attention is drawn to both the merits and drawbacks of existing means of reconfig-

uration to justify the exploration of the novel switching mechanism. The ring slot

antenna — the structure chosen for this project — is used as an example to explain

the theory behind the realization of circular polarization using perturbations.

2.1 Circularly Polarized Antennas

2.1.1 Circular Polarization

One of the characteristics associated with any antenna is the polarization of the

radiation it generates in the transmitting mode (or, the polarization of the radiation

it is most capable of receiving in the receiving mode). Elliptical polarization is the

most general case. The radiated field, over time, sketches an ellipse on the plane

perpendicular to the direction of field propagation. Linear polarization (LP) and

circular polarization (CP) are two special cases of elliptical polarization [4]. When

the length of the minor axis of the ellipse approaches zero, the ellipse becomes a line,

6

which is the case for LP. When the minor and major axes of the ellipse have the same

length, the ellipse becomes a circle, which is the case for CP.

Circular polarization, as the name suggests, is characterized by the ideally perfect

circle sketched out by the tip of the rotating field vector over time, when viewed

directly along the field propagation axis. CP is either left-handed (LHCP) or right-

handed (RHCP), as shown in Fig. 2.1, which shows a field vector rotating over time

as the field propagates in the +z-direction. A CP signal propagating towards an

observer while rotating in a clockwise fashion is an LHCP signal; and conversely

for counter-clockwise rotation and an RHCP signal. Eq. 2.1 shows how a CP field,

traveling in the +z-direction, can be decomposed into two orthogonal LP fields (Ex &

Ey) of equal magnitude and quadrature-phase (90 apart) [4]. Derivation and detailed

discussion can be found in Appendix A.2.1.

ERHCP =1√2(Ex + jEy) =

1√2(Ex + Eye

j(π/2))

ELHCP =1√2(Ex − jEy) =

1√2(Ex − Eye

j(π/2)) (2.1)

CP offers several advantages over its linear counterpart in some applications. For

instance, CP is preferred in satellite communications (such as GPS) because the signal

polarization purity is not affected by the Faraday rotation effect [4], [5, ch.54]. The

polarization of a LP wave, on the other hand, will rotate to a certain extent, often

unpredictably, during its propagation in the ionosphere, which can lead to polarization

inefficiency. LHCP and RHCP can also be utilized in polarization diversity schemes

to mitigate multipath fading [1],[6]. Modulation schemes using CP have also been

7

(b) RHCP

(a) LHCP

z

z

x

y

x

y

x

y

x

y

Field Vector

Figure 2.1: Illustration of circular polarization: (a) LHCP. (b) RHCP.

demonstrated [7]. A variety of antennas — specifically, printed antennas — for CP

applications are available, and the selection of an appropriate CP antenna for this

project will be discussed in Section 2.3.

2.1.2 Figures of Merit

This section introduces the figures of merit associated with circularly polarized an-

tennas, with which the antenna built in this project can be compared against existing

antennas published in literature.

2.1.2.1 Return Loss (−|S11|)

Generally, the majority of the incident power supplied from the power source to an

antenna is accepted by the antenna while a portion is reflected back to the power

8

source. Return loss, defined as −|S11| and expressed in the unit of decibels (dB)1, is

a metric for quantifying how well an antenna accepts power supplied to it (when it

is transmitting; the concept is the similar when the antenna is used as a receiving

antenna). It is essentially a ratio of the incident power to the reflected power. An

antenna that is well-matched to the source impedance — 50Ω is the standard in the

RF world — at a particular frequency will exhibit a return loss that is a large positive

dB value (while |S11| is a large negative dB value).

In standard antenna practice, a return loss greater than 10 dB (i.e. |S11| <

−10 dB) is considered acceptable for an antenna. The concept of impedance band-

width (BW) stems from this convention. The impedance bandwidth of an antenna is

the range of frequency within which |S11| < −10 dB or, in other words, within which

the antenna accepts at least 90% of the power supplied to it.

2.1.2.2 Axial Ratio (AR)

Axial ratio, which is a crucial parameter of a CP antenna, offers a quantitative de-

scription of how pure the CP radiation produced by the antenna is. It is the ratio of

the lengths of the major and minor axes of the ellipse [8], as expressed in Eq. A.12.

The lower the AR is, the purer the CP radiation is. The lowest achievable AR value is

1 (or 0 dB), which describes a perfect circle. The standard practice in designing a CP

antenna is to maintain AR < 3 dB in the desired frequency range of operation. This

gives rise to the metric of AR bandwidth. Typically, the AR bandwidth is smaller

than the impedance bandwidth because maintaining the condition of AR < 3 dB is

usually more challenging than upholding |S11| < −10 dB. Note that the impedance

1S11 represents the reflection coefficient at port 1 of a two-port network when port 2 is matched.

9

bandwidth generally does not encompass the AR bandwidth entirely by default. De-

sign parameters often need to be optimized in order for this condition to be satisfied.

2.1.2.3 Gain

When it is transmitting, a directional antenna has the property of having its radiated

energy more focused in certain direction(s), instead of being equally radiated in all

directions in free space. Conversely, when it is receiving, its ability to receive sig-

nal from certain direction(s) is greater, as opposed to being equally receptive in all

directions. The gain of an antenna describes how directive the antenna is.

Gain can be expressed as a numeric value or in the unit of dBi2. Gain of an

antenna in a particular direction is the ratio of the radiation intensity in that direction

to the average radiation intensity of the hypothetical isotropic antenna radiating the

same amount of energy. The greater the gain is in a direction, the more directive the

antenna is in that direction. Note that the gain of an antenna, by definition, accounts

for the radiation efficiency (how efficient the antenna is in minimizing dissipative

losses) but not the antenna mismatch [9].

Some antenna types are more directive than others, and generally each antenna

type has an expected range of gain it is capable to achieve. For instance, a typical

horn antenna may have a gain in the range of 15 to 20 dBi [10], while the directivity3

of a rectangular microstrip antenna with a thin substrate ranges from 5 to 8 dBi [5,

ch.7].

2The “i” stands for isotropic, which describes the hypothetical (idealized) antenna that canradiate its energy equally in all directions, such that its radiation intensity is constant over theentire sphere.

3Directivity is essentially gain but without accounting for the radiation efficiency. Therefore, thedirectivity value for an antenna is always higher than the gain value.

10

The gain of an antenna generally implies the gain in the co-polarization, which

is the polarization coinciding with the intended polarization of the antenna. Cross-

polarization, on the other hand, is the orthogonal polarization. For example, the

signal strength of the y-polarized radiation represents the level of cross-polarization

for an x-polarized LP antenna. As a second example, the RHCP signal is the cross-

polarized radiation for an LHCP antenna. It is in the interest of an antenna designer to

keep the cross-polarization level low in the intended direction(s) of signal propagation.

For a CP antenna, a low cross-polarization in a particular direction translates to a

low AR value, meaning that the antenna has good polarization purity in that specific

direction of propagation.

The main performance metrics that will be used to evaluate the antenna built

for this project have been described above. In the following section, the motivation

behind designing a reconfigurable antenna with switchable polarization, and a brief

history of the development of this type of antennas will be given.

2.1.3 Printed Antennas with Switchable Polarization

Reconfigurable CP antennas add a new dimension to the design of CP communica-

tions systems. Reconfiguration in the sense of polarization, the direction of maximum

radiation (boresight), or the frequency of operation can greatly increase the adapt-

ability of the antennas. With the latter two aspects of reconfiguration, electronically-

controlled target tracking and multiband operations become a possibility. The need

for switchable polarization arises from the existence of applications — such as polar-

ization diversity systems [6], the study of thickness sensitivity of anisotropic slab [11],

and magnetic field diagnostics in the corona [12] — in which the ability to transmit or

11

receive both LHCP and RHCP signals at the same location is required. In some sce-

narios, two antennas (one for each type of CP signal) need to function simultaneously,

where the concept of switchable polarization is not applicable. In other cases, how-

ever, only one type of CP signal needs to be transmitted or received at any particular

moment. A setup with a single reconfigurable antenna that can dynamically switch

between LHCP and RHCP is much more attractive than a system that requires two

fixed antennas and a dedicated switching network to perform the same function when

factors such as cost of production and maintenance, size, versatility, and efficiency

are considered. This is one of the sources of motivation to pursue reconfigurable

CP antennas with switchable polarization, besides the intention to implement a new

reconfigurable technology in a practical RF device, as outlined in Chapter 1.

As alluded to in Section 2.1.1, printed antennas are of interest because of:

• their relative ease of fabrication;

• their status of being a mature research topic, with a wealth of theoretical and

experimental knowledge available in literature; and

• the relative ease of integrating elements of reconfiguration, such as RF switches.

The simplest form of a printed antenna is a patterned metalized layer on one

surface of a dielectric substrate, with or without a second metalized layer on the other

surface. A printed antenna with only one metalized layer typically radiates on both

sides and has a symmetrical radiation pattern about the plane of the antenna. An

example is a slot antenna, depicted in Fig. 2.2(a). A printed antenna with metalized

layers on both sides (usually in the form of a patterned patch on top of a ground

plane) is called a microstrip antenna (Fig. 2.2(b)), and most of its radiated energy is

directed on one side (top hemisphere in the case of Fig. 2.2(b)). Due to their simple

12

structure and manufacturing process, which is very similar to that of printed circuit

boards (PCBs), printed antennas have been a popular choice for antenna designers

since 1970s [13].

Ground

Plane

(a) (b)Slot

Substrate

PatchBroadside Broadside

Broadside

Figure 2.2: Examples of printed antennas: (a) slot antenna, (b) microstrip antenna.

Printed antennas with simple geometric shapes such as rectangles, circles, and

annular rings are generally LP when they are single-fed (i.e. one point of excitation)

[14]. The boresight is generally in the broadside direction (see Fig. 2.2), meaning that

maximum radiation is along the normal direction to the plane of the antenna. The

most common method to achieve CP on a printed antenna is to simultaneously excite

two orthogonal linear modes which are of equal amplitude and phased 90 apart at the

operating frequency [13]. The theory behind the generation of CP will be discussed in

greater detail in Section 2.4 for the case of a ring slot antenna. Designing an antenna

with a single feed can be achieved either: by introducing perturbations strategically

on the metalized layer(s); by using a section of quarter-wavelength transmission line

or a hybrid power divider for quadrature-phase feed; or by other similar methods to

manipulate the currents on the metalized layer(s).

Designing a fixed (i.e. not reconfigurable) CP printed antenna can be considered

13

a mature research topic [13]. With the advances in RF switching elements such as

PIN diodes and MEMS switches, reconfigurable printed antennas with a switchable

sense of polarization have been successfully demonstrated as well; examples include:

• switching between the two ports to a semi-annular proximity feed line for a ring

antenna [15] (Fig. 2.3(a));

• switching the order of the quadrature-phase feed to a pair of crossed dipoles

[16];

• switching between shorting posts in the substrate of a square patch [17];

• switching between two pairs of triangular conductors on a corner-truncated

square patch [3] (Fig. 2.3(b));

• switching between rectangular openings attached to a ring slot [1] (Fig. 2.3(c));

• switching the orientation of an “L”-shaped arc slot near the edge of a circular

patch [18];

• switching the linear polarization of the aperture coupling for feeding a perturbed

circular patch [19]; and

• switching between a pair of crossed slots on a square patch [2] (Fig. 2.3(d)).

The list above describes how CP is generated for each reconfigurable antenna. It

also illustrates that the realization of reconfiguration lies in switching for majority of

the antennas with switchable polarization found in literature. However, other mech-

anisms besides switching have been reported, as demonstrated in [20] with changing

the excitation scheme of a substrate integrated composite right-/left-handed leaky-

wave structure. Also, antennas capable of both switchable polarization and frequency

agility (i.e. reconfigurable state of polarization and operating frequency) have been

published [21]. The focus of this thesis is on antennas with switchable polarization.

14

In the next section, various means of achieving reconfiguration will be introduced.

(a) (b)

(c) (d)

PIN diode

Shorting

capacitor

Biasing

slit

reconfigurablefeedline

foam substrate

FR-4 substrate

PIN diodeShorting

capacitor

Slot

Figure 2.3: Various designs of printed antennas with switchable polarization [1]c© 2003 IEEE, [2] c© 2002 IEEE, [3] c© 2004 IEEE, [15] c© 2008 IEEE.

2.2 Means of Reconfiguration

As mentioned in Section 2.1.3, CP antennas with a switchable sense of polarization

have seen a prevalent use of RF MEMS switches and PIN diodes as their switching

15

elements. Switching between the sense of CP can be performed simply by adjusting

the biasing voltages to these switching elements, essentially turning on and off the

various elements. This section compares these existing technologies, as well as some

of the more novel methods for creating reconfigurable RF devices.

2.2.1 DC-Biased Switches

The performance requirements of RF switches continue to become more stringent,

with many figures of merit being considered, such as insertion loss, isolation, power

handling, repeatability, switching speed, and ease of integration [22],[23]. Of the vari-

ety of existing switches in the market, PIN diodes, field-effect transistors (FETs), and

MEMS are amongst the most popular choices of RF designers [23],[24]. Though solid-

state switches — namely PIN diodes and FETs — boast fast switching speed in the

range of ns, which is important for telecommunications applications, they both suffer

from RF modulation, distortion, and noise. There is usually a trade-off between good

power handling capability and reasonable loss characteristics. For instance, Aeroflex

Metelics4 offers a range of PIN diodes, one with insertion loss of 0.25 dB at 6 GHz

and maximum power handling of only 10 W, which contrasts with another one capa-

ble of handling 150 W at the expense of higher insertion loss of 0.4 dB. RF MEMS

switches, on the other hand, have demonstrated their superior low-loss characteristics

(insertion loss < 0.1 dB up to 40 GHz) and immunity from RF distortion. However,

their switching speed is slower (µs) and they generally require higher actuation volt-

ages compared to solid-state switches. The power handling of MEMS devices is also

limited. It is obvious that the selection of the type of switches depends on the spe-

4www.aeroflex.com/ams/metelics/micro-metelics-prods-mhp-series-switches.cfm

16

cific applications and performance requirements. Note that varactor diodes, which

are voltage-controlled variable capacitors, may also be used as switching elements.

2.2.2 Liquid Metals

While research on MEMS switches is heading in the direction of smaller size and

greater efficiency [25], due in no small part to their usage in increasingly elaborate

circuit designs which require complex biasing networks, research on using unconven-

tional materials such as liquid metals has also received some attention. The use of

liquid metals such as mercury and Galinstan in tunable RF circuits as an alterna-

tive to traditional PIN diodes and MEMS switches has been demonstrated [26],[27],

as shown in Fig. 2.4. Varying the positions of the liquid metal droplets leads to a

change in the reactive loading of the RF circuitry, and can be utilized for reconfig-

urable antenna designs. A frequency reconfigurable antenna using a mercury droplet

has been demonstrated in [28]. By varying the voltage of electrostatic actuation, the

electrowetting profile of the droplet on the dielectric platform was altered, which gave

rise to a change in the capacitive loading and subsequently, a change in the resonant

frequency of the antenna. Such systems of dynamic control boast greater power han-

dling capability and better surface contact. However, the use of a toxic metal such

as mercury is not desirable, and relatively high voltages are often required to achieve

reasonable tuning. In addition, residue can be left behind unless the liquid metal

is bounded by Teflon coating or solution, and controlling the precise position of the

liquid metal in relation to the RF circuitry can be difficult unless binary control is

implemented. Also, using liquid metals introduces loss at high frequencies. Galin-

stan, for instance, has an electrical conductivity of 3.46 · 106 S/m, which is one order

17

of magnitude lower than that of copper [29].

(a) (b)

Figure 2.4: Examples of using liquid metals in RF devices [26] c© 2010 IEEE, [27]c© 2007 IEEE.

2.2.3 Permittivity Manipulation Techniques

As shown in Fig. 2.2, a microstrip antenna in its simplest form consists of a patch

(conductor), a substrate (dielectric), and a ground plane (conductor). RF switching

elements, as described in Section 2.2.1, manipulate the conducting layers. Another

way to create reconfiguration is to manipulate the dielectric substrate. Methods

utilizing materials such as ferroelectric thin films, liquid crystals, and nanoparticle-

oil mixture will be discussed next.

Phase shifters with ferroelectric thin films are based on varying the dielectric

constant of a thin film when it is subjected to a DC electric field. These thin-film

phase shifters were claimed to be capable of 360 of phase shift by sweeping the DC

bias voltage from 0 to 350 V to create a change in dielectric constant from 2000 to 800

[5, ch.21],[30]. However, this technology suffers from high dielectric loss and requires

high DC voltages. Maintaining the quality of the pulse laser ablated BaxSr1−xTiO3

films (one example of ferroelectric thin films) beyond a thickness of 0.5 µm is a difficult

18

challenge [5, ch.21]. The dielectric constant of the film is also subject to a significant

variation with changing temperature as opposed to bulk materials.

Similar to the ferroelectric thin-film technology, dynamic phasing with liquid crys-

tals is also a result of altering dielectric constant. The liquid crystal molecules are

pre-aligned such that they are perpendicular to the RF-field, resulting in a particular

effective dielectric constant of ǫ⊥. The molecules are rotated in a continuous fash-

ion under an applied DC voltage until reaching saturation, at which the state of the

molecules is parallel to the RF-field, producing the effective dielectric constant of ǫ‖.

Therefore, a range of effective dielectric constant between ǫ⊥ and ǫ‖ can be realized.

The liquid crystals in [31] has the following properties: ǫ⊥ = 2.39, ǫ‖ = 3.18,

tanδ⊥ = 0.0061, and tanδ‖ = 0.0016. By applying 0 to 40 V to the phase shifter

containing the liquid crystal mixture, the maximum phase shift was 117 with a

maximum loss of 4.03 dB. In order to realize a reasonable tuning range, it can be

seen that the liquid crystal technology also suffers from high dielectric loss. However,

it has been shown that the loss tangents of some nematic liquid crystals improve with

increasing frequency (measured from 30 to 60 GHz), which provides greater incentive

for using liquid crystals in high-frequency applications [32].

One common trait shared by MEMS switches, PIN diodes, FETs, ferroelectric

thin films, and liquid crystals is their voltage-controlled actuation, which is generally

a precise and repeatable actuation mechanism, although relatively high voltages may

be required. However, as the number of switches increases with more complex designs,

so does the complexity of the biasing layout on the conducting layer(s), which results

in limitations imposed on the RF functionality. This is exemplified by the presence

of four diagonal slits on the ground plane in Fig. 2.3(c), which were necessary for

19

biasing the four PIN diodes. Biasing slits and feedlines, depending on their locations

and the materials used, can interact with the fields of the radiating sections of an

antenna, which may lead to deterioration in the antenna patterns as well as losses.

Also, undesired slot modes can be excited in the biasing slits. Suppression of these

modes requires placing more shorting capacitors along the biasing slits, which is at

the expense of increased complexity and losses. Research on switches using other

actuation mechanisms, such as magnetostatic, thermal, and piezoelectric, has also

received attention [22],[33]–[35].

As an alternative to these voltage-controlled schemes, fluidic control offers a com-

pletely different approach. A coaxial stub microfluidic impedance transformer, also

called COSMIX, was described in [36], with potential applications in reflectarray an-

tennas [37]. A mixture of colloidal barium strontium titanate (BST) and silicone

oil in the coaxial cell is the crucial component of the stub loading, as illustrated in

Fig. 2.5. By varying the volume fraction (ϑ) of BST nanoparticles in the mixture, the

relative permittivity of the mixture can span from 2 to 8 for ϑ of 0 to 0.5; a change

in impedance loading is achieved. The mixture is injected into and out of the coaxial

cell using fluidic control (miniature pumps and valves), a method shared by some of

liquid metal technologies mentioned in Section 2.2.1.

However, a few drawbacks are apparent with this method. A dedicated mixing

component is needed, which increases the size of the device. For every change in

permittivity, the entire coaxial cell has to be completely emptied and replaced with

a new mixture, which seriously compromises the agility of this scheme. The repro-

ducibility of permittivity values is also questionable as the system relies heavily on

thorough dispersion of the nanoparticles in the oil and their precise proportion.

20

Figure 2.5: Reconfigurable reflectarray design using COSMIX [37] c© 2010 IEEE.

2.2.4 Flexible Materials

The development of flexible and wearable electronics has greatly promoted the explo-

ration of using flexible materials in designing reconfigurable antennas [38]–[43]. One

example is the use of thin films of solid metal deposited on a flexible dielectric ma-

terial, as described in [43] and [44], in which microscopic movement of the metalized

membrane was induced by pneumatic techniques. Fig. 2.6 illustrates a switch pro-

totype based on this technology. The movement of the membrane, fabricated from

polydimethylsiloxane (PDMS), is analogous to the movement of the metal bridge

of a MEMS switch. Since the metalized membrane is pneumatically controlled, DC

biasing feedlines are not required and therefore, eliminating the disadvantages of com-

plex biasing structures and the interference with the RF operation associated with

DC-controlled devices.

PDMS, which has become a popular choice as the flexible dielectric material be-

cause of its ease of processing, commercial availability, and flexibility at ambient

temperatures, has a relatively high loss tangent (tanδ in the range of 0.01 to 0.045

measured from 0.1 to 40 GHz [45]). Using this material would lead to higher losses

21

Figure 2.6: Switch design based on pneumatically actuated membranes [44] c© 2012IEEE.

in RF devices and reduced efficiencies of high-Q factor antennas (such as microstrip

patch antennas) [41]. Also, due to the inherent nature of physical movement of the

membrane, microcracks were observed in the thin metal film, which would undermine

the long-term reliability of the fabricated devices, unless the surface of the film was

prepared as an ordered wave to enhance its stretchability [42],[46].

Recent research has demonstrated the alternative of using liquid metals enclosed

in PDMS to create flexible electronics, including antennas [40]–[42]. The use of liquid

metals instead of solid metals allows these reconfigurable antennas to undergo drastic

shape deformation that is macroscopically visible without suffering from issues asso-

ciated with microcracks. Tuning of the resonant frequency of a dipole antenna by

stretching [40] and varying the impedance matching of a patch antenna via flexing

(Fig. 2.7) [41] were demonstrated. The disadvantage of using PDMS is evident in the

22

poor radiation efficiencies of 60% measured for the patch antenna in [41]. Creating

a large conductive area such as the patch or the ground plane using a liquid metal

in PDMS is challenging because directly filling channels that have low aspect ratios

may lead to the collapse of the channels and the uniformity of filling the channels

can be undermined by air pockets and uneven channel height. Sectoring the patch

into a long serpentine channel with rows of PDMS posts has been shown to mitigate

some of the issues in channel filling, but this was realized at the expense of increased

complexity of the structure and the time required to fill the channel [41].

Patch Ground

plane

Figure 2.7: Flexible patch antenna [41] c© 2012 IEEE. The embedded liquid metalforms two conductive layers (patch and ground plane) within the flexible PDMS.

The review of the previously reported means of reconfiguration presented above

reinforces the idea that there is no single means that is ideal for all applications, and

that the selection of a technology for a particular end-use application is driven by the

prioritization of various trade-offs. This review allows for comparison with the novel

switching mechanism proposed in this project, described in detail in Chapter 3.

23

2.3 Selecting a Circularly Polarized Antenna

Section 2.1.3 provided the motivation for developing a reconfigurable CP antenna

with switchable polarization. This antenna also serves the purpose of demonstrat-

ing the viability of the switching mechanism introduced in this thesis, which can be

applied to other reconfigurable antenna designs beyond switchable polarization. In

order to design a functional CP printed antenna with switchable polarization, se-

lecting a suitable antenna type is critical. CP can be achieved with various types

of antennas, some of which were mentioned in Section 2.1.3. Each one has its own

merits and drawbacks. Some prominent examples of CP printed antennas are listed

in Table 2.1 [5],[13],[14],[16],[47]–[51]. Note that the list is by no means exhaustive;

many variations and different types of antennas can be found in literature.

From Table 2.1, it is obvious that no single antenna type can be deemed the

perfect candidate, which reflects the reality of trade-offs in antenna designs. There

are some complementary merits and drawbacks. For instance, hybrid power dividers

or quadrature-phase feeds are not required for single-fed antennas, but identifying the

operating frequency of these antennas is not as straightforward as in the case of their

linearly-polarized versions. Achieving reasonable AR and impedance bandwidths is

important when designing CP antennas, as it increases the antennas’ tolerance to

manufacturing defects and other specification deviations. Many of the antenna types

in Table 2.1 use different techniques to achieve moderate bandwidths at the expense

of increased complexity. Some of the published antenna designs obtained respectable

bandwidths by employing a thick substrate and/or an air layer, thus increasing the

size of the antenna.

Some of the antenna types can be made reconfigurable more easily than others.

24

Table 2.1: Merits and drawbacks of various types of circularly polarized antennas.M

erits

(M)&

Drawback

s(D

)

Quad

rature-phasefeed

not

necessary

Sim

ple

stru

cture

Suitable

forpro

posed

mech

anism

Smallfootprint

Moderateto

largeban

dwidth

Operatingfreq.easily

determined

Narrow

ban

dwidth

Operatingfreq.not

easily

determined

Large

footprint

Trickyto

excite

correctmode

Com

plicatedforproposed

mechan

ism

Thickerstructure

Cou

plingdep

endenton

substrate

heigh

t

Rad

iation

onbothsides

Low

gain

1 M M M D D2 M M M M D D3 M M M D D D4 M M D D D5 M M M D D D6 M M D7 M M M M D D8 M M D D D9 M M D10 M D D

Antenna Types1: Single-fed square and circular patches with perturbations (see Fig. 2.3(b))2: Single-fed rings in TM11 mode with perturbations (see Fig. 2.3(a))3: Single-fed annular rings in TM12 mode with perturbations4: Single-fed rings with perturbations coupled to parasitic elements5: Single-fed stacked patches and rings with perturbations6: Single-fed printed loop with reactive loading7: Single-fed ring slots with perturbations (see Fig. 2.3(c))8: Printed spiral antennas9: Quadrature-phase-fed crossed-dipoles10: Cavity-backed crossed bowtie dipoles

25

For the purpose of selecting an antenna type compatible with the proposed switching

mechanism, types #1, #2, #3, #5, and #7 are shortlisted. Considering the fact that

a novel means of reconfiguration with custom fabrication technology is involved, a

prudent approach is to implement it on a simple structure, which narrows the choices

to a single-fed square or circular patch with perturbations (type #1), a single-fed

ring in TM11 mode with perturbations (type #2), and a single-fed ring slot with

perturbations (type #7). The last choice proves to be the most suitable because slot

antennas generally have greater impedance bandwidth compared to patch antennas

[52], which will be advantageous for this project. Perturbations can be in the form

of openings [1] or stubs [53], which are attached at specific positions to the outer

perimeter of the ring slot. The inherent nature of a slot antenna to radiate on both

sides is sometimes considered a drawback, but it depends on the applications.

With the ring slot chosen as the topology for realizing a CP antenna with switch-

able polarization, the next section will elaborate on the theory behind the generation

of CP specifically for this type of antenna.

2.4 Realizing Circular Polarization with Ring Slot Antenna

The basis for realizing circular polarization for a single-fed ring slot antenna, as

demonstrated in [1], is the perturbation of the two diagonal linear modes caused by

loading the ring slot with the appropriate pair of rectangular openings, as shown

in Fig. 2.8. Each opening is connected to the ring slot with a short narrow section

of slotline. The openings serve the purpose of lengthening the path of the current

flowing around the ring, and hence lowering the resonant frequency of the antenna.

To gain a better understanding of this idea of perturbation of linear modes, it

26

rectangular

opening

ring slot

pin diode across

a narrow slotline

Figure 2.8: Ring slot antenna with switchable polarization using PIN diodes [1]c© 2003 IEEE.

is important to have a closer look at the field distribution on the simple ring slot

antenna, shown in Fig. 2.9. Assuming the antenna is in transmit mode, power is

transferred from the microstrip line on the bottom side of the substrate to the slotline

feed on the ground plane. Radial stubs are used, which is a common practice to

realize a broadband microstrip-to-slotline transition [54]. From Fig. 2.9(a), it can

be seen that the location of the slotline feed dictates the direction of the overall

LP field being radiated from the ring slot antenna. In this particular case, the y-

oriented slotline feed leads to an x-polarized antenna (i.e. radiated field is x-directed),

as indicated in Fig. 2.9(b). This LP field can be decomposed into two orthogonal

diagonal components which are represented by the dashed arrows. The importance

of this field decomposition will become apparent below.

The positions of the rectangular openings with respect to the polarization direction

of the LP field are crucial. Fig. 2.10 shows the ring slot in linear polarization, with

an opening attached to the ring at different location. In Fig. 2.10(a), the rectangular

27

(a)

y

x

(b)

Slotline

feed

Microstrip feed Ring slot

Figure 2.9: Field distribution on the ring slot antenna. (a) Magnified view of theelectric field distribution at the transition from the slotline feed to the ring slot.(b) Decomposition of the overall x-directed LP field (solid black arrow) into twoorthogonal diagonal components (dashed arrows).

opening is placed on the north side of the ring, where current is the weakest. The

perturbation from the opening on the current is minimal in this case and the antenna

experiences a very minor shift in its resonant frequency. When the opening is placed

on the east side of the ring where current is the strongest, as shown in Fig. 2.10(b),

the current sees a longer path around the ring, which has the effect of lowering the

resonant frequency of the antenna.

This decrease in resonant frequency due to perturbation translates to a phase

lag in the radiated field at a particular frequency, as illustrated in Fig. 2.11. The

general phase response of the radiated field of an antenna near resonance is shown

in Fig. 2.11. The point of zero phase, where the curve crosses the frequency axis,

indicates the resonant frequency of the antenna. Loading the ring slot antenna with an

appropriately placed opening decreases the resonant frequency, which is represented

28

x

Rectangular

opening

(a)

y

(b)

Figure 2.10: Comparison of perturbation on the ring slot current for different positionof the rectangular opening. The black arrow represents the direction of the linearpolarization while the red dashed arrows represents the current flow around the ringslot.

by a shift of the entire phase curve to the left (solid curve to dashed curve). At a

particular operating frequency, the phase of the radiated field from the perturbed

antenna is delayed compared to that from the unperturbed antenna.

Consider the scenarios when a pair of openings are aligned 45 from the direction

of linear polarization, as shown in Fig. 2.12. For the sake of intuitive understanding,

first ignore the presence of the openings. As mentioned earlier, the overall radiated

field can be decomposed into two orthogonal diagonal components. In a similar sense,

the current distribution around the ring slot can also be viewed as the combination of

the current contributions that generate the two orthogonal field components. For both

scenarios in Fig. 2.12, the locations of the openings coincide with the points of current

minima of the component labelled “0”. Consequently, the “0” component in both

scenarios sees minimal influence from the rectangular openings, whereas the “−90”

component has a lower resonant frequency and therefore lags the “0” component

29

FrequencyPh

ase

Operating

Frequency

Phase

Difference

unperturbed

perturbed

Figure 2.11: Illustration of phase lag in the radiated field due to the lowering ofresonant frequency of an antenna. The solid curve represents the phase response ofthe antenna without perturbation, whereas the dashed curve represents that withperturbation.

in phase. With some adjustments to the parameters of the antenna, such as the

ring radius, the length of the connecting slotline, and the size of the opening, the

two orthogonal components (modes) can be made 90 apart with equal magnitude.

Under this condition, circular polarization (CP) is achieved and the type of CP (left-

or right-handed) is determined by the phase relation between the two components, as

shown Fig. 2.12. Unlike a microstrip antenna, an ideal slot antenna radiates almost

equal amount of power on both sides of the antenna, as mentioned in Section 2.1.3.

Radiation leaving from the top side of the antenna has a reverse CP compared to the

radiation leaving from the bottom side of the antenna. The convention used in this

thesis for referring to the polarization of the ring slot antenna is that of the radiation

leaving from the top side of the antenna (wave traveling in the +z-direction away

from the antenna). That is, an RHCP ring slot antenna is referring to an antenna

that radiates RHCP wave in the +z-direction and LHCP wave in the –z-direction.

If the two pairs of openings are simultaneously present in the antenna structure,

30

-90° 0° 0° -90°

LHCP RHCP

y

x

(a) (b)

Figure 2.12: Ring slot antenna operating in (a) LHCP and (b) RHCP states forradiation leaving the antenna in the +z-direction (out of the page). The polarizationis reversed in each case for radiation leaving the antenna in the –z-direction.

the antenna will generate an LP field in the x-direction, with a lower resonant fre-

quency compared to the case when none is present. The reconfiguration in polariza-

tion arises from selectively allowing the current to bypass the appropriate rectangular

openings. Referring to Fig. 2.8(a), the pair of openings are bypassed when the PIN

diodes connected across the narrow slotline are forward biased (behaving as shorted

switches). From the RF perspective, the pair of openings are not present and the

current flows around the ring slot to generate a linearly x-polarized radiation. When

the diodes are in the “off” state (reverse biased), the current of one of the diagonal

linear modes has to flow around the openings, leading to a phase lag compared to

the other diagonal mode. Given the locations of the openings with respect to the mi-

crostrip feed, the resulting CP is left-handed. For the antenna shown in Fig. 2.8(b),

with diodes 1 & 3 on and 2 & 4 off, LHCP radiation is generated, traveling in the

+z-direction. When diodes 1 & 3 are off and 2 & 4 are on, RHCP radiation traveling

in the +z-direction is produced.

31

Note that the two reconfigurable antenna designs (Fig. 2.8) in [1] could each

realize two polarization states (linear/LHCP and LHCP/RHCP), but not all three

polarization states in one antenna. This is because of the difficult challenge of arriving

at a combination of dimensions that sees the impedance bandwidths for all three

polarization states coinciding in the same range of frequency while maintaining a

reasonable AR bandwidth also within the same range. Generally, the AR bandwidth

of a circularly polarized antenna is noticeably smaller than its impedance bandwidth.

The performances of the reconfigurable ring slot antennas (LHCP/RHCP only)

in [1] and in [53], as well as two other similar reconfigurable antennas, are listed

in Table 2.2. All four antennas used PIN diodes as the switching elements. They

also operated in the same frequency range around 2.4 GHz, which falls in one of the

industrial, scientific, and medical equipment (ISM) bands [55]. They will serve as

the performance benchmarks for the antenna fabricated in this project, described in

Chapter 6. It should be noted that an obvious area of commercial applications for

switchable polarization is in GPS, which currently operates in the L1 (1.57542 GHz)

and L2 (1.2276 GHz) bands [5, ch.2]. However, the proposed antenna will be devel-

oped for operation in the 2.4 GHz ISM band for comparison with existing works.

Antenna designs published in [18] and [19] were capable of three or more polariza-

tion states (LP/RHCP/LHCP and LP/LP(orthogonal)/RHCP/LHCP, respectively),

which is impressive. However, such a feat came with some trade-offs. In [18], the over-

all three-state bandwidth — the strictest definition of bandwidth for antennas with

switchable polarization, which is defined as the bandwidth within which a reconfig-

urable antenna achieves both |S11| < −10 dB for LP/LHCP/RHCP and AR < 3 dB

for LHCP/RHCP — was actually zero. This means that there was no range of fre-

32

quency within which the antenna could truly achieve all three polarization states. In

order to achieve a three-state bandwidth of 2.2%, the thickness of the antenna in [19]

was significantly greater than the other antenna designs, which made it noticeably

bulkier. Also, both designs had complicated matching topologies.

Table 2.2: Performance of notable antennas in literature with switchable polarizationin the 2.4 GHz range.

Criteria [1] [53] [18] [19]

# of pol. states 2 2 3 4Thickness5 0.8 mm 1.6 mm 1.6 mm 8.2 mm|S11| BW 37.5% 20.9% 1.17% 13.1%

AR BW (LH) 4.3% 4.2% 0.74% 4.5%AR BW (RH) 3.4% 4.1% 0.72% 2.2%Min. AR (LH) 0.5 dB 1.0 dB 0.9 dB 1.0 dBMin. AR (RH) 1.2 dB 1.25 dB 0.5 dB 0.8 dBOperating Freq. 2.38 GHz 2.63 GHz 2.45 GHz 2.45 GHzBroadside Gain 4 dBi 3.1 dBi 5.85 dBi 5.87 dBi3-state BW N/A N/A 0% 2.2%

5Thickness accounts for the dielectric layers of the antenna structure and excludes

support structures such as measurement holders.

With the principles of operation of the reconfigurable ring slot antenna explained,

Chapter 3 introduces the novel means of pneumatically controlled reconfiguration,

while Chapter 5 describes in detail how this switching mechanism can be incorporated

into the design of a ring slot antenna.

33

Chapter 3

Pneumatically Controlled

Switching Mechanism

The concept of a pneumatically controlled switching mechanism, which is a major

focus of this thesis, shares one similarity with the fluidic control described in Sec-

tion 2.2.3 — the need of generating a pressure gradient in the fluid inside channels.

The key differences are the medium and the object being transported. The medium in

the proposed mechanism is air rather than a liquid, and the object being transported

is a solid slug instead of a liquid. The advantage of moving a solid slug instead of a

liquid metal or high-permittivity liquid mixture is the precise position control, since

a solid slug does not change shape or deform during its movement within the channel

like liquids do. The issue of having residue being left behind on the inner surface of

the channel after the displacement of the liquid matter is completely avoided in the

mechanism introduced here.

In this chapter, the concept of the basic switching mechanism is described. The

reasons behind the selection of the process and the material for constructing sample

channels are given, along with the overview of the process steps and the results of the

mechanical testing of the switching mechanism.

34

3.1 Concept

The switching mechanism, in its basic form, is illustrated in Fig. 3.1. RF reconfigu-

ration is a result of the capacitance variation in the region where a gap in the metal

layer is present. Initially, a series capacitance is associated with the gap, as shown

in Fig. 3.1(a). When a pressure gradient is generated by a stream of pressurized

air entering through one end of the channel, the dielectric slug is transported to the

other end of the channel in the direction of air flow. The slug width and thickness are

slightly less than those of the enclosing chamber, such that air flow is not completely

impeded and the movement of the slug is guided. As the slug rests on top of the gap,

as shown in Fig. 3.1(b), the metal coating on the underside of the slug bridges the

two halves of the metal layer. A thin layer of insulating material deposited on top of

the slug’s metal coating ensures a capacitive contact between the slug and the metal

layer instead of a metal-to-metal contact. The series capacitance across the gap is

now supplemented with additional capacitance, and this capacitance enhancement is

the basis of the RF reconfiguration exemplified by the devices presented in this thesis

(a switch in Chapter 4 and an antenna in Chapter 5).

The system for creating the pressure gradient and its integration in the fabri-

cated devices is beyond the scope of this thesis, although using miniature pumps,

which are a well-established technology in the field of microfluidics, is a promising

method. A preliminary study is presented in Appendix B. The minimum actuation

pressure required to move the slug will be investigated with a fabricated RF switch

in Section 4.4.

In order for this switching mechanism to be deemed viable for reconfigurable

antennas, its performance in both mechanical and RF criteria must be tested. Since

35

Substrate

Cover

Metal-plated slug Metal layer

Air

Channel wallGap

Air

Bottom View

Side View

(a) (b)

Figure 3.1: Pneumatic control of the slug position and the resulting capacitancevariation. Channel wall is not shown in the side view.

the mechanical criterion, i.e. the ability to move the slug back and forth within

a channel, can be evaluated independently of the RF criterion, the focus of this

chapter is on developing a reliable and repeatable process for fabricating channels

and on the mechanical testing of the switching mechanism. The understanding of

fabrication tolerances and constraints will be invaluable for designing the capacitive

switch (Chapter 4) and the antenna (Chapter 5).

The advantages and disadvantages of this switching mechanism will be discussed

in Section 4.3, after the RF performance of the fabricated switch is evaluated.

36

3.2 Process and Material Selection for Channel Fabrication

The channel is a layer of patterned polymer sandwiched between a cover and a met-

alized substrate. The channel widens as it meets the chamber within which the slug

is confined, as seen in Fig. 3.1. With the goal of testing only the mechanical aspect of

the switching mechanism in mind, it would be wise to begin with a simplified process

involving only the necessary layers to construct the physical channels. This would

facilitate troubleshooting and reduce the number of iterations required to realize a

dependable process. Layers related to the RF aspect, namely the metal layer on the

substrate and the metal coating on the slug, are hence omitted here.

In order to take advantage of the access to the Advanced Micro/nanosystems Inte-

gration Facility (AMIF) at the University of Calgary and the experience and training

on the equipment gained by the author in previous projects, the fabrication process

and material selection are tailored towards those of MEMS. Much of the equipment

in AMIF is compatible with MEMS production and specifically, the photolithography

process.

The choice to fabricate the channels in-house rather than submitting designs to

third-party fabrication facilities arises from the fact that the channel structure is not

compatible with any standardized fabrication technologies such as complementary

metal-oxide-semiconductor (CMOS) and silicon-on-insulator (SOI). The benefit of

developing a custom process in an on-campus fabrication laboratory is the flexibility

of modifying the process parameters and procedure, and the relatively short iterative

fabrication cycles in refining the process steps. However, the iterative nature of

arriving to a practical and repeatable process presents a surmountable yet difficult

challenge because of the substantial amount of time and varying factors associated

37

with developing a custom process.

The photolithography process is chosen for the fabrication of the channels because

it offers the flexibility of thickness variation, which is beneficial during the RF de-

sign and for the future consideration of miniaturizing a device. In a typical MEMS

photolithography process, a photoresist is deposited on a substrate, exposed to ultra-

violet (UV) light under a photomask, and developed to leave behind some patterned

features on the substrate. The patterned features, composed of solidified photoresist,

can either be a temporary fixture used for etching the pattern into the substrate or a

permanent fixture as part of a complete device. The thickness of a given photoresist

on the substrate can be varied within a finite range by adjusting the spin speed during

its deposition. Most of the commercially available photoresists stocked in AMIF are

capable of thicknesses between 0.5 µm and 250 µm with a single deposition. The

photolithography process, for a particular photoresist, is described in greater detail

in Section 3.3.

The channels should be constructed with a low-loss, low-permittivity dielectric,

such that their influence on the RF performance and the field distribution can be kept

at a minimum. In order to fabricate channels with a thickness between 150 µm and

250 µm, which is the lower limit of the slug thickness at which its structural integrity

remains acceptable, the ideal choice of photoresist is SU8-2100. Slugs thinner than

150 µm are deemed too fragile to handle. The datasheet of SU8-2100 suggests the

possibility of realizing 250 µm in one deposition. Thinner photoresists can be used

to attain similar thickness if consecutive stacking depositions are performed, but this

method of multiple depositions would compromise the repeatability in the thickness

achieved and complicate the overall process. SU8-2100 is selected also because it is

38

an epoxy, which is chemically and mechanically inert after processing. This makes it

an ideal candidate for creating features on a device.

However, SU8-2100 does suffer from two major drawbacks that need to be ad-

dressed before it becomes part of a reliable and repeatable fabrication process for

any device. The first disadvantage is the difficulty in its handling. SU8-2100 is an

extremely viscous photoresist, which requires special pouring technique during its

deposition. The second issue with the photoresist is its inconsistent adhesion to the

substrate due to the buildup of internal stress during its processing. Strict compliance

to an established set of process steps would generally increase the yield of devices.

Patience and careful observation, along with troubleshooting skills gained only from

experience and repeated trials, are key to the successful application of SU8-2100.

As a result, the channels described in this chapter serve two important purposes

— testing the feasibility of moving slugs confined inside channels of different config-

urations, and refining the fabrication steps of processing the SU8 photoresist.

3.3 Processing SU8 Photoresist

As mentioned in Section 3.2, SU8 is one of a few highly viscous and commercially

available photoresists that can be used to attain chemically inert structures with a

thickness of 200 µm. However, because of its viscosity, it is also very difficult to

work with. Detailed steps for processing the photoresist in AMIF, customized by the

author, can be found in Appendix C. In this section, an overview of the process is

described, with the introduction of a photomask designed specifically to produced

channels for mechanical testing of the switching mechanism.

39

Processing SU8 photoresist (in a controlled cleanroom environment with yellow

lighting) generally involves:

1. Cleaning a 4” circular wafer with a Piranha etching solution.

2. Exposing to hexamethyldisilazane (HMDS) in a vacuum oven, which will pro-

mote the adhesion between the photoresist and the wafer.

3. Pouring an appropriate amount of the photoresist at the centre of the wafer.

This is one of the most critical steps of the entire process which requires practice.

4. Spreading the photoresist evenly at a targeted thickness by spinning the wafer

at a set speed in a spin coater.

5. Soft-baking the deposited photoresist on a programmable hot plate to evaporate

the majority of the solvent in the photoresist gradually. Drastic temperature

changes would cause thermal stress in the photoresist, which leads to adhesion

issues.

6. Exposing the photoresist to UV light under a photomask using the mask aligner,

for a predetermined duration corresponding to the targeted thickness.

7. Post-exposure baking the photoresist on the programmable hot plate gradually

helps facilitate the cross-linking of the photoresist in the exposed area, which

becomes insoluble in the developer solution.

8. Developing the photoresist in an agitated bath of SU8 developer solution. Unex-

posed photoresist is dissolved in the solution, leaving behind patterned features

identical to those on the photomask.

SU8 is a negative photoresist, which means that any UV-exposed area of the

photoresist remains on the wafer. Therefore, a photomask designed for a negative

photoresist (identified as non-inverted/dark) is mainly opaque. A screen-shot of the

40

photomask design used to fabricate the sample channels is shown in Fig. 3.2. The

design configurations used are discussed in detail in Section 3.4. The photomask was

drawn using Cadence’s Virtuoso, a layout editing tool, and the mask was fabricated

by the Nanofab at the University of Alberta (precision within 2 µm). The dark area

shows up as opaque on the square-shaped photomask which is made of a 0.09” thick

glass plate. The areas shown as red are clear on the photomask, so allowing UV

light to pass through during exposure. Having channel structures populating most

of the wafer surface also serves the purpose of providing structural support when the

cover layer (a glass wafer) is stacked on top of the channels after the photoresist is

patterned.

Various issues were encountered while refining the photolithography process, and

after numerous iterations, most were either eliminated or kept under control:

• Any bubbles trapped in the photoresist during the pour led to localized un-

evenness. Unevenness causes adhesion issue and inconsistent thickness of the

patterned features. Practice-pouring the photoresist was key to refining the

technique.

• Even with an evenly spread photoresist, unevenness was observed during the

soft bake. The cause was determined to be the slight uneven heat distribution

on the wafer. The problem was mostly alleviated with placing a thin piece

of aluminum plate between the hot plate and the wafer to facilitate the heat

transfer, as well as removing as much resist removal solution residue as possible

from the bottom side of the wafer.

• Stress marks around the exposed area were observed during the post-exposure

bake. The stress marks caused localized unevenness, as well as issues with

41

Figure 3.2: Computer screen-shot of the photomask design used to build the testchannels.

completely removing unexposed photoresist. White residue was left despite

extended period of submersion in the developer solution. Two possible causes

for the stress marks are the density of features on the photomask and the

temperature ramp rate. Instead of designing and ordering a new mask, the

temperature ramp rates for both the heating and cooling steps were lowered by

about 15%, which resolved this issue in most trials.

• Hard baking, which is a recommended step after developing when the photore-

sist is used for creating permanent fixtures, is supposed to further strengthen

the photoresist. However, the high temperature (165C) required caused sig-

nificant amount of thermal stress in the patterned photoresist, which led to a

42

complete delamination and curling of the features, as shown in Fig. 3.3. Hard

baking was not performed for any subsequent fabrication trials.

• Adhesion issue, due to the aforementioned reasons as well as causes unknown,

was largely eliminated but still appeared in isolated areas on a small number of

processed wafers.

Good Poor adhesion

Stress-induced delamination

and curling

Complete detachment with

portion of silicon ripped

from wafer surface

Good

adhesionPoor adhesion

in most area

Figure 3.3: Adhesion issue in the development of the SU8 process.

A successfully fabricated wafer with good SU8 thickness uniformity (200± 30µm)

is shown in Fig. 3.4. In this particular instance, SU8 was processed on the rough side

of a silicon wafer to improve the chance of achieving good adhesion. The wafer was

diced into sections and the pneumatically controlled switching mechanism was tested

for various channel configurations, as described in Section 3.4.

43

Figure 3.4: SU8 resist successfully deposited and developed with minimal adhesionissue.

3.4 Channel Configurations

With the channels built on a wafer, slugs were required to complete the simple testing

structures. Cover slips of thickness #1.5 (0.16 to 0.19 mm) were diced into slugs of

various lengths and widths. In general, the slugs were diced such that they were

slightly narrower (80 µm) than the chambers inside which they would be placed.

This provided an average clearance of 40 µm on each side of the slug, which was

deemed large enough for unhindered linear movement of the slug inside the chamber

and small enough to prevent any rotational movement of the slug. For instance, a

chamber that is 1 mm wide is fitted with a 0.92-mm-wide slug. Subsequently in this

section, a slug with a listed width of 1 mm is in fact 0.92 mm wide; a listed chamber

width of 2 mm equates to a slug width of 1.92 mm.

The actual thickness of the few cover slips used fell in the range of 0.175 to 0.18 mm

44

consistently , which was quite ideal for the SU8 channel thickness of 200 µm, offering

about 20 µm of clearance.

As mentioned in Section 3.2, the slugs were not coated with a metal layer as the

thin coating was not expected to interfere with the testing of the slug movement. The

difference in friction characteristics was not a significant concern since the slug was

expected to be lifted slightly by the in-rush of pressurized air causing its movement

inside the chamber. Though outside of the scope of this work, a fluid dynamics based

simulation could be performed for comparison.

3.4.1 Simple Channels

A section on the wafer was dedicated to testing simple channels and determining

the range of channel dimensions with which good SU8 adhesion could be attained. A

picture of the setup is shown in Fig. 3.5. A circular feature terminated each end of the

channels. These 3-mm-diameter features identified the locations where holes would

be drilled in the wafer to provide access for air flow from the backside of the wafer.

The slugs would be placed in the chambers and a second wafer would be adhered on

top of the channels to complete a sealed prototype. In order to minimize the number

of fabrication steps involved during this preliminary test stage, some modifications

were made that would simplify fabrication, yet still allow the mechanical testing of

the switching mechanism. Hole drilling was not performed as the forces involved

had a tendency to crack the wafer. A microscope slide was used to substitute the

covering wafer, also shown in Fig. 3.5, such that the area directly above the chambers

was still covered while leaving the ends of the channel uncovered to allow air flow.

Instead of permanently adhering the microscope slide to the channels, clamps were

45

used to secure the slide in place, which allowed easy swapping of slugs and layer

adjustments in this preliminary prototype. The essence of testing the slug movement

due to pressure gradient inside the chambers remained valid with this setup. A light

layer of marker ink was applied on the topside of the slugs to help identify their

positions in the chambers during testing.

Slug lengths:

Channel intake widths:

0.5mm & 0.8mm

Slug lengths:

3.5mm to 7mm

Chamber widths:

1mm & 2mm

10mm

Channel wall

widths:

0.7mm & 1mm

Figure 3.5: Testing of the switching mechanism with simple channels. The rectangularslugs can be identified by the wiggly lines drawn with a marker pen. At this particularinstant, four slugs have already been moved, and the remaining two are about to bemoved.

Multiple combinations of slug lengths, chamber widths, channel intake widths,

and channel wall widths were planned with the design of the photomask. As observed

in Fig. 3.5, good adhesion was obtained, giving confidence in the range of channel

46

dimensions viable for future designs of RF devices. As a rule of thumb, channel intake

width should be about half of the chamber width, such that the ends of the chamber

have a well-defined wall to stop the moving slug, while still allowing a sufficient volume

of air to pass and move the slug. In this instance, the pressurized air originated from

a can of household compressed-gas dust-cleaner (through the red tube in Fig. 3.5). A

constant stream of gas was applied, and the tube was moved from the bottommost

channel towards the topmost channel. It was observed that the slug did not move

until the tube was directly aimed at the end of its corresponding channel, indicating

that the air flow was directed along the channel as designed with no air leakage (or

very minimal leakage that the slug in the adjacent channel was not influenced). Each

slug was found to be capable of moving back and forth inside its channel smoothly.

3.4.2 More Complex Configurations

It can be expected that the channels used in a functional RF device, such as a switch

or an antenna, would involve more complex features beyond the case of a single slug

in a straight channel demonstrated in Section 3.4.1. The ability to manipulate the

direction of air flow is the subject of this section, in which bends in channels and

multiple slugs in one channel are described.

Fig. 3.6 shows a meandering channel with several 90 bends. The slug moved with

the air flow back and forth inside the chamber without any issue, which confirms that

bends do not impede the air flow or affect air leakage.

A combination of slug positions for a three-port three-slug prototype is shown in

Fig. 3.7. When pressurized air was applied at port A, the air flow first pushed the

leftmost slug away from its port and then split off at the junction to move the other

47

Figure 3.6: A meandering channel.

two slugs towards their corresponding ports. Similar behaviour was observed when

pressurized air was applied at ports B and C. This test demonstrated the feasibility

of incorporating multiple slugs that move in orthogonal directions in a single channel,

and that airflow around a slug was capable of moving other slugs.

The configuration shown in Fig. 3.8 is a proof of concept of channel blocking. The

device consisted a blocking slug (1 mm × 7 mm), a main slug (1 mm × 8 mm), and

four access ports. The main slug was free to move within its chamber (positions #1

and #2), as long as the blocking slug resided closer to port A. This was accomplished

by applying pressurized air at port C. When pressurized air was applied at port A, the

blocking slug would slide past the intersecting channel BD and stop at the position

closer to port C. Channel BD would be blocked, as shown in the “Block mode”

picture in Fig. 3.8. With the blocking slug in its blocking position, even if the supply

of pressurized air at port A was discontinued, the blocking slug would remain in place

regardless of any pressure gradient in channel BD. One of the conditions required to

realize a functional channel blocking device is that the blocking slug must be long and

narrow, which guarantees its forward progress as it approaches the intersection and

avoids the potential issue of having one of the slug corners caught at the intersection.

48

A

B

Slugs

B

C

Figure 3.7: A device with multiple chambers in each channel path. Each letterindicates the port at which pressurized air was applied in each instance.

If another channel is constructed parallel to channel BD, which is also intersected

by channel AC, the blocking slug in channel AC can control the two parallel channels,

and a four-state device can be achieved. Expanding on this idea, a nine-state device

is proposed, as shown in Fig. 3.9.

Each of the horizontally-sliding (main) slugs can rest in three positions, while the

vertically-sliding (blocking) slugs can block either channel CH or channel DG in one

instance. Nine different combinations of the main slugs can be achieved. Fig. 3.9

shows how the device can transform from state “1-4” to state “3-5” in four sequential

steps of applying pressurized air at the appropriate port.

The dimensions of the slugs and the separation distance between channels were

49

C

D

Position #1

Block mode

B

C

Position #2 A

Blocking

slug

Main

slug

Figure 3.8: Prototype demonstrating the concept of channel blocking.

given some careful considerations to minimize the chance of slugs getting caught

while crossing the junctions. Two design rules should be adhered to when designing

networks of intersecting channels:

1. The length of a slug should be at least 2.5 times the width of its intersecting

channel. This ensures that the slug will not catch one of the corners of the

intersecting channel due to the possible rotation (very slight but not completely

zero) of the slug during its movement across the intersecting channel.

For instance, the length of the main slugs is 4.5 mm given the width of their

intersecting channels AF and BE being 1.5 mm. Similarly, this rule is applied

to the blocking slugs (length 8 mm) and their intersecting channels CH and

DG (width 3 mm).

2. For a blocking slug that is required to span across two intersecting channels

during its movement, the separation distance between its intersecting channels

50

A B

Port sequence for

applying pressurized air:

B G A H

1 2 3

4 5 6

A B

C

D

EF

H

G

1.5

8

4.5

35 5

Blocking slug

Main slug

1 2 3

4 5 6

A B

C

D

EF

H

G

Dimensions in mm

4

1.5

Figure 3.9: Concept of a nine-state device and its fabricated channels.

51

should be greater than or equal to half the length of the block slug.

In this case, the separation distance between channels CH and DG is 4 mm

while the length of the blocking slugs is 8 mm.

Other interesting configurations, which are beyond the scope of this thesis project,

can be explored in the future. The RF devices (a switch and an antenna) that are

described in Chapter 4 and Chapter 5 do not require the more complex channel

configurations such as the multi-state topologies, but this preliminary feasibility study

of elaborate channel designs offers an interesting direction of future work.

In this chapter, the novel switching mechanism of pneumatically controlling solid

slugs inside channels was introduced, with the process and material selection described

in detail. Prototypes of both simple and complex channel configurations were fab-

ricated and tested, which demonstrated the viability of the concept of manipulating

the positions of enclosed slugs with pressure gradient.

With the mechanical aspect of the switching mechanism tested, Chapter 4 docu-

ments the viability test of the mechanism with respect to RF performance, for which

a capacitive switch was designed and fabricated.

52

Chapter 4

Design and Fabrication of a

Capacitive Switch

The design and fabrication of a pneumatically controlled capacitive switch is described

in detail in this chapter. The consistency achieved between simulated and measured

RF results can be considered an intermediate milestone preceding the implementa-

tion of the switching mechanism in more elaborate and practical designs, such as a

reconfigurable antenna with switchable polarization (Chapter 5).

As mentioned in Section 2.2.1, a variety of RF switches are available on the market,

and the selection of a switch depends on the application and the priority of different

figures of merit and trade-offs. The pneumatically controlled capacitive series switch

offers a low-loss alternative actuation mechanism, with great topological flexibility

due to the absence of electrical biasing structures. This mechanism replaces biasing

feedlines in the conducting layer with air channels in a separate dielectric layer, which

offers greater flexibility in the RF design of reconfigurable devices, such as antennas.

Substantial part of the work in this chapter has been published in [56] and [57].

53

4.1 Structure and Fabrication Process

The switch consists of a gap in a microstrip line that can be bridged by a movable

copper-coated slug, as illustrated in Fig. 4.1, and with the switching concept as de-

scribed in Section 3.1. The slug is free to move within an enclosed channel formed in

a polymer layer situated between the substrate and the cover. Pneumatic ports are

incorporated at the ends of the channel structure, which both limit the slug move-

ment and also allow pressure gradients to be applied to the channel. Alternating the

direction of pressure gradient in the channel will cause the slug to move, and so the

device will switch between closed and open states, as shown in Fig. 4.2. When the

slug rests on top of the gap in the closed state, the significant increase in capacitance

due to the overlapping area between the slug and the microstrip across the gap creates

an RF capacitive short.

The slug and the microstrip have similar widths to minimize discontinuity in the

closed state. The copper-coated side of the slug faces the microstrip to maximize

the capacitive coupling in the closed state. A thin layer (200 nm) of silicon dioxide

is sputter-deposited on top of the copper coating of the slug to ensure a capacitive

contact instead of a metal-to-metal contact.

The fabrication process, developed by the author, is depicted in Fig. 4.3. The

substrate can either be a 30-mil-thick (0.762 mm) Rogers RO4350B hydrocarbon

ceramic laminate board (ǫr = 3.66, tanδ = 0.004) or two stacked borosilicate glass

wafers (each 0.5 mm thick, ǫr = 4.6, tanδ = 0.005), offering good design flexibility.

The laminate board comes with a 35-µm-thick copper cladding. Additional steps

of sputter-depositing/electroplating 5 µm of copper on the top and the bottom side

are required if the stacked glass wafers are used. The microstrip line with a gap

54

Cover

(glass)

Copper-coated slug

(glass & copper)

Channel

(polymer)

Microstrip

(copper)

Substrate

(glass or laminate)(glass or laminate)

Ground

(copper)

35mm

20mm

25mm

Port A

Port B

Figure 4.1: Exploded view of the switch showing various layers involved.

is patterned on the substrate. Copper is coated on top of a 0.18-mm-thick cover

slip and 1.87-mm-wide slugs of various lengths are diced out. SU8 photoresist with a

thickness of 0.2 mm is deposited and patterned on a glass wafer to produce the covered

channel. 0.8-mm-wide holes are drilled on the glass to serve as inlet ports. The three

components of the switch — the substrate, the slug, and the covered channel — can

be processed simultaneously, which allows greater adaptability and easy replacement

of defective parts. Adhesive can then be applied at specific locations outside the

channel, the components aligned, and the glue left to cure to produce a prototype

55

Closed Switch

Substrate

Cover

Open Switch

Channel wallAir

AirPort B

Gap in microstripCopper-

coated slug

Ground

Microstrip

Port A

Figure 4.2: Side profile illustrating two states of the switch.

switch. The width of the microstrip is 1.75 mm on the laminate to yield Zo = 50Ω.

The channel chamber has an inner width of 1.95 mm, which offers 40 µm of clearance

on each side of the slug. Coax connectors (not shown in figures) are then soldered on

the ends of the microstrip to allow for RF measurement.

56

Substrate

Cover

Deposit & pattern resist

on glass wafer

Pattern copper on substrate

Deposit copper on cover slip

& dice out slug

Cover

Drill holes in glass

Cover

Apply adhesive

Align layers & assemble

Substrate

Cover

Figure 4.3: Custom fabrication process.

4.2 Simulation

The switch was simulated for two different slug lengths and two different microstrip

gap sizes in HFSS (a commercial finite element method solver). Since the precise

profile of the contact between the slug and the microstrip was difficult to predict,

the effects arising from the deposited silicon dioxide on the copper-coated slug and

the microstrip’s copper surface roughness have been combined into one quantity,

called hair. It represented the average air gap separation between the slug’s and

the microstrip’s copper layers in simulations. Due to the lateral dimensions of the

switch being much greater than the thicknesses of its layers, it was expected that the

performance of the switch is much more sensitive to variations in layer thicknesses,

especially hair. The simulated results, which were de-embedded to the edge of the

57

glass cover and ignored the effect of the coax-to-microstrip transitions, are shown in

Fig. 4.4 and Fig. 4.5.

0 5 10 15 20 25−1

−0.8

−0.6

−0.4

−0.2

0

Freq (GHz)

|S21

| (dB

)

Ls=6, h

air=15

Ls=8, h

air=15

Ls=6, h

air=10

Ls=8, h

air=10

Ls= 6 & 8, h

air= 0.5 & 1

Figure 4.4: Simulated insertion loss of the closed switches, for two different sluglengths (Ls in mm) and several average air gap separations (hair in µm), given mi-crostrip gap of 2 mm.

From this it can be seen that the losses from the dielectric and the conductor were

accounted for. For the switch in the closed state, a large air gap separation poses a

lower limit in the switch’s frequency range. But even for an unlikely high gap value

of 15 µm, the insertion loss remains below 0.5 dB within the range of 4 to 15 GHz.

For the switch in the open state, the isolation is solely determined by the size of the

microstrip gap, g, and as expected, the larger gap of 2 mm offers better isolation.

Performance degradation from misalignment of the layers was also investigated in

simulations and it was determined that an error as large as 0.1 mm has negligible

effect, which is not surprising for this simple microstrip structure with a slug in the

58

0 5 10 15 20 25−40

−35

−30

−25

−20

−15

−10

−5

0

Freq (GHz)

|S21

| for

ope

n sw

itch

(dB

)

|S11

| for

clo

sed

switc

h (d

B)

Ls=6, h

air=15

Ls=8, h

air=15

Ls=6, h

air=10

Ls=8, h

air=10

OpenClosedg = 2mm Closed

Ls = 6 & 8

hair

= 0.5 &1

g = 2mm

g = 2mm

g = 1mm

Figure 4.5: Simulated return loss of the closed switches (scenarios similar to those inFig. 4.4); and simulated isolation of the open switches, for two different microstripgaps (g in mm).

scale of millimeters. This knowledge of a good tolerance to misalignment is beneficial

during the design and production of the device.

4.3 Fabrication and RF Measurement

Different samples of the covered channels for housing 6-mm and 8-mm long slugs

were fabricated. As mentioned in Section 3.4.1, the glass cover could potentially

crack during mechanical hole drilling. An example of a cracked cover is shown in

Fig. 4.6. The yield of successfully drilled glass covers improved with practice.

The use of glass wafers as the substrate was first considered. A copper layer with

59

Figure 4.6: Cracked glass cover from mechanical drilling.

a thickness of 4 µm was coated on a 0.5-mm-thick glass wafer using a magnetron

sputtering deposition system, as described in Section 4.1. A thin layer (100 nm) of

titanium was used as the seed layer to enhance the adhesion between the glass surface

and the coated copper. Adhesion appeared to be good initially and passed adhesion

tests. The copper-coated wafer was then patterned via photolithography, using a

photomask populated with several microstrip designs and etched in a ferric chloride

bath. The resulting wafer is shown in Fig.4.7. Pockets of air appeared underneath the

microstrip line, which was a copper adhesion issue likely caused by thermal stresses

in the copper film from the sputter deposition process. This issue may be mitigated

by heating the glass wafer during the sputter deposition, which should reduce the

thermal stress on the copper film after the process.

The alternative of using a laminate board as the substrate was explored, with the

microstrip lines milled out from the board. Fig. 4.8 shows a successfully fabricated

switch with a 2-mm microstrip gap on the laminate board. The initial prototype

60

Delamination of

deposited copper

Figure 4.7: Copper adhesion issue on the patterned glass wafer.

device was clamped together, so allowing easy reconfiguration and adjustments to

the layer alignment and channel geometries. The measurement setup is illustrated in

Fig. 4.9.

After the assembly of the prototype device, coaxial cables were attached to the

SMA connectors to allow RF measurements to be performed. The state of the switch

was then successfully changed between open and closed, using pneumatic control of

the slug, as described in Section 3.1. This demonstration of the actuation mecha-

nism was accomplished by alternately applying short bursts of air from a household

compressed gas duster at the two inlet ports (similar to the method employed in Sec-

tion 3.4.1). The results from three consecutive measurements of the closed switch are

plotted in Fig. 4.10 (up to 15 GHz, which is near the upper operating limit of SMA

cables and connectors). The overlapping curves demonstrate the great repeatability

61

ClosedPort B Port A

Support

features

Alignment

markChamberSlug

50mm

20mm

Open 25mm

Figure 4.8: Fabricated switch in closed and open positions.

of the switch behaviour. This repeatability test also served to settle the concern of

the slug potentially bouncing too far away from the end of its chamber. The good

alignment of measured data after numerous switching actuations demonstrates that

any variation in slug resting position has a negligible effect on the performance of the

switch, and that the switch operating repeatability (from open to closed) is good, i.e.

the mechanical performance of the switch is as anticipated. The results, however, also

expose the main drawback of the measurement design of the switch, which is the pres-

ence of two coax-to-microstrip transitions. The insertion loss becomes significantly

higher than the simulated values at frequencies higher than 3 GHz.

To present an accurate report on the RF performance of the switch alone, a

62

Figure 4.9: Experimental measurement setup of the switch.

Thru-Reflect-Line (TRL) calibration [58] was performed to remove the undesirable

effects of the coax-to-microstrip transitions as well as the microstrip sections from the

transition to the edge of the cover. The idea behind the TRL calibration in virtually

moving the reference planes of the measurement is depicted in Fig. 4.11. The custom

TRL calibration standards are shown in Fig. 4.12. The “reflect” standard was an

0 5 10 15−2

−1.5

−1

−0.5

0

Freq (GHz)

|S21

| (dB

)

0 5 10 15−40

−30

−20

−10

|S11

| (dB

)

Figure 4.10: Overlapping curves showing good repeatability of the switch. Each tracerepresents a separate measurement taken after closing the switch.

63

open-ended microstrip line. Two “line” standards were used to cover the measured

range of 0.5 to 15 GHz. Six separate measurements — switch, thru, reflect (port1),

reflect (port2), line#1, and line#2 — were performed to collect the necessary data

for the TRL program to generate the de-embedded results.

DUT(Device

Under Test)

Coax-

to-

Micro-

strip

Micro-

strip

Line

Coax-

to-

Micro-

strip

Micro-

strip

Line

De-embedded

reference

plane

De-embedded

reference

plane

Initial

reference

plane

Initial

reference

plane

Figure 4.11: Concept of TRL calibration.

The TRL de-embedded results of the open switch are compared against the direct

measurement in Fig. 4.13. The narrow discontinuity near 3.5 GHz in the de-embedded

results is simply a gap in the valid data range associated with using only two “line”

standards, which does not impact the general trend observed. The de-embedded

64

Reflect

Thru

Line #1

Line #2

25mm

5mm

12.5mm

Figure 4.12: Custom TRL calibration standards.

results have a return loss much closer to 0 dB at higher frequencies, which is an

indication that the directly measured results are subject to a non-negligible amount

of power lost due to radiation and dissipation at the coax-to-microstrip transition.

The difference between the de-embedded and the directly measured insertion loss

of the closed switch is significant, as seen in Fig. 4.14. The insertion loss at 14 GHz

is adjusted from 1.7 dB to a respectable value of 0.7 dB after de-embedding. In fact,

the insertion loss is lower than 0.7 dB for the entire measured frequency range. The

de-embedded insertion loss is consistent with the simulated result for hair = 1µm,

and they match very closely up to 8 GHz, which suggests that the surface contact

between the slug and the microstrip equates to a very small air gap separation. The

65

0 5 10 15−50

−40

−30

−20

−10

Freq (GHz)

|S21

| & |S

12| (

dB)

0 5 10 15−2

−1.5

−1

−0.5

0

Freq (GHz)

|S11

| & |S

22| (

dB)

TRL de−embedded

TRL de−embedded

Direct measurement

Direct measurement

Simulation

Figure 4.13: Measured performance of the open switch (g = 2mm and Ls = 8mm),before and after de-embedding using TRL calibration.

oscillating ripples, found at higher frequencies in the directly measured results, have

been noticeably reduced after de-embedding.

It should be noted that the simulated and the de-embedded measured results for

the 6-mm long slug did not show the same consistency that the 8-mm long slug did.

This can largely be attributed to the adhesion issue of the coated copper on the diced

slug, similar to that experienced when attempting to fabricate microstrip lines on a

glass wafer. The copper film became slightly detached from the edges of the slug, as

shown in Fig. 4.15, which lifted the slug away from the microstrip in the assembled

switch and thereby increasing the air gap separation (hair). As mentioned before, this

issue may be mitigated by heating the cover slip during the sputter deposition.

66

0 5 10 15−2

−1.5

−1

−0.5

0

Freq (GHz)

|S21

| & |S

12| (

dB)

0 5 10 15−40

−30

−20

−10

Freq (GHz)

|S11

| & |S

22| (

dB)

TRL de−embeddedDirect measurement

TRL de−embedded

Simulation

Direct measurement

Simulation

Figure 4.14: Measured performance of the closed switch, before and after de-embedding using TRL calibration.

Figure 4.15: Copper-coated glass slugs showing film peeling at edges.

67

4.4 Pressure Measurement

The prototype device was tested for the minimum actuation pressure required to move

the slug, using the setup shown in Fig. 4.16. One end of a tube (3 mm inner diameter)

was bonded to the inlet port, the other end was connected to a y-piece splitter, which

allowed a pressure-regulated supply of nitrogen to be applied to the device, as well as

the pressure applied to be monitored simultaneously using a pressure gauge. It was

determined that a minimum pressure of approximately 4 kPa (0.6 psi) was required

to reliably move the slug. However, the actuation pressure used was subsequently

increased to 10 kPa (1.5 psi), to ensure positive and responsive slug motion. The

effects of device orientation on the slug activation pressure were also investigated. It

was found that the minimum operation pressure of 10 kPa was able to satisfactorily

move the slug with the device mounted either horizontally or vertically. It should

be noted that after a short burst of higher-pressure air (≈ 10 kPa) to move the slug

into the required position, a lower pressure supply (a few kPa) could then be used to

“hold” the slug in position.

68

From

pressure-regulated

nitrogen supply

Switch (layers clamped together)

Inlet ports

To

pressure gauge To

inlet port

Figure 4.16: Setup of pressure measurement.

4.5 Discussion

A summary of the de-embedded RF performance of the fabricated switch is tabulated

in Table 4.1. The results are comparable to some of the existing low-loss switches

[22]–[24]. Given the current dimensions of the switch and the physical displacement

of a slug associated with its switching mechanism, the switch can be a reliable and

low-loss option from low MHz to 20 GHz with reasonable power handling capability

and immunity to RF modulation. One limitation of the current design is that the

switching speeds are not as fast as other alternatives, which excludes it from high-

speed applications. Using multiple pneumatically controlled switches in a device such

as a reconfigurable antenna would require careful design to optimize the channel

69

network and pneumatic port layout. However, due to the channel network being on a

separate polymer layer, and there being no requirement for any electrostatic actuation

circuitry near the slug, the layout in the conducting layer is less restricted when using

pneumatic control. In order to realize a more complete solution for reconfigurable

devices in future research, pumps and valves — technologies that are commercially

available in the field of microfluidics — could be connected directly to the ends of the

channels, away from the RF devices. Such pneumatic components could then provide

the source of the actuation pressure gradient required for slug positioning and control,

as well as the means to address the individual switches in a network.

A thorough evaluation of any emerging RF switching technology includes a reli-

ability test that subjects the switch to millions of on-off cycles [22]. This is beyond

the scope of this project, but will serve to answer the questions of when the slug or

the channel wall will wear out and how the performance of the switch will deteriorate

over time with use. Since this switch is not intended for high-speed applications, the

number of switching cycles it will experience should be significantly smaller.

Table 4.1: Summary of de-embedded measured results.

Parameter At 5 GHz Up to 15 GHz

Insertion Loss (closed) 0.2 dB < 0.7 dBReturn Loss (closed) 24 dB > 15 dB

Isolation (open) 25 dB > 15 dB

As described in Section 4.1, the increase in capacitance in the closed state relies on

the overlapping between the slug and the microstrip, which may be prone to vibration.

Some kind of pressure locking mechanism may be required if the performance of the

70

switch is found to be noticeably influenced by vibration in likely application settings.

Also, a sizable overlapping area is crucial for realizing a reasonable capacitance. This

can be an issue for applications in which form factor cannot be compromised. But

for reconfigurable antennas, in which there are generally large expanses of conducting

layer to serve as radiating elements, this pneumatically controlled switch can easily be

integrated without increasing the antenna size. A circularly polarized antenna with

its sense of polarization controlled by pneumatic means is the focus of Chapters 5

and 6.

71

Chapter 5

Ring Slot Antenna with Switchable

Polarization

The concept of this reconfigurable antenna with switchable polarization using pneu-

matic control was inspired by the reconfigurable ring slot antenna demonstrated in

[1]. The theory behind the generation of circular polarization from a single microstrip

feed for the ring slot antenna is discussed in Section 2.4. Its particular structure and

implementation of PIN diodes lends itself to a relatively straightforward adaptation

of the sliding slug mechanism, as explained in this chapter.

The antenna structure and its operation of reconfigurable polarization states is de-

scribed, along with the simulation results to justify the selection of various dimensions

and its simulated performance. The design for two streams of fabrication process will

be presented. The first one involves using glass wafers as the antenna substrate and

SU8 photoresist as the channel layer, while the second one utilizes low-loss laminate

boards for the substrate and the channel layer. Both choices of process were intro-

duced in the fabrication of the switch, described in Section 4.1. The benefits and

drawbacks of each will be compared at end of this chapter. Substantial part of the

work in this chapter has been published in [59], [60], and [61].

72

5.1 Changing Polarization Using Pneumatic Control

In order to realize the pneumatic control, a multilayered structure, similar to the

capacitive switch described in Chapter 4, is required. Fig. 5.1 illustrates the various

layers of the proposed antenna. The channel in which the slugs reside, sits on top of

the ground plane. The channel is sealed with a cover layer on top. Holes are drilled

in the cover to create the ports for pressurized air. In contrast to the work published

in [1], in which two separate reconfigurable antennas were fabricated with each hav-

ing two possible polarization states as shown in Fig. 2.8, this proposed antenna has

all three polarization possibilities (LP/RHCP/LHCP). The respective port through

which pressurized air is applied to realize a particular state is indicated in Fig. 5.1.

The positions of the slugs leading to each polarization state are discussed below.

5.1.1 Linear Polarization

The top view of the antenna in LP mode is shown in Fig. 5.2 (the cover layer is

omitted). Pressurized air enters the channel from the port labeled “LP” in Fig. 5.1,

which aligns with the corresponding circle of the channel as seen in Fig. 5.2. Air flows

towards the other three exits (two ports in the cover layer and one gap in channel),

and in the process pushes each slug to one end of the chamber it resides in. The

narrow slotlines are not overlapped by the slugs, and therefore all four rectangular

openings are not bypassed. Linearly-polarized radiation oriented in the x-direction is

generated.

The small gap in the channel ensures that there is air flow into the right half of

the channel. Without the gap, air would likely escape the channel only through the

two ports.

73

Cover

tcover

Channel / Air

tchannel

Slugs

tslug

Port for

pressurized air

LP

RHCP

LHCP

Ground / Slot

tcopper

Substrate

tsub

Microstrip feed

tcopper

Figure 5.1: Structure of the switchable polarization ring slot antenna.

5.1.2 Right-Hand and Left-Hand Circular Polarizations

Similarly for RHCP, when pressurized air is applied to the corresponding port, the

slugs will be positioned as shown in Fig. 5.3. The phase relation between the two

diagonal modes leads to RHCP radiation leaving from the antenna in the +z-direction

(and LHCP radiation leaving from the antenna in the –z-direction). The case for

LHCP is illustrated in Fig. 5.4.

A ring slot antenna that uses pneumatic control would present the advantages

discussed in Chapter 3. A prototype of a practical antenna using this mechanism of

74

y

x

Copper-

plated slug

Channel

air out

LPair in

air out

Figure 5.2: Direction of air flow and the resulting positions of slugs for generatinglinear polarization.

reconfiguration would serve the purpose of demonstrating the feasibility of implement-

ing this scheme as an alternative to the existing methods of reconfiguration in other

reconfigurable antenna designs. With the reasons for implementing the pneumatic

control scheme justified, the other question to ask is: Why is the ring slot antenna

with rectangular openings chosen for demonstrating the viability of this scheme of

reconfiguration?

As hinted in Chapter 3, the mechanism of sliding dielectric slugs is well suited for

creating a distinct change in capacitance as in the case of RF switches. The capacitive

change originates from “macroscopic” movement of the slug as opposed to “micro-

scopic” movement of the cantilever in a MEMS switch or charges in a PIN diode.

The nature of this “macroscopic” movement contributes to increased uncertainty of

75

y

xCopper-

plated slug

Channel

0° -90°

air out

RHCPair in

air out

Figure 5.3: Direction of air flow and the resulting positions of slugs for generatingRHCP.

the realized capacitance values when compared to those achieved by the finer “mi-

croscopic” movement associated with MEMS switches and PIN didoes. Therefore, a

reconfigurable antenna structure that is less sensitive to capacitance values realized

by the switching mechanism should be selected for this project, which is currently

a proof of concept. The ring slot antenna with rectangular openings connected via

narrow slotlines is an ideal candidate. This is because ground current around the

ring can effectively bypass an opening, as long as a certain level of capacitance value

is achieved by the slug overlapping the two sides of the slotline. The absolute value

of the capacitance realized is not critical as far as antenna performance is concerned.

This particular attribute of this chosen antenna structure also helps reduce the influ-

ence on the behaviour of the antenna due to possible vibration of the slugs.

76

y

x

Copper-

plated slug

Channel

-90° 0°

air out

LHCPair in

air out

Figure 5.4: Direction of air flow and the resulting positions of slugs for generatingLHCP.

With the means of achieving reconfigurable polarization states serving as the pro-

logue to designing the antenna, the specific parameters required to fulfill the condition

for circular polarization — two linear orthogonal modes with equal magnitude and

90 phase separation — will be the focus of the rest of this chapter.

5.2 Design using glass wafers and photoresist

There are many sets of parameter values that could yield an antenna with acceptable

AR and impedance bandwidths for a specific range of operating frequency. These

parameters can mainly be divided into three groups: thicknesses, material permittiv-

ities, and lateral dimensions.

77

5.2.1 Dimensions

The thicknesses of the layers, as listed in Fig. 5.1, are given in Table 5.1. These values

are chosen mainly because of fabrication constraints and the materials available, as

mentioned in Chapter 4.

Table 5.1: Layer thickness of the ring slot antenna with glass substrate.

tsub (2 wafers) 1mm tcover (1 wafer) 0.5mmtchannel 0.2mm tslug 0.18mm (plus 5um)tcopper 5um

Similar to the thicknesses, material permittivities are dictated by the materials

chosen for the fabrication process. The relative permittivity of the SU8 photoresist,

which is used to construct the channel, is taken to be 2.89 with tanδ of 0.04 [62]. Each

fused-silica glass wafer (ǫr,sub = 3.78, tanδsub = 0.0004) has a thickness of 0.5 mm.

The substrate consists two stacked wafers, as the resulting overall thickness of 1 mm

dictates a reasonable microstrip width.

The lateral dimensions of various features can be designed properly to allow for

reconfigurable polarization operation of the antenna at the design frequency, which

is the 2.4 GHz range as in [1]. These dimensions are illustrated in Fig. 5.5. Some

parameters, such as the ring radius and the size of the openings, are more influential

than others, the channel wall width for instance. The radius of the ring, ro plays a

dominant role in deciding the resonant frequency of the antenna. The circumference

of the ring is generally one guided wavelength [1], [53].

Lateral dimensions that are more pertinent to impedance matching, namely the

slotline and microstrip stubs, are shown in Fig. 5.6. The resonant frequency of the

78

wchamberwslug

gring

wchannel

wwall

lslug

rport

gsl

y

x

ro

lopen

wopen

lchamber

gchannel

To coax

connector

Figure 5.5: Dimensions of the ring slot with glass substrate.

antenna sees a lesser degree of influence from these parameters, but they obviously

have a strong effect on how well the antenna is impedance matched and how deep the

resonance is. The width of the microstrip line (wms) can be determined analytically

to realize a Zo = 50Ω line given ǫr,sub and tsub.

With such a long list of parameters to be determined, different simulations were

set up in HFSS to progressively arrive at the final set of dimensions. The first sim-

ulation was a LP ring slot antenna without the rectangular openings, with lateral

dimensions similar to those in [1] but thicknesses and material permittivities specific

79

wms

wmitre

θstub,sl

θstub,ms

rstub,sl

rstub,ms

lsl

wsl

lms

Figure 5.6: Dimensions of microstrip-to-slotline transition with glass substrate.

to the fabrication process of this project. After a number of iterations and some

optimization, a tentative set of parameters related to impedance matching (shown in

Fig. 5.6) were determined. Next, the LHCP antenna in Fig. 2.12(a) was simulated to

determine the size of the rectangular openings, while tuning the impedance matching

parameters and ro for maximum AR bandwidth at the centre frequency of 2.4 GHz.

Further adjustments to these parameters were required to ensure the performance of

the RHCP antenna was comparable since the microstrip line and stub presented an

asymmetry to the structure.

Finally, the reconfigurable antenna in Fig. 5.5 was simulated, with the slugs,

channel, and cover in place. The HFSS simulation setup is illustrated in Fig. 5.7.

80

As with the switch simulations in Section 4.2, the microstrip line was excited with a

lumped port and radiation boundary was set up as the end limit of the simulation

domain to emulate the infinite extension to free space. To ensure the simulation

results would be realistic, the 4-inch-diameter glass substrate was included in the

setup, with the dielectric losses of the substrate, cover, and channel as well as the

metal losses of the microstrip and ground accounted for. In order to simplify the

simulation setup, the coax connection to the microstrip line was omitted. Observing

the comparison between directly measured and TRL de-embedded results in Fig. 4.14,

the effect of the coax-to-microstrip transition was not pronounced at low frequencies

near 2.4 GHz. Also, the air ports (drilled holes) on the cover layer for applying

pressurized air were not included.

Lumped

port

yx

z

85mm

Figure 5.7: Simulation model of the reconfigurable ring slot antenna with glass sub-strate in HFSS.

81

Table 5.2 lists the various parameters shown in Fig. 5.5 and Fig. 5.6 for the final

antenna design. As mentioned before, many parameters were obtained progressively

by simulating a linearly polarized antenna and a fixed circularly polarized antenna.

The ones associated with the channel were largely determined by fabrication con-

straints. The remaining parameters would be finalized by observing the results from

parametric sweeps, which will be discussed below. Note that wslug is slightly smaller

than wchamber to allow the slug to slide within its chamber smoothly while keeping its

direction of travel aligned to the length of the chamber.

Table 5.2: Dimensions of the ring slot antenna with glass substrate.

From Fig. 5.5 From Fig. 5.6

ro 15mm rstub,sl 4.75mmgring 2mm θstub,sl 140

wslug 3.75mm wsl 1.55mmlslug 6mm lsl 6.05mmwopen 3.27mm rstub,ms 5.5mmlopen 4.14mm θstub,ms 75

gsl 0.4mm wms 2.1mmwchamber wslug + 0.12mm lms 13.05mmlchamber 1.5lslug + lopen wmitre 2.344mmwwall 0.8mm

wchannel wchamber/2gchannel 1.5mmrport 1.5mm

5.2.2 Parametric Analysis

Some of the parameters that are considered crucial for the resonant frequency and the

AR performance of the reconfigurable antenna include: ro, gring, wslug, lslug, wopen,

and lopen.

82

A methodical approach to identify the optimal set of parameters would be to

set gring, wopen, and lopen to the values determined with the simulation of the fixed

circularly polarized antenna, and perform parametric analyses on ro, wslug, and lslug

to observe their effects on the resonant frequency and the AR performance. Trade-offs

between the parameters are discussed next.

It should be noted that a fourth parameter was included in the parametric anal-

yses. The thin air/dielectric gap between the coated copper layer on the slug and

the ground layer, denoted as hair in Section 4.2, will be present in the fabricated

antenna. It is this layer that makes the switching contact capacitive rather than

metal-to-metal. Similar to the switch simulations in Section 4.2, a few possible hair

values were swept in the antenna simulations to observe the effect of variation in the

air gap separation. From the simulation results of the switches in Section 4.2, the

effect on RF performance is negligible if the air gap is small (around 0.5 to 1 µm,

which is deemed reasonable for the fabricated antenna). The two values of hair used

in the parametric sweeps were 0.5 and 10 µm, with the intention to compare the

degree of performance between a realistic value and an outlying value.

Rather than simply presenting the results of the parametric sweeps, a prior discus-

sion on some of the expected trends and influences of the parameters would facilitate

the interpretation and understanding of the results.

The obvious effect of varying the radius of the ring slot, ro, is the shift in the

resonant frequency of the antenna, as mentioned in Section 5.2.1.

The area occupied by each slug (wslug · lslug) is directly proportional to the capaci-

tance enhancement when the slug bridges across the narrow slotline that connects the

rectangular opening and the ring slot. Much like the scenario of a simple parallel-plate

83

capacitor, the sensitivity of the capacitance is dominated by the air gap separation

as opposed to the slug area. Therefore, it can be expected that for a well-designed

antenna, its performance should not be overly sensitive to changes in the slug area.

However, a distinction needs to be made between varying the slug width (wslug) and

the slug length (lslug). From the geometry of the antenna in Fig. 5.5, it seems reason-

able to assume that increasing the length (lslug) of the slugs beyond the length of the

rectangular opening (lopen) would yield negligible effect on the current flow around

the ring slot. Varying wslug, on the other hand, would have multiple effects, besides

changing the slug area. This is because the length of the narrow slotline is pegged to

the slug width by design, and the length of the narrow slotline is one of the factors

that decides the length of current path around the ring slot (which ultimately affects

the resonant frequency and the AR performance).

Fig. 5.8 and Fig. 5.9 illustrate the minimum AR achieved in simulations and the

frequency at which it is obtained, respectively, over a range of wslug, lslug, and hair

while keeping ro at 15.5 mm. All other parameters were kept constant, as given in

Table 5.2. In general, it can be seen that a small minimum AR value in the range

of 0.1 to 0.2 can be reached in the vicinity of the design frequency of 2.4 GHz given

an appropriate combination of parameters. This remains true even when the air gap

separation is assumed to be unrealistically high (10 µm). This certainly provides

reassurance for the decision to select the ring slot antenna with rectangular openings

as the candidate for the reconfigurable antenna.

It is immediately obvious, from Fig. 5.8, that for the realistic air gap separation

of 0.5 µm, increasing the length of the slug from 4 mm to 10 mm has negligible effect

on the minimum AR achieved since the three curves overlap closely. This confirms

84

1 2 3 4 5 60

0.5

1

1.5

2

2.5

3

Slug Width (mm)

Min

. AR

at B

road

side

(dB

)

lslug

=4mm, hair

=0.5um

lslug

=7mm, hair

=0.5um

lslug

=10mm, hair

=0.5um

lslug

=4mm, hair

=10um

lslug

=7mm, hair

=10um

lslug

=10mm, hair

=10um

Figure 5.8: Minimum AR achieved for various slug widths, slug lengths (lslug), andair gap separations (hair).

the hypothesis discussed earlier. The trend diverges when the separation is increased

to 10 µm. In this case, it is observed that a small minimum AR value can still be

reached when the slugs are longer (7 mm and 10 mm), though the corresponding slug

width at which the minimum AR is obtained is shifted. When hair is set at 10 µm,

the curve for lslug = 10 mm is the closest to the overlapping set of hair = 0.5 µm

curves. This can be explained by the fact that longer slugs help compensate for the

reduction in capacitance from the increased air gap separation.

The linear relationship observed in Fig. 5.9 between the slug width and the fre-

quency at which the minimum AR is obtained is a result of the length of the narrow

slotline being pegged to the width of the slug. The narrow slotline decides how far

85

1 2 3 4 5 62.2

2.25

2.3

2.35

2.4

2.45

2.5

Slug Width (mm)

Fre

quen

cy o

f Min

. AR

(G

Hz)

lslug

=4mm, hair

=0.5um

lslug

=7mm, hair

=0.5um

lslug

=10mm, hair

=0.5um

lslug

=4mm, hair

=10um

lslug

=7mm, hair

=10um

lslug

=10mm, hair

=10um

Figure 5.9: Frequency of minimum AR for various slug widths, slug lengths (lslug),and air gap separations (hair).

the rectangular opening is from the ring slot, and hence it directly dictates how much

further current has to flow around the ring and subsequently the resonant frequency.

The overlapping curves indicate that the resonant frequency is mostly independent

of the length of the slugs and the air gap separation.

The 3-dB AR bandwidth of the antenna under different scenarios, with ro =

15.5 mm, is shown in Fig. 5.10. The antenna achieves a bandwidth of over 4% for

slug width greater than 3 mm. This is similar to those (around 3.4% to 4.3%) obtained

by [1] and [53]. This particular antenna design has the potential of reaching over 5%

bandwidth if the slug width is increased beyond 3.5 mm.

The ring outer radius, ro, was the next parameter to be swept. Different values

86

1 2 3 4 5 62

2.5

3

3.5

4

4.5

5

5.5

6

Slug Width (mm)

AR

Ban

dwid

th a

t Bro

adsi

de (

%)

lslug

=4mm, hair

=0.5um

lslug

=7mm, hair

=0.5um

lslug

=10mm, hair=0.5um

lslug

=4mm, hair

=10um

lslug

=7mm, hair

=10um

lslug

=10mm, hair

=10um

Figure 5.10: AR bandwidth obtained for various slug widths, slug lengths (lslug), andair gap separations (hair).

of the slug width and the air gap separation were also compared. From Fig. 5.8, the

minimum AR was reached when wslug was in the range of 3 to 4 mm. This range of

values would therefore be used for this parametric sweep. Since the influence of the

slug length was determined to be diminishing for values greater than lopen, it was set

to 8 mm for the subsequent simulations.

Fig. 5.11 shows the minimum AR obtained when a range of ro was swept. For

both values of hair, it is apparent that increasing the slug width moves the curve to

the right. Imagine the following scenario: starting at hair = 0.5 µm, ro = 14.5 mm,

and wslug = 3 mm, a respectable minimum AR value of 0.26 dB is achieved, and an

optimal design is found. If wslug were to be increased to 3.5 mm, which would lead

87

to a deviation from the optimal design, the minimum AR obtained would increase

to 0.55 dB. In order to regain a lower minimum AR value, one has to trace the

wslug = 3.5 mm curve in the direction of increasing ro to the point where ro = 15 mm,

which gives a minimum AR value of 0.24 dB.

13.5 14 14.5 15 15.5 16 16.50

0.5

1

1.5

2

2.5

3

Ring Outer Radius (mm)

Min

. AR

at B

road

side

(dB

)

w

slug=3mm, h

air=0.5um

wslug

=3.5mm, hair

=0.5um

wslug

=4mm, hair

=0.5um

wslug

=3mm, hair

=10um

wslug

=3.5mm, hair

=10um

wslug

=4mm, hair

=10um

Figure 5.11: Minimum AR achieved for various ring outer radii, slug widths (wslug),and air gap separations (hair).

In essence, when the slug width is increased (within a small range), the ring radius

also needs to be increased to maintain a similar AR value. This is a demonstration

of the intricate component sizing required to fulfill the criteria for generating circular

polarization: equal magnitude and quadrature phase difference between two linear

orthogonal modes. It should be noted that this discussion does not consider the

88

deviation of the resonant frequency and the change in AR bandwidth. The frequency

at which the new minimum AR is realized would certainly be lower because of the

increase in the length of the narrow slotline and the ring radius.

Fig. 5.12 affirms the trends observed in Fig. 5.9 in two ways. For each particular

wslug value, the two curves for the two hair values overlap, which again indicates the

independence of the resonant frequency from the air gap separation. The wslug curves

are spaced equally apart for each 0.5-mm increment in wslug, exemplifying the linear

relationship between the slug width and the frequency at which the minimum AR is

obtained, as seen in Fig. 5.9. A linear relationship between the frequency at which

the minimum AR is obtained and the ring radius is also observed in Fig. 5.12.

13.5 14 14.5 15 15.5 16 16.52.2

2.25

2.3

2.35

2.4

2.45

2.5

2.55

2.6

Ring Outer Radius (mm)

Fre

quen

cy o

f Min

. AR

(G

Hz)

wslug

=3mm, hair

=0.5um

wslug

=3.5mm, hair

=0.5um

wslug

=4mm, hair

=0.5um

wslug

=3mm, hair

=10um

wslug

=3.5mm, hair

=10um

wslug

=4mm, hair

=10um

Figure 5.12: Frequency of minimum AR for various ring outer radii, slug widths(wslug), and air gap separations (hair).

89

Finally, the sweep of the small range of ro reveals that the AR bandwidths achieved

under different scenarios are fairly consistent, as shown in Fig. 5.13.

13.5 14 14.5 15 15.5 16 16.52

2.5

3

3.5

4

4.5

5

5.5

6

Ring Outer Radius (mm)

AR

Ban

dwid

th a

t Bro

adsi

de (

%)

wslug

=3mm, hair

=0.5um

wslug

=3.5mm, hair

=0.5um

wslug

=4mm, hair

=0.5um

wslug

=3mm, hair

=10um

wslug

=3.5mm, hair

=10um

wslug

=4mm, hair

=10um

Figure 5.13: AR bandwidth obtained for various ring outer radii, slug widths (wslug),and air gap separations (hair).

Tracing each curve in Fig. 5.8 until a minimum point is reached is essentially

optimizing the slug width for the lowest possible AR value when all other parameters

are fixed. This optimization process is at the expense of potentially deviating from

the design frequency and lower bandwidth. Referring to the plots from Fig. 5.8 to

Fig. 5.13 generated with parametric sweeps, the best compromise between a wide AR

bandwidth and a low minimum AR at the design frequency of 2.4 GHz was decided

with the following parameters: wslug = 3.75 mm, lslug = 6 mm, and ro = 15 mm.

90

5.2.3 Simulated Performance

The final antenna design with the parameters given in Table 5.2 was simulated for

the three polarization states: LP, RHCP, and LHCP. The air gap separation was set

at 1 µm.

2 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8−25

−20

−15

−10

−5

0

Frequency (GHz)

|S11

| (dB

)

LPRHCPLHCP

Figure 5.14: |S11| of the simulated ring slot antenna (glass substrate) under differentpolarization states.

Impedance Matching (|S11|)

The impedance matching of the antenna is plotted in Fig. 5.14. The antenna is

impedance matched for all three states from 2.17 to 2.41 GHz. The capability of this

antenna to operate in all three polarization states was not demonstrated in [1] and

91

[53].

Figure 5.15: S11 (on the Smith Chart) of the simulated ring slot antenna (glasssubstrate) from 2 to 2.8 GHz under different polarization states.

Except for a small range of frequency where |S11| is slightly higher than −10 dB,

the impedance bandwidth of the antenna operating in circular polarization has a

wide impedance bandwidth, starting from 2.16 GHz and extending beyond 2.8 GHz.

This is the result of having the two resonances of the two orthogonal linear modes

being close to each other, causing the formation of a loop in the frequency sweep near

the centre of the Smith Chart (see Fig. 5.15). From Fig. 5.14, the two resonances

are located at roughly 2.31 GHz and 2.73 GHz. As mentioned in Section 5.1.1, the

LP state was realized with all four slugs positioned away from the narrow slotline.

92

Therefore, the rectangular openings are not bypassed and the current path around

the ring is increased, resulting in a lower resonant frequency. This is confirmed with

the antenna in the LP state having a resonant frequency coinciding with the lower

resonance of the circularly polarized antenna.

2.25 2.3 2.35 2.4 2.45 2.50

0.5

1

1.5

2

2.5

3

3.5

4

4.5

5

Frequency (GHz)

AR

(dB

)

RHCPLHCP

Figure 5.16: Broadside AR of the simulated ring slot antenna (glass substrate).

Axial Ratio (AR)

The AR of the simulated antenna at broadside (in the +z-direction) over a range

of frequency is shown in Fig. 5.16. The minimum AR achieved is lower than 0.7 dB

at about 2.36 GHz (RHCP) and 2.38 GHz (LHCP). The slight deviation in the AR

93

curve between the LHCP and RHCP cases (less than 20 MHz) can be attributed to

the asymmetric microstrip feedline. The 3-dB AR bandwidth is from 2.31 to 2.42 GHz

for RHCP and from 2.32 to 2.44 GHz for LHCP, which is about 5% when considering

2.38 GHz as the operating frequency.

−80 −60 −40 −20 0 20 40 60 800

1

2

3

4

5

6

7

8

9

10

θ (°)

AR

(dB

)

RHCP, φ = 0°

RHCP, φ = 90°

LHCP, φ = 0°

LHCP, φ = 90°

Figure 5.17: AR of the simulated ring slot antenna (glass substrate) in the twoprincipal planes at 2.38 GHz.

The AR on the two principal planes in both CP states at 2.38 GHz is given in

Fig. 5.17. The asymmetry in the curves about the z-axis (θ = 0) can again be

explained by the asymmetric feed structure. The performance is generally acceptable

(AR below 3 dB) for |θ| ≤ 30. Note that the RHCP performance is not as good as

that of LHCP because the frequency of minimum AR for the antenna operating in

94

the RHCP state is not at 2.38 GHz as in the case of the LHCP state, which is evident

in Fig. 5.16.

2 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.82

2.5

3

3.5

4

4.5

5

5.5

6

Frequency (GHz)

Gai

n (d

Bi)

LPRHCPLHCP

Figure 5.18: Broadside gain of the simulated ring slot antenna (glass substrate).

Antenna Gain and Radiation Pattern

The broadside gain of the antenna at 2.38 GHz in all three polarization states

reaches 5 dBi, as shown in Fig. 5.18. The antenna gain drops by less than 1 dB from

its maximum value within the impedance and AR bandwidths, which is respectable.

The radiation pattern in the RHCP state is illustrated in Fig. 5.19. A clarification

in terminology should be made here. RHCP radiation is as described in Section 2.1.1.

95

Figure 5.19: Simulated antenna pattern for φ = 0 and φ = 90 planes when operatingin the RHCP state at 2.38 GHz (glass substrate).

96

Figure 5.20: Simulated antenna pattern for φ = 0 and φ = 90 planes when operatingin the LHCP state at 2.38 GHz (glass substrate).

97

An antenna operating in the RHCP state is referring to a reconfigurable antenna that

is configured to produce pure RHCP radiation, ideally. In the context of this work

with a ring slot antenna, it is further specified as RHCP radiation in the +z-direction.

In reality, the antenna produces a small amount of cross-polarization — which in the

case of an antenna operating in the RHCP state — LHCP radiation. Therefore,

Fig. 5.19 shows both RHCP and LHCP radiation, and in the +z-direction (θ = 90),

RHCP radiation is significantly larger than LHCP radiation for this antenna operating

in the RHCP state, as expected.

The broadside gain is 5 dBi for RHCP wave traveling in the +z-direction. The

broadside cross-polarization level is –25 dB. As mentioned in Section 2.4, the reverse

CP wave (LHCP) is produced on the backside for a ring slot antenna, which is evident

in Fig. 5.19. The radiation pattern in the LHCP state is very similar to that in the

RHCP state, as shown in Fig. 5.20.

The radiation pattern in the LP state is noticeably different from those in the

CP states (see Fig. 5.21). Energy remains directed in the broadside direction, but

the backside sees the same dominating polarization as opposed to the case in the CP

states where the cross-polarization is dominant on the backside.

A summary of the simulated performance is presented in Table 5.3. Comparing

with Table 2.2, the simulated performance of the proposed antenna is similar to, if

not slightly better than, those of [1] and [53]. The key advantage of the proposed

antenna, as mentioned in Section 5.1, is its capability to achieve all three polarization

states.

98

Figure 5.21: Simulated antenna pattern for φ = 0 and φ = 90 planes when operatingin the LP state at 2.38 GHz (glass substrate).

99

Table 5.3: Simulated performance of the ring slot antenna using glass substrate.

Impedance bandwidth 2.17 to 2.41 GHz (10%)AR bandwidth (RHCP) 2.31 to 2.42 GHz (4.6%)AR bandwidth (LHCP) 2.32 to 2.44 GHz (5%)Minimum AR (RHCP) 0.67 dBMinimum AR (LHCP) 0.4 dB

Frequency of min. AR (RHCP) 2.36 GHzFrequency of min. AR (LHCP) 2.38 GHz

Broadside gain 5 dBiBroadside cross-pol. level < −25 dB

Ground Current

The cycle of the ground current at 2.38 GHz for each of the three states is presented

in “snapshots,” as shown in Fig. 5.22 to Fig. 5.24.

The resonant nature and the standing wave pattern in the LP state is clear in the

animation of the current distribution in Fig. 5.22. The animation shows a continuous

forward power transfer along the microstrip feedline to the ring slot, which confirms

the good impedance match at 2.38 GHz as indicated in Fig. 5.14.

The “hurricane”-like current distribution observed in Fig. 5.23 and Fig. 5.24 indi-

cates the generation of circularly polarized radiation of the antenna.

A prudent antenna design should include a sensitivity analysis as a result of fab-

rication tolerances. This is especially important for this particular project since a

custom fabrication process is used. The next section will discuss the results of the

sensitivity analysis of several parameters.

100

0°

90° 135°

45°

Figure 5.22: Ground current at different phases of a cycle in the LP state (glasssubstrate).

101

0°

90° 135°

45°

Figure 5.23: Ground current at different phases of a cycle in the RHCP state (glasssubstrate).

102

0°

90° 135°

45°

Figure 5.24: Ground current at different phases of a cycle in the LHCP state (glasssubstrate).

103

5.2.4 Sensitivity to Fabrication Tolerances

Given the nature of the custom fabrication process, fabrication tolerances require

greater attention compared to a standard proven manufacturing process. For this

process, three parameters that can potentially cause noticeable degradation to per-

formance are (referring to Fig. 5.25):

• the air gap separation between the copper-coated slugs and the ground — hair;

• the misalignment between the narrow slotlines on the ground and the slugs

(essentially between the ground wafer and the channel) — ∆xslug & ∆yslug; and

• the misalignment between the radial slot stub on the ground and the microstrip

feedline (essentially between the ground wafer and the microstrip wafer) —

∆xms & ∆yms.

Sensitivity analyses were performed in simulation to investigate the influence of

these parameters. From Fig. 5.14 and Fig. 5.16, it is observed that the performance

of the antenna operating in the RHCP state is not as good as that in the LHCP state.

Hence, the performance shown below is for the antenna operating in the RHCP state,

because it would serve as a more stringent test.

It was seen in Section 5.2.2 that an air gap separation of 10µm could cause a small

but noticeable change in the antenna performance. Also mentioned was that hair of

10 µm would be far too large. A realistic range for hair is ≈ 1 µm. While keeping

other parameters constant, the simulated AR, |S11|, and gain for this range of hair are

plotted in Fig. 5.26, along with the case of hair = 15 µm for comparison. Negligible

difference in the antenna performance was observed, with the exception of when hair

was increased to a very large value of 15 µm, which is beyond the clearance in the

channel.

104

∆yslug∆xslug

∆yms∆xms

hair

Ground

wafer

Microstrip

Channel

& slugs

Microstrip

wafer

Figure 5.25: Parameters considered in the sensitivity analysis.

The radiation pattern of the antenna when operating in the RHCP state, given

various air gap separations, is shown in Fig. 5.27. Only RHCP radiation is shown to

avoid overcrowding the plot. LHCP radiation, not shown here, looks very similar to

that shown in Fig. 5.19. The overlapping curves suggest that the radiation pattern,

especially in the +z-direction, is largely unaffected by varying hair. The radiation

patterns also see minimal change for the cases of misalignment discussed below.

In understanding the influence of the misalignment between the ground and the

slugs, a combination of ∆xslug and ∆yslug from –200 to 200 µm in steps of 50 µm

105

2.3 2.35 2.4 2.45

1

2

3

Frequency (GHz)

AR

(dB

)

2.1 2.15 2.2 2.25 2.3 2.35 2.4 2.45 2.5−15

−12.5

−10

−7.5

Frequency (GHz)

|S11

| (dB

)

2.1 2.15 2.2 2.25 2.3 2.35 2.4 2.45 2.53

4

5

Frequency (GHz)

Gai

n (d

Bi)

hair

=0.5um

hair

=1um

hair

=5um

hair

=15um

Figure 5.26: Performance variation due to changing hair (glass substrate).

were used in simulations. With the alignment of the layers being performed under

a microscope, and considering the attainable precision of the alignment marks on

the SU8 layer, misalignment in the x- and the y-directions should remain below

200 µm. The performance degradation, as shown in Fig. 5.28, is not significant when

compared to the case of perfect alignment. The AR increases by no more than 0.2 dB

at 2.38 GHz for the worst cases of misalignment (∆xslug = ∆yslug = ±200µm), with

|S11| and gain almost unchanged. The greatest impact from the misalignment of the

slugs is the change in capacitance between the slugs and the narrow slotlines, which

would affect the purity of the circularly polarized radiation. This is confirmed with

the change in AR being slightly more noticeable than the change in |S11|.

106

hair = 0.5, 1, 5um

hair = 15um

Figure 5.27: Simulated antenna pattern for φ = 0 (solid lines) and φ = 90 (dashedlines) planes when operating in the RHCP state at 2.38 GHz, given different air gapseparations hair (glass substrate).

107

2.3 2.35 2.4 2.45

1

2

3

Frequency (GHz)

AR

(dB

)

2.1 2.15 2.2 2.25 2.3 2.35 2.4 2.45 2.5−15

−12.5

−10

−7.5

Frequency (GHz)

|S11

| (dB

)

2.1 2.15 2.2 2.25 2.3 2.35 2.4 2.45 2.53

4

5

Frequency (GHz)

Gai

n (d

Bi)

Figure 5.28: Performance variation due to misalignment between the ground waferand the channel/slugs (glass substrate). The solid lines represent two combinationswith the largest misalignment (∆xslug = ∆yslug = ±200µm). The short dashed linerepresents the case of perfect alignment (∆xslug = ∆yslug = 0µm). The long dashedlines represent combinations of moderate misalignment.

The misalignment between the ground and the microstrip wafers was treated in a

similar fashion. However, due to the more precise alignment marks achievable with

the etched copper layers, the worst cases of misalignment were expected to be less

than 100 µm. Different combinations of misalignment in the x- and the y-directions,

in steps of 50 µm, were simulated, and the performance is shown in Fig. 5.29. Since

this particular misalignment would affect the microstrip feeding to the antenna, it is

obvious that the impedance matching sees slightly greater impact than the purity of

108

the circularly-polarized radiation. |S11| shifts slightly up by 0.5 dB at 2.38 GHz for

the worst cases of misalignment, while the AR and gain remain unaffected.

2.3 2.35 2.4 2.45

1

2

3

Frequency (GHz)

AR

(dB

)

2.1 2.15 2.2 2.25 2.3 2.35 2.4 2.45 2.5−15

−12.5

−10

−7.5

Frequency (GHz)

|S11

| (dB

)

2.1 2.15 2.2 2.25 2.3 2.35 2.4 2.45 2.53

4

5

Frequency (GHz)

Gai

n (d

Bi)

Figure 5.29: Performance variation due to misalignment between the ground waferand the microstrip wafer (glass substrate). The solid lines represent two combinationswith the largest misalignment (∆xms = ∆yms = ±100µm). The short dashed linerepresents the case of perfect alignment (∆xms = ∆yms = 0µm). The long dashedlines represent combinations of moderate misalignment.

The sensitivity analyses above offered some reassurance in the inherent ability of

this antenna design to withstand some of the expected fabrication tolerances. Thus

far, a simple simulation model of the antenna was used for parametric sweeps and

sensitivity analyses with the aim of saving resources and time. It is important to

109

understand the extent of the change in performance when the fabricated antenna is

placed in a realistic environment during measurement. This is the subject of the next

section.

5.2.5 Modeling for Measurement of Fabricated Antenna

Performing antenna simulations offers antenna designers the opportunity to under-

stand the behaviour of their antennas and the potential pitfalls without committing

to the cost and time in physically building the antennas. However, the simulations

are only useful at predicting accurately the performance of the fabricated antennas

if they are set up properly with features closely resembling those of the antennas in

their measurement environments.

For this particular antenna design, migrating from the simple simulation model

to a more realistic measurement model requires adding the following features, which

are illustrated in Fig. 5.30:

• the microstrip-to-coax transition;

• the antenna holder (made from Lexan, a polycarbonate resin thermoplastic with

ǫr ≈ 3); and

• the alignment marks on the SU8, ground, and microstrip layers.

Each of the above features was first incorporated separately to observe its individ-

ual influence on the antenna performance. In the most realistic model of the antenna

shown in Fig. 5.30, all these features were included. No significant change in the

antenna and the radiation pattern was observed.

The change in AR is plotted in Fig. 5.31. The “Simple” curve represents the result

from the simple simulation model while the “Realistic” curve shows the result with

110

Antenna

holder

Coax connector

25.4mm

175mm

SU8 support posts

& alignment marks for

metal layers

25.4mm

Figure 5.30: Simulation setup for antenna measurement (glass substrate).

all the additional features included. It is obvious that the alignment marks alone

would have minimal effect on the purity of the circularly polarized radiation, since

the “Align. marks” curve overlaps the “Simple” curve. This is expected because

the alignment marks were designed to be small and intentionally placed away from

the ring slot. The presence of the antenna holder would up-shift the frequency of

minimum AR, while the coax-to-microstrip transition would increase the AR. The

combined effect of these features, represented by the realistic model, is a shift in the

111

frequency of minimum AR from 2.36 to 2.38 GHz and an increase of AR from 0.67

to 1.19 dB. The change in AR bandwidth is negligible.

2.3 2.35 2.4 2.450.5

1

1.5

2

2.5

3

3.5

Frequency (GHz)

AR

(dB

)

SimpleAlign. marksHolderCoax connect.Realistic

Figure 5.31: Comparison of various additional features in the simulation model onthe AR performance (glass substrate).

The effect of the additional features on |S11| is less, which is evident in Fig. 5.32.

|S11| actually improves slightly, lowering from –11.6 to –12.4 dB at 2.38 GHz.

This chapter, thus far, has documented the logical and thorough approach used

to design the reconfigurable ring slot antenna, with glass wafers as the substrate and

SU8 photoresist as the channel layer. The following section presents the design of the

antenna, with laminate boards replacing the glass wafers and the photoresist.

112

2.1 2.15 2.2 2.25 2.3 2.35 2.4 2.45 2.5−15

−12.5

−10

−7.5

Frequency (GHz)

|S11

| (dB

)

SimpleAlign. marksHolderCoax connect.Realistic

Figure 5.32: Comparison of various additional features in the simulation model onthe |S11| performance (glass substrate).

5.3 Design Using Laminate Boards

The theory behind the reconfigurable ring slot antenna with switchable polarization

remains the same when the glass wafers and the photoresist are replaced with low-

loss laminate boards (Rogers RO4350B hydrocarbon ceramic laminate, introduced in

Section 4.1). However, because laminate boards have greater physical and processing

flexibility than that of glass wafers, certain structural features are added and modified

to take advantage of this flexibility.

113

5.3.1 Structure

The layers of the antenna and the magnified plan view are shown in Fig. 5.33 and

Fig. 5.34, respectively. The RO4350B laminate board (ǫr = 3.66, tanδ = 0.004) is a

good candidate for replacing the fused-silica wafers because of their similar dielectric

constants, which facilitates the re-optimization of the antenna dimensions. The lam-

inate boards are available in several standard thicknesses, with 30 mil (0.762 mm)

being the closest option for replacing the 1-mm-thick glass substrate.

One noticeable difference between this alternative antenna structure and the al-

ternative structure of the switch described in Section 4.3 is that the combination of

SU8 photoresist (channel) and glass wafer (cover) is replaced with Rexolite plastic

and RO4350B laminate board. As shown in Fig. 4.6 and described in Section 3.3, the

brittle glass cover is susceptible to cracking and the SU8 photoresist is very difficult

to work with.

The combination of Rexolite plastic (ǫr = 2.53, tanδ = 0.0002) and laminate board

offers a better option from the perspective of handling and processing. Rexolite, which

is a cross-linked polystyrene plastic, is chosen because of its consistent permittivity

and low-loss characteristic over a wide range of frequency and its transparency for

allowing visual inspection of the movement of the slugs. The laminate board used

for the channel layer was stripped of the copper cladding on both sides before being

laser-machined.

The glass slugs, described in Sections 4.1 and 5.2, were also replaced with slugs

cut from a laminate board with copper on one side removed. The remaining copper

cladding on the other side of the laminate board did not have the adhesion issues

suffered by the copper-coated glass slugs (Fig. 4.15).

114

Cover tcoverRexolite

Ground & Microstrip tcopperCopper

Substrate tsubLaminate

Slugs tslugChannel tchannelLaminate

Side view

Figure 5.33: Various layers composing the ring slot antenna, with laminate substrate.

The additional features in the structure are the mounting screw holes at the

four corners and the centre screw hole, as shown in Fig. 5.33. As mentioned in

Section 4.3, the various layers of the switch were not bonded permanently during

measurement, but instead were clamped together to allow easy reconfiguration and

adjustments to the layer alignment and channel geometries. A more elegant way to

maintain this flexibility in assembling the antenna while avoiding the cumbersome

clamp (Fig. 4.9) is to hold the layers together with nylon screws, washers, and nuts.

115

ltab

y

x

l

wtab

rhole,cover

rhole,sub

To coax

connector

lsl

Figure 5.34: Magnified plan view of the ring slot antenna with laminate substrateand channel. Note that all four holes shown would be drilled through the cover layer.Only the centre hole would be drilled through the substrate as well.

The holes are strategically positioned away from the ring slot and the microstrip line

such that their influence on the EM behaviour of the antenna should be negligible.

This method of assembly is not feasible with the design using glass wafers because of

the risk of cracking the glass during hole drilling, screw tightening, and the limited

space available with the restrictive 4” wafer size. A larger wafer size (6”) could be

considered, but any device fabricated on a wafer of this size would be very fragile to

handle.

116

Another advantage, which is very important, is the introduction of the centre

screw. It allows for fine adjustment to the channel height and a means to control

the air gap separation (hair) between the slugs and the ground. With the glass

substrate design, the antenna layers would be held together at the edges by the

antenna holder, as shown in Fig. 5.30. The possibility of such adjustment after

assembling the layers together does not exist. Given the symmetry of the ring slot

and relatively insignificant presence of ground current at the centre of the structure

(as observed in Fig. 5.22, Fig. 5.23, and Fig. 5.24), the relatively small hole at the

centre is not expected to alter the EM behaviour of the antenna.

The thicknesses of the layers, as denoted in Fig. 5.33, are listed in Table 5.4.

The antenna was first optimized using a simple simulation model without the four

screw holes at the corners, the coax-to-microstrip transition, and the antenna holder.

Instead of performing the parametric sweep analysis as documented in Section 5.2.2, a

quicker approach was used — optimization with the initial starting dimensions being

those of the glass substrate design (listed in Table 5.2).

Table 5.4: Layer thickness of the ring slot antenna with laminate substrate.

tsub 0.762mm tcover 0.787mmtchannel 0.508mm tslug 0.422mm (plus 35um)tcopper 35um

The lateral dimensions of the optimized design are given in Table 5.4. Some of

the additional and modified features different from the glass substrate design are

specifically labelled in Fig. 5.34.

The dimensions obtained after optimization were then transferred to the realistic

117

Table 5.5: Dimensions of the ring slot antenna with laminate substrate.

ro 15.54mm rstub,sl 4.2mmgring 2mm θstub,sl 140

wslug 3.4mm wsl 1.59mmlslug 6mm lsl 7.35mmwopen 3.27mm rstub,ms 4.85mmlopen 4.14mm θstub,ms 75

gsl 0.4mm wms 1.6mmwchamber wslug + 0.12mm lms 13.05mmlchamber 1.5lslug + lopen wmitre 1.786mmwwall 0.8mm rhole,cover 1mm

wchannel wchamber/2 rhole,sub 1.25mmgchannel 1.5mm ltab 3mmrport 1.5mm wtab 2mm

model, as illustrated in Fig. 5.35, to assess the influence of the additional features

present in the fabricated antenna during measurement. Since the ring slot would be

patterned on the ground plane using either laser machining, milling, or photoetching,

as opposed to using the process of photolithography in the cleanroom, the shape of

the substrate is not restricted to circular and the size is not restricted to 4” or 6”.

As shown in Fig. 5.35, the substrate is square and it is larger than the glass wafers,

which allows the corner screw holes and the antenna holders to be further away from

the ring slot. The antenna holder was designed to accommodate a larger substrate

by expanding the distance between the arms.

5.3.2 Simulated Performance

The simulated antenna performance, for both the simple and realistic models, is given

in Fig. 5.36, Fig. 5.37, and Fig. 5.38.

From Fig. 5.36, it is evident that the additional features in the realistic model do

118

Coax

connector

Antenna

holder

Nylon screw

through hole

in substrate225mm

136mm

100mm

Figure 5.35: Simulation setup for antenna measurement (laminate substrate).

not have a significant effect on the return loss of the antenna, as in the case of the

glass substrate design. In fact, the impedance matching demonstrated in the realistic

model is slightly better than the simple model.

The discrepancies in performance are more noticeable with the AR and the gain.

As in the case of the glass substrate design, the frequency of minimum AR is shifted

up in the realistic model, which is evident in Fig. 5.37. However, the minimum

AR achieved remains similar if not slightly better for the realistic model. The AR

bandwidths shift accordingly. The appropriate operating frequency for this design

would be 2.43 GHz, compared to 2.38 GHz in the glass substrate design. However,

the minimum AR achieved remains similar, if not slightly better for the realistic

model.

119

2.1 2.2 2.3 2.4 2.5 2.6 2.7−25

−20

−15

−10

−5

0

Frequency (GHz)

|S11

| (dB

)

LP (simple)RHCP (simple)LHCP (simple)LP (realistic)RHCP (realistic)LHCP (realistic)

Figure 5.36: |S11| of the simulated ring slot antenna (laminate substrate) under dif-ferent polarization states, for both simple and realistic models.

The broadside gain of the antenna in all three polarization states, as plotted in

Fig. 5.38, is also similar to that of the glass substrate design. A gain reduction of

about 1 dB and a slight up-shift in the frequency of maximum gain is observed for

the realistic model. The coax-to-microstrip transition can be attributed as one of the

major causes of this gain reduction when comparing the simple model against the

realistic model. This is evident in the gain comparison in Fig. 5.39, which shows the

changes in gain due to the addition of the antenna holder and the addition of the

coax connector separately.

120

2.3 2.35 2.4 2.45 2.5 2.550

0.5

1

1.5

2

2.5

3

3.5

4

4.5

5

Frequency (GHz)

AR

(dB

)

RHCP (simple)LHCP (simple)RHCP (realistic)LHCP (realistic)

Figure 5.37: Broadside AR of the simulated ring slot antenna (laminate substrate)over a range of frequency, for both simple and realistic models.

The simulated antenna radiation efficiency, using the realistic model, is 93.7%. As

mentioned in Section 2.1.2, radiation efficiency accounts for the dissipative losses of

the antenna (which are mainly conductor and dielectric losses in this case). The high

efficiency was expected because low-loss laminate was chosen as the substrate and the

copper thickness of 35 µm equated to over 25 skin depths at 2.4 GHz. This confirmed

the low-loss characteristics of this scheme of reconfigurable antenna as demonstrated

in the low insertion loss of the switch in Chapter 4.

The radiation pattern and the ground current of the antenna operating in the

121

2.1 2.2 2.3 2.4 2.5 2.6 2.72

2.5

3

3.5

4

4.5

5

5.5

6

Frequency (GHz)

Gai

n (d

Bi)

LP (simple)RHCP (simple)LHCP (simple)LP (realistic)RHCP (realistic)LHCP (realistic)

Figure 5.38: Broadside gain of the simulated ring slot antenna (laminate substrate)over a range of frequency, for both simple and realistic models.

RHCP state at 2.4 GHz are provided in Fig. 5.40 and Fig. 5.41, respectively.

A summary of the simulated antenna performance for the realistic model is given

in Table 5.6.

It should be noted that the antenna does suffer from the inability to align the

impedance bandwidth with the AR bandwidth, a goal which is very difficult to achieve

given the objective of attaining a reconfigurable antenna capable of all three polar-

ization states (as mentioned in Section 2.4). The limiting factors here are the upper

limit of the impedance bandwidth imposed by the LP state and the lower limit of the

122

2.1 2.2 2.3 2.4 2.5 2.6 2.72

2.5

3

3.5

4

4.5

5

5.5

6

Frequency (GHz)

Gai

n (d

Bi)

SimpleHolderCoax connect.Realistic

Figure 5.39: Comparison of various additional features in the simulation model ongain performance (laminate substrate) for the RHCP state.

AR bandwidth when operating in the LHCP state, as shown in Fig. 5.42.

The ideal scenario is to have the overall impedance bandwidth of the antenna

encompassing the overall AR bandwidth. When considering the antenna as a recon-

figurable antenna with only two polarization states — RHCP and LHCP without LP,

the performance is very respectable, since the impedance bandwidths of RHCP and

LHCP have very high upper limits that reach far beyond 2.7 GHz and encompass the

entire AR bandwidths. However, when the antenna is subject to the more stringent

bandwidth standard of a reconfigurable antenna capable of all three states — i.e.

the overall impedance bandwidth of the antenna is defined as the frequency band at

123

Figure 5.40: Simulated antenna pattern for φ = 0 and φ = 90 planes when operatingin the RHCP state at 2.4 GHz (laminate substrate).

124

Table 5.6: Simulated performance (realistic model) of the ring slot antenna usinglaminate substrate.

Impedance bandwidth 2.21 to 2.44 GHz (9.5%)AR bandwidth (RHCP) 2.37 to 2.49 GHz (4.9%)AR bandwidth (LHCP) 2.39 to 2.51 GHz (4.9%)Minimum AR (RHCP) 0.39 dBMinimum AR (LHCP) 0.67 dB

Frequency of min. AR (RHCP) 2.43 GHzFrequency of min. AR (LHCP) 2.45 GHz

Broadside gain 4.6 dBiBroadside cross-pol. level < −20 dB

Antenna radiation efficiency 93.7%

which |S11| remains below −10 dB regardless of the polarization state, it is clear that

this particular antenna cannot claim as great of a performance. In the strictest sense,

the simulated overall bandwidth of this reconfigurable antenna — capable of three

polarization states (LP/RHCP/LHCP) — is 39 MHz (2.394 to 2.433 GHz), which is

about 1.6%, as indicated in Fig. 5.42.

125

0°

90° 135°

45°

Figure 5.41: Ground current at different phases of a cycle in the RHCP state (laminatesubstrate).

126

2.15 2.2 2.25 2.3 2.35 2.4 2.45 2.5 2.55 2.6 2.65−19

−16

−13

−10

−7

−4

|S11

| (dB

)

2.15 2.2 2.25 2.3 2.35 2.4 2.45 2.5 2.55 2.6 2.650

1

2

3

4

5

AR

(dB

)Freq (GHz)

|S11|, LP|S11|, RHCP|S11|, LHCPAR, RHCPAR, LHCP

3−stateBW

Figure 5.42: Comparison of impedance and AR bandwidths (laminate substrate,realistic model).

127

5.4 Design Advantages and Disadvantages

With both substrate designs capable of producing reasonable simulated antenna per-

formance, it is important to evaluate their merits and drawbacks in terms of man-

ufacturability and ease of handling during assembly. The advantages of the design

using laminate boards include:

• Conventional materials, used in typical PCB designs, lead to greater predictabil-

ity of their mechanical behaviour during processing.

• With conventional materials, the overall cost of manufacturing should be less,

since Rexolite plastic and laminate boards are generally cheaper than glass

wafers and SU8 photoresist, and less effort is required in tweaking parameters

and steps of the fabrication process.

• Laminate is not brittle like glass, which makes it much easier to handle during

the assembly of the antenna. The antenna is also more robust as a result.

• Ground plane (antenna size) is not bounded by the standard 4” or 6 ” glass

wafer size.

• Channel thickness can be greater than 200 µm.

• Misalignment between the ring slot and microstrip is smaller and more pre-

dictable since the two layers are on the top and bottom sides of the same piece

of substrate instead of two separate glass wafers.

• Copper layers are thicker, and thus reducing metal loss.

• No copper adhesion issue.

• Alternative manufacturing methods, such as laser machining, screen printing,

photoetching, or milling, can be used depending on precision required and cost.

• Centre screw allows for fine adjustment to the channel height.

128

Despite the advantages listed above, there are some disadvantages associated with

this alternative design. They are as follows:

• The channel is patterned and then adhered to the cover, which increases the

possibility of distorting the shape of channel pattern.

• Quality of the channel wall depends strongly on the manufacturing method

chosen.

• The inherent variability in thickness and permittivity of the laminate boards

is larger than that of the glass wafers (reflected in the price difference between

the two materials).

The evaluation above shows that the design using laminate boards is more viable

and cost-effective when compared to using glass wafers. With the reconfigurable

antenna designed and its performance predicted using simulations, the fabrication

result using both materials is discussed in Chapter 6, along with a comparison between

simulated and measured performance of the antenna.

129

Chapter 6

Fabrication and Measurement of

Ring Slot Antenna

This chapter is divided into two parts. The first part documents the results of fab-

ricating the antenna using the glass and the laminate substrates. Some of the issues

encountered during fabrication are discussed. The setup used for testing the pneu-

matic control of the slugs and measuring the pressure required to move the slugs is

given. The second part provides a brief description of the antenna measurement setup

and an in-depth discussion on the measured results. The figures of merit — return

loss, AR, and gain — as well as the antenna patterns will be compared against the

simulated results in Chapter 5. Substantial part of the work in this chapter has been

published in [59], [60], and [61].

6.1 Fabrication

Referring to the evaluation in Section 5.4, using glass wafers as the substrate and

the cover layer offers greater accuracy at the expense of greater fabrication cost and

difficulty in handling. The fabrication results shown below offers convincing evidence

for the hypothesis.

130

6.1.1 Glass Wafers as Substrate

As mentioned in Section 4.1, the substrate, slugs, and covered channel can be coated

simultaneously without the requirement of building one layer on top of another. Each

of these components was fabricated by the author in AMIF.

SU8 photoresist was patterned on a glass wafer to create the channel on a cover,

which is the same process as described in Section 4.1. Fig. 6.1 illustrates the fragility

of the glass wafer and the recurring adhesion issue of the SU8 resist on glass. The

perimeter of the wafer, which was not supported by the cluster of square posts,

cracked during the dicing process. Some of the square posts drifted away from their

original positions due to delamination. Nevertheless, the precisely patterned channel

wall exhibited the advantage of the photolithography process.

Holes for

access ports

Broken

edges of

wafer

Diced edges

of waferPhotoresist

delamination

2mm

Figure 6.1: Fabricated channel layer on glass wafer.

131

The holes on the glass wafer, through which pressurized air would be applied, were

produced with laser machining instead of drilling (recall the poor result in Fig. 4.6).

No issue was experienced using laser machining because the stress on the glass wafer

associated with this manufacturing technique was minimal. The only disadvantage

compared to drilling is the longer time required to laser-machine holes through 0.5 mm

thick glass.

Copper with a thickness of 5 µm was deposited on a glass wafer using evaporation

deposition instead of sputtering deposition (refer to Section 4.3) with the intention

of improving the adhesion of copper on glass. Unfortunately, the effort proved to be

futile, as shown in Fig. 6.2, with the delamination of copper occurring after patterning

the ring slot on the layer.

5 µm of copper deposited on glass using evaporation or sputtering was relatively

thick for this technique and suffered from significant thermal stress (even with the

addition of titanium as adhesion layer in between copper and glass). Thinner copper

is not desirable because it would lead to greater metal loss due to the relatively

high skin depth at the frequency of interest (skin depth is 1.33 µm at 2.4 GHz).1

Electroplating is a promising alternative process for realizing thick layer of copper on

glass, which could be investigated in the future if the fabrication of the antenna using

glass substrate were to be explored further.

Although the process of photolithography yielded very precise patterns (precision

of 3 µm for copper), the complexity and the steps involved led to a number of issues

that could not be eliminated entirely, even with repeated trials. Compounded with

the inherent fragility of glass wafers, fabricating the antenna using a glass cover and

1Copper thickness of at least several multiples of skin depth is deemed necessary to keep metalloss of the antenna reasonably low.

132

Figure 6.2: Delamination of copper after etching.

substrate for 2.4 GHz operation was deemed nonviable. Handling 4” glass wafers

that are only 0.5 mm thick during antenna measurements would also likely lead to

more issues. For future work, scaling down the overall size of the antenna would be

the right way forward for this particular fabrication method, which would require

applying this technology at a higher frequency of operation.

In the next section, the antenna fabricated using laminate substrate, which was

predicted to be more viable and cost effective in Section 5.4, is presented.

133

6.1.2 Laminate Boards as Substrate

The fabricated antenna is shown in Fig. 6.3. The method of laser-scribing the out-

lines of the ring slot and the microstrip line on 35-µm-thick copper layers achieved a

precision within 10 µm, which is better than the precision by using photolithography

for this thickness of material. Such a high precision was more than adequate for this

antenna operating at 2.4 GHz.

RHCP port

LHCP port

LP port

slugs

Figure 6.3: Fabricated antenna with laminate substrate (ground plane side). Slugs,visible through the cover layer, are positioned for operating in the RHCP state. Withthe cover layer removed, inset shows the ground plane with the ring slot, slugs, andchannel.

134

The channel layer was also produced via laser machining. The precision was found

to be in the range of 30 µm. A small amount of lateral deformation in the shape of

the channel was observed, which was due to the thin structure of the channel. This

small deformation of the channel layer became negligible after the channel was aligned

and glued onto the ground plane. Referring to the result of the sensitivity analysis in

Fig. 5.28, the antenna performance of the device was expected to have a reasonable

tolerance to channel misalignment and discrepancy in patterning the channel.

Fig. 6.4 illustrates the backside of the antenna, on which the microstrip feedline

was patterned. The laminate substrate was thin enough to reveal the ring slot when

light was shone from the opposite side.

Figure 6.4: Fabricated antenna with laminate substrate (microstrip side). Light isbeing shone through the ring slot from the opposite side.

The profile of the channel was examined more closely under a calibrated optical

microscope, and the magnified view, shown in Fig. 6.5, demonstrates the accurate

135

transfer from the design pattern to the fabricated channel achieved by using laser

machining. The slugs were also cut from a laminate sheet using laser machining.

3.52mm

3.4mm0.8mm

Channel wall

SlugSlug

Channel wall

Figure 6.5: View under the microscope of a slug resting in the channel.

The three different combinations of slug positions for the three polarization states

are shown in Fig. 6.6, which are identical to those in Fig. 5.2, Fig. 5.3, and Fig. 5.4.

To clearly show the positions of the slugs, the cover layer was removed for illustration

purposes. When the various layers of the antenna were assembled together, the slugs

remained visible through the translucent cover layer, as seen in Fig. 6.3.

RHCP LHCP LP

Figure 6.6: Positions of slugs for different polarization states. The cover layer wasremoved.

136

In practice, the slugs would be moved into position by the pressure gradient gen-

erated from the application of pressurized air at the appropriate port. The pneumatic

actuation test is discussed in the next section.

6.1.3 Pneumatic Control and Pressure Measurement

Preliminary tests of the three-port device shown in Fig. 6.3 revealed an issue with the

hypothesized air flow described in Fig. 5.2, Fig. 5.3, and Fig. 5.4. In each scenario,

it was found that the two air-flow exits closer to the air-flow input port allowed too

much air to escape the channel. The remaining air flow inside the channel past the

two exits did not sustain enough pressure to move the two slugs further away from

the input port. Referring to Fig. 5.2 for the LP case, there was difficulty in reliably

moving the two slugs on the right side. In Fig. 5.3, it was the two slugs at the bottom

half of the channel.

A modification was made to the structure to resolve this issue, which did not affect

the electromagnetic behaviour of the antenna. A fourth port was added to the cover

layer, as shown in Fig. 6.7. 90 elbows and flexible pneumatic tubes were attached to

the top of the cover layer.

The influence of the pneumatic test setup (i.e. the port connectors and the tubes)

on the antenna performance was not expected to be significant because the compo-

nents were not of high permittivity and were not directly adjacent to the ground

plane. In future designs, the port connectors and the tubes will be replaced with

channels entirely embedded between the ground plane and the cover layer, such that

the connections to the pneumatic source will be away from the ring slot. This will

bring the influence of the pneumatic actuation setup on the antenna performance to

137

3

4

1

2

Actuation SequenceLP: i) 1&3, ii) 4LHCP: i) 2&4, ii) 3RHCP: i) 2&4, ii) 1

Additional

port

Figure 6.7: Setup for the pneumatic control test of the antenna and the actuationsequence of air inflow for each polarization state. Current slug positions correspondto the LHCP state.

a minimum.

Besides the physical modification to the antenna structure, the actuation was

changed from a one-step action to a two-step sequence, as illustrated in Fig. 6.8. This

increased the actuation time, but ensured the reliable control of the slug positions.

Since this scheme of reconfiguration is not designed for high-speed applications, as

mentioned in Section 4.5, the increase in the actuation time due to the extra step is

not considered a disadvantage. With the use of solenoid valves to control the source

of inflow, the actuation time could be decreased to under 500 ms.

138

LP

RHCP

LHCP

air in

air out

yx

1 2

1 2

1 2

(a)

(b)

(c)

(d)

Figure 6.8: Two-step sequence of air input and slug movement for realizing (a) LP,(b) LHCP, and (c) RHCP states. Slugs and chambers are highlighted for clarity. Thecover was removed (d) to clearly show the slug positions in each step of the sequenceto arrive at the RHCP state.

139

The pressure required to move the slugs was measured. The measurement setup

was similar to that shown in Fig. 4.16. It was found that the pressure required

was less than 2 kPa (0.3 psi) with the antenna placed horizontally (i.e. its plane

perpendicular to gravity). This was half of the pressure measured for the switch in

Section 4.4. The main reason for less pressure required for the antenna is the use of

the Rexolite cover and laminate slugs instead of a glass cover and slug. The stiction

between glass surfaces is higher and this was observed in the case of the switch with

glass cover and slug, when 4 kPa was required to move the slug.

The pneumatic control of the laminate-based antenna was deemed successful af-

ter the addition of the fourth port for air inflow and the change from a one-step

actuation to a two-step actuation. Section 6.2 documents the investigation of the

electromagnetic behaviour of the antenna and its measured performance.

6.2 Antenna Measurements

In order to evaluate the performance of the fabricated ring slot antenna and compare

it against the results obtained from simulations in Chapter 5, measurements were

performed inside an anechoic chamber. The anechoic chamber provides the necessary

environment to characterize an antenna under test (AUT) with minimal external in-

terfering signals and multipath errors. The antenna measurement setup is illustrated

in Fig. 6.9.

A standard-gain pyramidal-horn antenna (model 3160-03) by ETS Lindgren [10],

with its antenna gain known for its operating range from 1.7 GHz to 2.6 GHz, was

set up at one end of the chamber and aligned with the ring slot antenna (the AUT)

at the opposite end of the chamber. The two antennas were separated by a distance

140

Standard

Gain HornAntenna Under

Test (AUT)

Turntable

Absorbing Cones

Figure 6.9: Conceptual diagram of the antenna measurement setup inside the anechoicchamber.

of 3.1 m, which was sufficient to keep the antennas at a far-field distance of each

other. Fig. 6.10 shows the two antennas inside the chamber. Additional absorbing

material was attached to the arm of the turntable on which the AUT was mounted

to minimize the reflection and scattering of signals that could distort the measured

results.

Standard

Gain Horn

AUT (Ring Slot Antenna)

on Turntable

Figure 6.10: Standard gain horn and the ring slot antenna (AUT).

Since the turntable rotated in the azimuth direction only, in order to obtain the

141

antenna patterns in both φ = 0 and φ = 90 planes, remounting the AUT with

an orthogonal rotation was necessary, as depicted in Fig. 6.11. Detailed procedure

and description of the antenna measurements in the anechoic chamber is provided in

Appendix A.

x

y

x

yφ = 90°

φ = 0°

φ = 0°

φ = 90°

Figure 6.11: Setup of the ring slot antenna on the turntable for scanning φ = 0 (left)and φ = 90 (right) planes.

Since the goal of the antenna measurements was to evaluate the RF performance

of the antenna, the dynamic nature of reconfiguration (i.e. pneumatic actuation)

was not tested simultaneously during the antenna measurements in order to focus

on understanding the electromagnetic behaviour of the antenna. Therefore, the slugs

were temporarily secured in their designated positions for each state of configuration

(LP/RHCP/LHCP) during each measurement to ensure that none of them would

move out of position. An alternative setup, which would be a more realistic scenario,

is to supply a constant pneumatic pressure gradient (i.e. a constant stream of air flow

from the corresponding port) during each measurement. This is an area of future

work. Testing of the pneumatic control was performed separately and is presented in

142

Section 6.1.3.

6.2.1 Antenna Patterns

The measured antenna patterns (co-polarization and cross-polarization) for both φ =

0 and φ = 90 planes when the antenna was configured for the LP state are given

in Fig. 6.12. These are gain patterns and are given in the unit of dBi. The patterns

are a close match to the simulated patterns, except for the cross-polarization in the

φ = 90 plane. However, the cross-polarization level remains below –20 dB in almost

all directions in the φ = 90 plane, which is very respectable.

−20

−10

0

10

−60

120

−30

150

0

180

30

−150

60

−120

90 −90

φ=0° plane

θ (degrees)

Gai

n (d

Bi)

−20

−10

0

10

−60

120

−30

150

0

180

30

−150

60

−120

90 −90

φ=90° plane

θ (degrees)

Gai

n (d

Bi)

co−pol,sim.co−pol,meas.x−pol,sim.x−pol,meas.

Figure 6.12: Comparison between measured and simulated antenna patterns for φ =0 and φ = 90 planes when operating in the LP state at 2.4 GHz.

For obtaining the circular polarization patterns of the antenna when it was con-

figured for the RHCP or LHCP state, it was possible to use the same standard gain

horn in Fig. 6.10, which is linearly polarized. Ideally, for each CP pattern in a par-

143

ticular plane, a CP measurement is performed by using a standard antenna that can

generate a CP signal or by using a spinning dipole antenna. In this case, where only

a linearly polarized standard antenna was available, two separate linear polarization

measurements, with the horn oriented orthogonally in each measurement, were re-

quired to obtain one CP pattern. One set of the measured data was mathematically

phase-shifted by 90 before the two sets of measured results were either combined or

subtracted to obtain the RHCP or LHCP pattern, respectively [8]. Detailed steps

and derivation of this procedure are given in Appendix A.2.

When the antenna was configured to operate in the RHCP and LHCP states,

the measured patterns were consistent with the simulated patterns, as illustrated in

Fig. 6.13 (RHCP) and Fig. 6.14 (LHCP).

−20

−10

0

10

−60

120

−30

150

0

180

30

−150

60

−120

90 −90

φ=0° plane

θ (degrees)

Gai

n (d

Bi)

−20

−10

0

10

−60

120

−30

150

0

180

30

−150

60

−120

90 −90

φ=90° plane

θ (degrees)

Gai

n (d

Bi)

LHCP,sim.LHCP,meas.RHCP,sim.RHCP,meas.

Figure 6.13: Comparison between measured and simulated antenna patterns for φ =0 and φ = 90 planes when operating in the RHCP state at 2.4 GHz.

It should be noted that in Fig. 6.13 the measured LHCP pattern in the φ = 90

144

−20

−10

0

10

−60

120

−30

150

0

180

30

−150

60

−120

90 −90

φ=0° plane

θ (degrees)

Gai

n (d

Bi)

−20

−10

0

10

−60

120

−30

150

0

180

30

−150

60

−120

90 −90

φ=90° plane

θ (degrees)

Gai

n (d

Bi)

LHCP,sim.LHCP,meas.RHCP,sim.RHCP,meas.

Figure 6.14: Comparison between measured and simulated antenna patterns for φ =0 and φ = 90 planes when operating in the LHCP state at 2.4 GHz.

plane did deviate noticeably from the simulated pattern from θ = −90 to −60 and

from θ = −45 to 0. Also, the measured LHCP pattern in the φ = 90 plane from

θ = −150 to −90 in Fig. 6.14 also demonstrated noticeable deviation. Repeated

measurements confirmed that these deviations from the simulated patterns were not

the result of inconsistent measurements. These deviations were most likely caused by

145

the following:

• inherent measurement errors due to multipath signals in the asymmetric cham-

ber;

• the presence of the coax connector near the direct path between the horn an-

tenna and the AUT when the rotation of the turntable was around θ = −90

during the φ = 90 plane scan (see the right diagram in Fig. 6.11), which could

lead to unwanted scattering of signals; and

• the inherent characteristics of the antenna due to fabrication tolerances.

Nevertheless, the simulated gain values in that range of θ values were from −5 to

−20 dBi, where the accuracy of measurements was not expected to be as good as in

other regions. Overall, the measured patterns confirmed that the fabricated antenna

behaved as expected under different polarization states.

6.2.2 Return Loss (−|S11|)

The measured return loss of the antenna when operating in each polarization state

is compared against the simulated value in Fig. 6.15. The measured results were

quite consistent with those from simulations, except for the 2.5 to 3 dB discrepancy

beginning at 2.35 GHz for the CP states.

The phenomenon, which was not fully understood, could be attributed to the

soldering connection of the coax connector to the microstrip line, the surface con-

tact between the slugs and the ground plane, slight permittivity anisotropy in the

substrate, slight curvature in the substrate, or a combination of the aforementioned

factors. Different simulations were run to test these hypotheses, however, none pro-

duced return loss that would match up well with the measured results over the entire

146

2.1 2.2 2.3 2.4 2.5 2.6 2.7−25

−20

−15

−10

−5

0

Frequency (GHz)

|S11

| (dB

)

LP (meas.)RHCP (meas.)LHCP (meas.)LP (sim.)RHCP (sim.)LHCP (sim.)

Figure 6.15: Comparison between measured and simulated |S11| of the ring slot an-tenna under different polarization states.

frequency range. Nevertheless, |S11| for the CP states remained below or at −10 dB

above 2.24 GHz, which kept the measured impedance bandwidth of each polarization

state the same as the simulated bandwidth.

As mentioned in Section 6.1.3, the presence of the pneumatic port connectors

and tubes was expected to have an insignificant effect on the antenna performance.

Fig. 6.16, which shows the measured |S11| of the antenna with and without the pneu-

matic actuation setup, confirmed that the influence of these components was negligi-

ble. Note that these plots are slightly different from the measured |S11| in Fig. 6.15

because the coax connector had to be re-soldered onto the substrate due to a handling

147

error when connecting the cable during one of the antenna measurements.

2.1 2.2 2.3 2.4 2.5 2.6 2.7−25

−20

−15

−10

−5

0

Frequency (GHz)

|S11

| (dB

)

LP (without)RHCP (without)LHCP (without)LP (with)RHCP (with)LHCP (with)

Figure 6.16: Comparison between measured |S11| with and without pneumatic actu-ation setup.

6.2.3 Axial Ratio (AR)

The measured and simulated broadside AR for both CP states are given in Fig. 6.17.

The method of measurements and data analysis is described in greater detail in Ap-

pendix A.2.

Referring to Fig. 6.17 for the comparison with the simulated AR, the frequencies

at which the minimum AR were recorded shifted to the left by 20 MHz for the

148

2.3 2.35 2.4 2.45 2.5 2.550

0.5

1

1.5

2

2.5

3

3.5

4

4.5

5

Frequency (GHz)

AR

(dB

)

RHCP (meas.)LHCP (meas.)RHCP (sim.)LHCP (sim.)

Figure 6.17: Comparison between measured and simulated broadside AR of the ringslot antenna under different polarization states.

RHCP state, and 10 MHz for the LHCP state. These shifts were less than 0.8% of

the operating frequency. Though the measured AR values did not reach as low as

in simulations, the 3-dB AR bandwidths were 5% for both CP states, which were

consistent with the simulated results in Table 5.6.

6.2.4 Gain

The measured broadside gain values in different polarization states are compared

against the simulated values in Fig. 6.18. The results show great consistency with

simulations, with discrepancy no more than 0.5 dB in almost the entire measured

149

range. It is also worth noting that the measured gain values were at their maxima

and remained mostly flat within the measured AR bandwidths (2.35 to 2.5 GHz).

2.1 2.2 2.3 2.4 2.5 2.6 2.72

2.5

3

3.5

4

4.5

5

Frequency (GHz)

Gai

n (d

Bi)

LP (meas.)RHCP (meas.)LHCP (meas.)LP (sim.)RHCP (sim.)LHCP (sim.)

Figure 6.18: Comparison between measured and simulated broadside gain of the ringslot antenna under different polarization states.

Finally, to verify the low-loss characteristic of the antenna, the radiation efficiency

was measured using the improved Wheeler Cap method [63], [64]. More details on the

measurement setup and procedure can be found in Appendix D. The measured radi-

ation efficiency was 92.1% at 2.4 GHz, which compares very well with the simulated

value of 93.7%.

Determining the phase centre of the antenna and its variation from one polar-

ization state to another is an interesting topic of future work. The location of the

antenna phase centre and its variation play an important role in high-precision GNSS

150

ranging applications and other engineering applications in the areas of imaging and

antenna arrays [65].

6.2.5 Performance Summary

A summary of the measured antenna performance, compared to the performance of

the realistic simulation model, is given in Table 6.1.

Table 6.1: Measured and simulated performance of the ring slot antenna.

Criteria Simulation Measurement

Impedance BW 2.21 to 2.44 GHz (9.5%) 2.24 to 2.45 GHz (8.8%)AR BW (RHCP) 2.37 to 2.49 GHz (4.9%) 2.35 to 2.47 GHz (5%)AR BW (LHCP) 2.39 to 2.51 GHz (4.9%) 2.385 to 2.505 GHz (5%)

Minimum AR (RHCP) 0.39 dB 0.65 dBMinimum AR (LHCP) 0.67 dB 1.25 dB

Freq. of Min. AR (RHCP) 2.43 GHz 2.41 GHzFreq. of Min. AR (LHCP) 2.45 GHz 2.44 GHz

Broadside Gain 4.6 dBi 4.8 (LP) / 4.4 (CP) dBiBroadside Cross-pol. level < −20 dB < −16 dB

Rad. Efficiency 93.7% 92.1%

The comparison between the impedance and AR bandwidths, similar to the sim-

ulated one made in Fig. 5.42, is provided in Fig. 6.19. The overall bandwidth of the

fabricated reconfigurable antenna, under the strictest definition of bandwidth (see

Section 2.4), is 58 MHz (2.386 to 2.444 GHz) or equivalently, 2.4% given the oper-

ating frequency of 2.4 GHz. This is larger than the simulated bandwidth of 1.6%

mentioned previously in Section 5.3.2.

Table 6.2 compares the measured performance of this antenna with the existing

antennas in literature described in Section 2.4. The existing antennas utilized PIN

151

−19

−16

−13

−10

−7

−4|S

11| (

dB)

2.15 2.2 2.25 2.3 2.35 2.4 2.45 2.5 2.55 2.6 2.650

1

2

3

4

5

AR

(dB

)

Freq (GHz)

|S11|, LP|S11|, RHCP|S11|, LHCPAR, RHCPAR, LHCP

3−stateBW

Figure 6.19: Comparison of measured impedance and AR bandwidths.

diodes, which are an established and commercially available technology.

In this work, a novel scheme of reconfiguration was implemented, and the mea-

sured antenna performance was similar, if not slightly better, when considering the

AR and overall three-state bandwidths. More importantly, the antenna developed in

this work was designed to switch between three polarization states, which increased

the realization challenge and the level of difficulty in attaining the respectable per-

formance. The two existing antennas capable of three or more polarization states

either had AR and impedance bandwidths that did not overlap, which resulted in a

0% three-state bandwidth [18], or required a thick air layer between conducting layers

to broaden its AR and impedance bandwidths in order to guarantee an overlap [19].

The antenna presented here attained a respectable 2.4% three-state bandwidth with

152

a reasonable device thickness of 2.1 mm.

Table 6.2: Comparison of antenna performance with similar reconfigurable antennasin literature.

Criteria [1] [53] [18] [19] This Work

# of pol. states 2 2 3 4 3Thickness2 0.8 mm 1.6 mm 1.6 mm 8.2 mm 2.1 mm|S11| BW 37.5% 20.9% 1.17% 13.1% 8.8%

AR BW (LH) 4.3% 4.2% 0.74% 4.5% 5%AR BW (RH) 3.4% 4.1% 0.72% 2.2% 5%Min. AR (LH) 0.5 dB 1.0 dB 0.9 dB 1.0 dB 1.25 dBMin. AR (RH) 1.2 dB 1.25 dB 0.5 dB 0.8 dB 0.65 dBOperating Freq. 2.38 GHz 2.63 GHz 2.45 GHz 2.45 GHz 2.42 GHzBroadside Gain 4 dBi 3.1 dBi 5.85 dBi 5.87 dBi 4.4 dBi3-state BW N/A N/A 0% 2.2% 2.4%

2Thickness accounts for the dielectric layers of the antenna structure and excludes support

structures such as measurement holders.

153

Chapter 7

Conclusion

Amongst the many options available to realize RF reconfigurable devices, the preva-

lent use of DC-controlled devices (varactor diodes, PIN diodes, FETs, and MEMS

switches) is a result of their ease of use, fast switching speed, and the vast number of

commercially available products. However, there are trade-offs in power handling ca-

pability, loss characteristics, and linearity. The main disadvantage of DC-controlled

devices is the increasing complexity in arranging the biasing structures associated

with an increasing number of devices used. Careful design is required as the bias-

ing structures compete with RF structures for limited area on the conducting layer.

Compromises in the RF functionality and performance are often necessary as the

metalized biasing structures do lead to a certain amount of loss through dissipation

and undesired radiation. It is for these reasons that there is continual exploration

on alternative schemes of reconfiguration — from the use of liquid metals, liquid

crystals, and high-permittivity mixtures, to the implementation of pneumatically ac-

tuated membranes. Each methodology boasts its own advantages and disadvantages,

and its selection over another is dependent on the application.

The scheme of reconfiguration using pneumatic control of solid dielectric slugs

was introduced in this work, which serves as an alternative to the aforementioned

schemes. The concept of this scheme is simple: a copper-backed dielectric slug, when

154

bridging over a narrow gap on the conducting layer of the substrate, provides the

necessary capacitive coupling required to effectively close the gap.

An evaluation of this technology is summarized below. The advantages are:

• Simple concept;

• Low loss, since low-loss materials (laminate boards) were used;

• Different materials and fabrication methods can potentially be adopted;

• Mass production possible;

• Good repeatability;

• High linearity and immunity to RF distortion due to the absence of semicon-

ductor materials;

• Absence of DC biasing structures, replaced by pneumatic actuation components.

This methodology does suffer from the following limitations:

• Slow switching speed (tens to hundreds of milliseconds), which excludes imple-

mentation in high-speed applications in its current form;

• Relatively large footprint of conducting layer required for capacitive coupling,

which makes this technology more suitable for reconfigurable antennas given

the large expanse of conductor already available;

• Integration of pneumatic components, which is more feasible for miniatur-

ized structures since commercially available micro-valves and pumps (see Ap-

pendix B) can be utilized.

• Requirement to secure the slugs against gravity and vibration, which is ad-

dressed in Section 7.2.2).

In this project, the scheme was brought from the concept stage to the significant

milestone of successfully fabricating two practical reconfigurable RF devices.

155

7.1 Contributions

1. A novel scheme of reconfiguration was introduced (Chapter 3) and successfully

demonstrated with the fabrication of a capacitive switch and an antenna with

switchable polarization. A review of the current landscape on the technologies

of reconfiguration was provided in Chapter 2 and how the scheme developed in

this project compares under different criteria was given in Section 4.5.

2. A custom process for patterning the pneumatic channel layer using SU8 pho-

toresist was developed by the author (Section 3.3 and Appendix C). Though the

antenna designed for applying this process on glass substrate was not successful,

the process developed will be useful for the miniaturization of future designs at

higher operating frequencies.

3. The capacitive switch described in Chapter 4, which offered the first physical

demonstration of the viability of pneumatic control for a reconfigurable RF

device, was presented in [56] and published in [57].

4. The design and fabrication of the reconfigurable ring slot antenna was presented

in [59] and [60], and resulted in an accepted journal publication [61]. The

measured performance of the antenna was comparable to some of the notable

designs found in literature, and the antenna showed better performance in AR

and three-state bandwidths, as summarized in Section 6.2.5. This work was

awarded second place in the Student Paper Competition in the 2014 IEEE

International Symposium on Antennas and Propagation in Memphis, USA [60].

5. This project has also led to two other conference presentations [66],[67].

156

7.2 Future Work

7.2.1 Further Characterization of Switching Mechanism

The main project goal of implementing this pneumatic slug control technology in a

practical RF device has been demonstrated. Additional characterization will be im-

portant for gaining greater understanding of this switching mechanism, as mentioned

in Section 4.5. It will offer a more comprehensive comparison against other existing

reconfiguration technologies, which can serve as a guide in determining the suitabil-

ity of this technology for a particular application. The following attributes will be

considered:

• Number of switching cycles before failure;

• Limits of operating temperature and the effects of thermal expansion;

• Power handling capability; and

• Precise profiling of the surface contact between the slug and the metal patch.

7.2.2 Slug Position Control

Securing the positions of the slugs after switching is a topic to be addressed in the

future development of this technology. This is important for enhancing the versatility

of a reconfigurable device since it may be subjected to physical movement and rota-

tion. Given the current dimensions of the reconfigurable antenna, the gravitational

force on a slug is much stronger than any electrostatic force associated with stiction.

When the antenna is not positioned with its plane perpendicular to the force

of gravity, one method of ensuring that the slugs remain in their positions is the

continued presence of pressure gradient (supply of pressurized air). As discussed in

157

Section 4.4, the holding pressure required is noticeably less than the actuation pres-

sure. However, an orthogonal pressure gradient is also required to keep the copper

side of a slug in contact with the copper ground plane to maintain capacitive cou-

pling.1 This may be achieved with the addition of a smaller air intake drilled on the

cover layer directly adjacent to the side of slug without copper, as shown in Fig. 7.1.

This, however, will require additional layers on top of the cover layer to direct air

flow. Design improvement will be needed to keep the structural complexity and the

profile of the device low.

Cover

Ground plane on

laminate substrate

Channel

Ring slot

Orthogonal

air intake

Air in

Air in

Air out

Slug

Figure 7.1: Introducing pressure gradient in the orthogonal direction.

7.2.3 A Complete System

The pneumatic control setup used in this project served well for the antenna pro-

totype, which was a proof of concept. A more elegant setup is to replace the ports

1This is not a concern when the antenna lies flat topside up since the slugs naturally rest on theground plane because of gravity.

158

and the tubes with embedded channels. The benefits will be twofold. The practical

antenna profile will be thinner without the 90 elbows and pneumatic tubes, and the

small amount of RF influence on the antenna performance from these components

will be eliminated. Section 3.1 alluded to the use of miniature pumps as the source of

pneumatic pressure, and a preliminary study is provided in Appendix B. The integra-

tion of miniature pumps and valves will be important for realizing a complete system.

Demonstrating a stand-alone device will be a significant milestone in furthering this

technology.

7.2.4 Miniaturization

Another future direction is the miniaturization of the technology and its application

in reconfigurable devices at higher frequencies. The benefits are several-fold:

• Smaller slugs inside a smaller channel (scaled accordingly) experience a differ-

ent combination of forces; stiction becomes more dominant compared to the

gravitational force. A continuous pressure gradient after switching may not be

required to hold the smaller slugs in position against gravity.

• The actuation pressure required to move the slugs may be lower, so the size and

capacity of the miniature pumps could possibly be reduced.

• Given the smaller size and potentially lower actuation pressure required, the

possibility of successfully integrating miniature pumps and valves that are com-

mercially available in the field of microfluidics is greater.

Associated with the potential benefits of miniaturization of the technology are new

obstacles in design and fabrication. Some of the anticipated issues and challenges are:

• Repeated switching will likely lead to a buildup of charge, and stiction in this

159

case will work against the intended operation of the device. Periodic adjustment

to the applied pressure for moving the slug may be necessary depending on the

time it takes for charges to naturally dissipate.

• A new characterization of the contact profile between the slug and the ground

plane, similar to the one performed during the design of the capacitive switch

in Chapter 4, will be necessary. It is not yet known whether the contact profile

will show similar repeatability when the size of the device shrinks.

• A different set of materials will likely be necessary. The standard thicknesses of

commercially available laminate boards become too restrictive during the design

process. The precision in thickness and the consistency in permittivity may also

become inadequate for designing smaller devices. Therefore, custom fabrication

will need to be explored again and the possibility of better fabrication results

is greater for small devices, as mentioned in Section 6.1.1.

• Greater precision in aligning the layers will be required while maintaining the

precision attained in constructing the channel (see Fig. 6.5). Combining the use

of laser machining to fabricate the channel and the use of a flip-chip bonder to

align the various layers presents a promising avenue to achieve the greater fab-

rication precision required for miniaturization. With the advent of 3D printing

technology and its trend of attainable greater precision and wider material se-

lection, it may someday offer a simpler alternative in fabricating pneumatically

controlled devices.

On the opposite note, the current fabrication process in combination with the use

of laminate boards can be readily adapted for antennas operating at lower frequen-

cies, namely, in the L1 or L2 band for applications in GPS (see Section 2.4). Most

160

components and features, such as the slugs and the ring slot, will be larger, which

eases the fabrication precision required.

7.2.5 Other Antenna Designs

The work performed on more complex channel configurations, described in Sec-

tion 3.4.2, can be expanded and integrated into the reconfigurable design of RF

devices. The concepts of channel blocking and multiple slugs in single channels can

lead to some interesting reconfigurable designs. In this thesis, the reconfiguration

of an antenna in the aspect of changing polarization has been demonstrated. An-

other aspect, such as frequency agility, could be explored. This concept has been

applied to a frequency switchable dielectric resonator antenna by another student in

this research group [68], in which an antenna prototype was fabricated and measured

without any pneumatic control test. Besides slot antennas and dielectric resonator

antennas, microstrip antennas and reflectarray elements are also good candidates for

serving as platforms to implement this reconfigurable scheme.

This thesis will be concluded with the introduction of an aperture coupled patch

antenna capable of switching its operating frequency from 3 GHz to 5 GHz, which

is currently being developed by the author. Its structure is shown in Fig. 7.2. The

width (14.57 mm) and the length (28 mm) of the patch provide the resonant condi-

tions for the upper and lower frequencies, respectively. The frequency switching is

achieved by selectively exciting one dimension of the patch. The swinging arm, which

is pneumatically actuated as shown in Fig. 7.3, is coated with copper on the bottom

side. At a given state (operating at 3 GHz or 5 GHz), the arm covers one of the two

slots, such that the energy can be coupled from the microstrip feedline to the patch

161

via the other slot. The multiple layers are aligned using precision dowels through

tight-fitting alignment holes, such that performing the alignment under a microscope

is no longer necessary. The analysis and performance evaluation of this antenna will

be published in the future, and will serve to further demonstrate the versatility of

this scheme of reconfiguration.

Slots

Swinging

arm

Patch

Air ports

Substrate

- ground (top)

- microstrip (bottom)

Channel

Cover 14.57mm

28mm

Figure 7.2: Exploded view of the frequency-switchable aperture-coupled antenna witha pneumatically controlled swinging arm.

162

UF

LF

Microstrip

Patch

Air in

Air out

Air port

UF

Nylon

screw

and nut

Channel

(a) (b)

Figure 7.3: Operation of the frequency-reconfigurable antenna. (a) With the coverlayer removed, this top view shows the position of the swinging arm and the directionof air flow for operation in upper frequency (UF) and lower frequency (LF). (b) Planview of all features on all layers. The microstrip feedline is situated on the bottomside of the substrate, whereas the patch is on the top side of the cover.

163

Bibliography

[1] M. K. Fries, M. Gran, and R. Vahldieck, “A reconfigurable slot antenna with

switchable polarization,” IEEE Microw. Wireless Compon. Lett., vol. 13, no. 11,

pp. 490–492, Nov. 2003.

[2] F. Yang and Y. Rahmat-Samii, “A reconfigurable patch antenna using switchable

slots for circular polarization diversity,” IEEE Microw. Wireless Compon. Lett.,

vol. 12, no. 3, pp. 96–98, Mar. 2002.

[3] Y. J. Sung, T. U. Jang, and Y. S. Kim, “A reconfigurable microstrip antenna for

switchable polarization,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 11,

pp. 534–536, Nov. 2004.

[4] B. Y. Toh, R. Cahill, and V. F. Fusco, “Understanding and measuring circular

polarization,” IEEE Trans. Education, vol. 46, no. 3, pp. 313–318, Aug. 2003.

[5] J. L. Volakis, Antenna Engineering Handbook, 4th ed. McGraw-Hill, 2007.

[6] D. Manandhar and R. Shibasaki, “Possibility analysis of polarization diver-

sity scheme for multipath mitigation in GPS receivers,” in Proc. Int. Symp.

GNSS/GPS, Dec. 2004.

[7] M. A. Kossel, R. Kung, H. Benedickter, and W. Biichtokd, “An active tagging

system using circular-polarization modulation,” IEEE Trans. Microw. Theory

164

Tech., vol. 47, no. 12, pp. 2242–2248, Dec. 1999.

[8] T. A. Milligan, Modern Antenna Design, 2nd ed. Wiley, 2005, ch. 1.

[9] W. L. Stutzman and G. A. Thiele, Antenna Theory and Design, 2nd ed. Wiley,

1998.

[10] Model 3160 Series Pyramidal Horn Antenna: User Manual, ETS-Lindgren, 2011.

[11] Y.-C. Huang and K.-H. Lin, “A study of the thickness sensitivity of the

anisotropic-slab LP to CP by using the inverse circular polarization ratio,” in

Proc. IEEE Antennas and Propagation Society Int. Symp., Jun. 2003.

[12] C. E. Alissandrakis, “Magnetic field diagnostics in the low corona from microwave

circular polarization inversion,” in Proc. Nobeyama Symp., Oct. 1998.

[13] R. Garg, P. Bhartia, I. Bahl, and A. Ittipiboon, Microstrip Antenna Design

Handbook. Artech House, 2001.

[14] C. A. Balanis, Antenna Theory: Analysis and Design, 3rd ed. Wiley-

Interscience, 2005, ch. 14.

[15] K.-F. Tong and J. Huang, “New proximity coupled feeding method for recon-

figurable circularly polarized microstrip ring antennas,” IEEE Trans. Antennas

and Propagat., vol. 56, no. 7, pp. 1860–1866, Jul. 2008.

[16] W.-S. Yoon et al., “Crossed dipole antenna with switchable circular polarisation

sense,” Electron. Lett., vol. 45, no. 14, pp. 717–718, Jul. 2009.

[17] D. H. Schaubert, F. G. Farrar, A. Sindoris, and S. T. Hayes, “Microstrip antennas

with frequency agility and polarization diversity,” IEEE Trans. Antennas and

Propagat., vol. 29, no. 1, pp. 118–123, Jan. 1981.

[18] B. Kim et al., “A novel single-feed circular microstrip antenna with reconfigurable

polarization capability,” IEEE Trans. Antennas and Propagat., vol. 56, no. 3, pp.

165

630–638, Mar. 2008.

[19] R.-H. Chen and J.-S. Row, “Single-fed microstrip patch antenna with switchable

polarization,” IEEE Trans. Antennas and Propagat., vol. 56, no. 4, pp. 922–926,

Apr. 2008.

[20] Y. Dong and T. Itoh, “Substrate integrated composite right-/left-handed leaky-

wave structure for polarization-flexible antenna application,” IEEE Trans. An-

tennas and Propagat., vol. 60, no. 2, pp. 760–771, Feb. 2012.

[21] J.-S. Row and C.-J. Shih, “Polarization-diversity ring slot antenna with frequency

agility,” IEEE Trans. Antennas and Propagat., vol. 60, no. 8, pp. 3953–3957,

Aug. 2012.

[22] G. M. Rebeiz, RF MEMS: Theory, Design, and Technology, 1st ed. Wiley, 2003.

[23] Agilent Technologies, “Understanding RF/microwave solid state switches and

their applications,” App. Note: 5989-7618EN, May 2010.

[24] P. Hindle, “The state of RF and microwave switches,” Microwave Journal,

vol. 53, no. 11, pp. 20–36, Nov. 2010.

[25] C. D. Patel and G. M. Rebeiz, “RF MEMS metal-contact switches with mN-

contact and restoring forces and low process sensitivity,” IEEE Trans. Microw.

Theory and Tech., vol. 59, no. 5, pp. 1230–1237, May 2011.

[26] M. Li and N. Behdad, “A technique for designing liquid-tunable RF lenses,” in

Proc. IEEE Antennas and Propagation Society Int. Symp., Jul. 2010.

[27] C.-H. Chen and D. Peroulis, “Liquid RF MEMS wideband reflective and absorp-

tive switches,” IEEE Trans. Microw. Theory Tech., vol. 55, no. 12, pp. 2919–

2929, Dec. 2007.

[28] Y. Damgaci and B. A. Cetiner, “A frequency reconfigurable antenna based on

166

digital microfluidics,” Lab Chip, no. 13, pp. 2883–2887, May 2013.

[29] C. Karcher, V. Kocourek, and D. Schulze, “Experimental investigations of elec-

tromagnetic instabilities of free surfaces in a liquid metal drop,” in Int. Sci.

Colloquium Modelling for Electromagnetic Processing, Mar. 2003.

[30] R. R. Romanofsky, “Advances in scanning reflectarray antennas based on fer-

roelectric thin-film phase shifters for deep-space communications,” Proc. IEEE,

vol. 95, no. 10, pp. 1968–1975, Oct. 2007.

[31] A. Moessinger, C. Fritzsch, S. Bildik, and R. Jakoby, “Compact tunable Ka-band

phase shifter based on liquid crystals,” in IEEE MTT-S Int. Microwave Symp.

(IMS) Digest, May. 2010.

[32] S. Bulja, D. Mirsheka-Syahkal, R. James, S. E. Day, and F. A. Fernandez, “Mea-

surement of dielectric properties of nematic liquid crystals at millimeter wave-

length,” IEEE Trans. Microw. Theory and Tech., vol. 58, no. 12, pp. 3493–3501,

Dec. 2010.

[33] M. G. Wong, “Piezoelectrically actuated liquid metal switch,” U.S. Patent US

6 756 551, Jun. 29, 2004.

[34] T.-H. Lin, S. Paul, S. Lu, and H. Lu, “A study on the performance and reliabil-

ity of magnetostatic actuated RF MEMS switches,” Microelectronics Reliability,

vol. 49, no. 1, pp. 59–65, Jan. 2009.

[35] R. G. Polcawich et al., “Surface micromachined microelectromechancial ohmic

series switch using thin-film piezoelectric actuators,” IEEE Trans. Microw. The-

ory and Tech., vol. 55, no. 12, pp. 2642–2654, Dec. 2007.

[36] G. H. Huff and S. Goldberger, “A coaxial stub microfluidic impedance trans-

former (COSMIX),” IEEE Microw. Wireless Compon. Lett., vol. 20, no. 3, pp.

167

154–156, Mar. 2010.

[37] S. A. Long and G. H. Huff, “Experiments on a fluidic loading mechanism for

beam-steering reflectarrays,” in Proc. IEEE Int. Conf. Wireless Information

Technology and Systems, Aug. 2010.

[38] L. Yang, L. J. Martin, D. Staiculescu, C. P. Wong, and M. M. Tentzeris, “Con-

formal magnetic composite RFID for wearable RF and bio-monitoring applica-

tions,” IEEE Trans. Microw. Theory and Tech., vol. 56, no. 12, pp. 3223–3230,

Dec. 2008.

[39] G. J. Hayes, Y. Liu, M. D. Dickey, and G. Lazzi, “Light-responsive, heat-

activated, reconfigurable microstrip structures,” in IEEE Int. Symp. Antennas

and Propagation and URSI National Radio, Jul. 2012.

[40] J.-H. So et al., “Reversibly deformable and mechanically tunable fluidic anten-

nas,” Adv. Funct. Mater., vol. 19, no. 22, pp. 3632–3637, Nov. 2009.

[41] G. J. Hayes, J.-H. So, A. Qusba, M. D. Dickey, and G. Lazzi, “Flexible liq-

uid metal alloy (EGaIn) microstrip patch antenna,” IEEE Trans. Antennas and

Propagat., vol. 60, no. 5, pp. 2151–2156, May 2012.

[42] M. Jobs, K. Hjort, A. Rydberg, and Z. Wu, “A tunable spherical cap microfluidic

electrically small antenna,” Small, vol. 9, no. 19, pp. 3230–3234, Apr. 2013.

[43] W. S. T. Rowe and X. Tang, “Pneumatically switched microwave and antenna

structures,” in Proc. European Microwave Integrated Circuits Conference (Eu-

MIC), Oct. 2013, pp. 480–483.

[44] C. M. Shah et al., “Elastomer-based pneumatic switch for radio frequency mi-

crodevices,” J. Microelectromech. S., vol. 21, no. 6, pp. 1410–1417, Dec. 2012.

[45] N. J. Farcich, J. Salonen, and P. M. Asbeck, “Single-length method used to

168

determine the dielectric constant of polydimethylsiloxane,” IEEE Trans. Microw.

Theory and Tech., vol. 56, no. 12, pp. 2963–2971, Dec. 2008.

[46] S. P. Lacour, J. Jones, S. Wagner, T. Li, and Z. Suo, “Stretchable interconnects

for elastic electronic surfaces,” Proc. IEEE, vol. 93, no. 8, pp. 1459–1467, Aug.

2005.

[47] S.-W. Qu, C. H. Chan, and Q. Xue, “Wideband and high-gain composite cavity-

backed crossed triangular bowtie dipoles for circularly polarized radiation,” IEEE

Trans. Antennas and Propagat., vol. 58, no. 10, pp. 3157–3164, Oct. 2010.

[48] A. K. Bhattacharyya and L. Shafai, “A wider band microstrip antenna for cir-

cular polarization,” IEEE Trans. Antennas and Propagat., vol. 36, no. 2, pp.

157–163, Feb. 1988.

[49] R. L. Li, J. Laskar, and M. M. Tentzeris, “Wideband probe-fed circularly po-

larised circular loop antenna,” Electron. Lett., vol. 41, no. 18, pp. 997–999, Sep.

2005.

[50] T. Sudha, T. S. Vedavathy, and N. Bhat, “Wideband single-fed circularly po-

larised patch antenna,” Electron. Lett., vol. 40, no. 11, pp. 648–649, May 2004.

[51] R. L. Li, N. A. Bushyager, J. Laskar, and M. M. Tentzeris, “Determination

of reactance loading for circularly polarized circular loop antennas with a uni-

form traveling-wave current distribution,” IEEE Trans. Antennas and Propagat.,

vol. 53, no. 12, pp. 3920–3929, Dec. 2005.

[52] M. K. Fries and R. Vahldieck, “Uniplanar circularly polarized slot-ring antenna

architectures,” Radio Sci., vol. 38, no. 2, Nov. 2002.

[53] Y.-J. Kim, J.-K. Kim, J.-H. Kim, and H.-M. Lee, “Reconfigurable annular ring

slot antenna with circular polarization diversity,” Asia Pacific Microwave Con-

169

ference (APMC 2007), Dec. 2007.

[54] M. M. Zinieris, R. Sloan, and L. E. Davis, “A broadband microstrip-slotline

transition,” Microwave and Optical Technology Letters, vol. 18, no. 5, pp. 339–

342, Aug. 1998.

[55] National Archives & Rec. Admin., Code of Federal Regulations Title 47, Telecom-

munications, Parts 0-19, Revised Oct. 1, 2009. U.S. Govt. Printing Office, 2010.

[56] B. Wu, M. Okoniewski, and C. Hayden, “Design and fabrication of pneumati-

cally controlled capacitive switches,” in Proc. European Microwave Conference

(EuMC), Oct. 2013, pp. 424–427.

[57] ——, “A pneumatically controlled capacitive switch,” Microwave and Optical

Technology Letters, vol. 56, no. 2, pp. 489–493, Feb. 2014.

[58] P. Colestock and M. Foley, “A generalized TRL algorithm for s-parameter de-

embedding,” Fermi National Accelerator Laboratory Tech. Memo: TM-1781,

Apr. 1993.

[59] B. Wu, M. Okoniewski, and C. Hayden, “Comparing the simulated and measured

results of a pneumatically controlled antenna,” in Proc. Int. Conf. Microwave,

Radar and Wireless Comm. (MIKON), Jun. 2014.

[60] ——, “Reconfigurable ring slot antenna with polarization diversity using pneu-

matic control,” in Proc. IEEE Antennas and Propagation Society Int. Symp.,

Jul. 2014.

[61] ——, “A pneumatically controlled reconfigurable antenna with three states of

polarization,” IEEE Trans. Antennas and Propagat., 2014, accepted.

[62] A. Ghannam, C. Viallon, D. Bourrier, and T. Parra, “Dielectric microwave char-

acterization of the SU-8 thick resin used in an above IC process,” in Proc. Eu-

170

ropean Microwave Conference, Sep. 2009.

[63] R. H. Johnston and J. G. McRory, “An improved small antenna radiation-

efficiency measurement method,” IEEE Antennas and Propagation Magazine,

vol. 40, no. 5, pp. 40–48, Oct. 1998.

[64] R. H. Johnston, “Efficiency simulations on a Vivaldi antenna in a Wheeler Cap,”

in Proc. IEEE Antennas and Propagation Society Int. Symp., Jun. 2009.

[65] W. Kunysz, “Antenna phase center effects and measurements in GNSS ranging

applications,” in Proc. Antenna Tech. Applied Electromag. Int. Symp. & Amer-

ican Electromag. Conf. (ANTEM-AMEREM), Jul. 2010.

[66] B. Wu and M. Okoniewski, “A novel scheme for realizing a microstrip antenna

with switchable circular polarization,” in Proc. Eur. Conf. Antennas and Prop-

agat. (EuCAP), Mar. 2012, pp. 3278–3282.

[67] B. Wu, T. Apperley, M. Okoniewski, and C. Hayden, “Reconfigurable antenna

designs using micro-pneumatic control of dielectric slugs,” in Proc. Int. Conf.

Microwave, Radar and Wireless Comm. (MIKON), May 2012, pp. 69–72.

[68] T. Apperley and M. Okoniewski, “An air gap based frequency switching method

for the dielectric resonator antenna,” IEEE Antennas and Wireless Propagat.

Letters, vol. 13, no. 1, pp. 455–458, 2014.

[69] A. Sutinjo, R. H. Johnston, and M. Okoniewski, “Improved efficiency measure-

ment and analysis methods using an overmoded Wheeler Cap,” in Proc. IEEE

Antennas and Propagation Society Int. Symp., Jul. 2010.

171

Appendix A

Antenna Measurements in

Anechoic Chamber

The antenna being characterized, also known as the antenna under test (AUT), was

measured in the anechoic chamber in the ICT Building at the University of Calgary.

The basic chamber setup is shown in Fig. 6.9 and described in Section 6.2. There

are a number of methods for measuring the gain of the AUT. The more common

types require either a standard-gain antenna with a known gain, a pair of antennas

with unknown gains (in addition to the AUT), or near-field/far-field transformation

[14]. The measurement method selected in this project requires a standard-gain

antenna with a known gain, and the procedure is described below. As long as the

gain of the standard-gain antenna is known over the frequency band of interest, the

measurement method of using a standard-gain antenna is preferred because of its

simplicity. Section A.1 describes how this measurement method makes use of the

Friis transmission equation to determine the gain and obtain the antenna patterns

when the AUT is a linearly polarized antenna. Section A.2 then expands the method

to perform circular polarization measurements.

172

A.1 Linear Polarization Measurements

Generally, the gain of a linearly polarized antenna, unless specified, implies the con-

dition of maximum polarization match — i.e. a hypothetical test antenna used to

measure the antenna is oriented in line with the polarization direction of the antenna

for maximum field reception.

To find the broadside gain of the AUT, the Friis transmission equation is used.

One form of the Friis transmission equation is given as [9]:

Pr

Pt

= εGtGr

(

λo

4πR

)2

(A.1)

where Pr is the power accepted by the receiving antenna, Pt is the power delivered to

the transmitting antenna, Pr/Pt is the power ratio, Gr is the maximum gain of the

receiving antenna, Gt is the maximum gain of the transmitting antenna, λo is the free-

space wavelength, R is the distance between the antennas, and ε is the efficiency factor

(0 < ε < 1), which accounts for the polarization and impedance mismatches between

the antennas. For this project, the transmitting antenna is the standard-gain horn

while the receiving antenna is the AUT, although reversing the roles of the antennas

leads to the same result (theoretically). Also, for both the standard-gain horn and

the AUT, the maximum gain direction is in the broadside direction, therefore, the

antennas are aligned facing each other for the calculation of the broadside gain of the

AUT. The broadside gain of the standard-gain horn from 1.7 GHz to 2.6 GHz can be

obtained from the user manual [10].

To perform a valid gain measurement, the antennas must be in the far-field region

of each other. At a particular frequency, an antenna has an associated Fraunhofer

173

distance — 2D2/λo — beyond which any object is considered to be located in the

antenna’s far field region [9]. D is the maximum dimension of the antenna or, max-

imum dimension of the aperture in the case of an aperture antenna. If one antenna

is placed within the Fraunhofer distance of the other antenna (i.e. the separation

distance is not large enough), the presence of reactive field may be significant enough

to affect the accuracy of the measured results. For this project, the associated Fraun-

hofer distances for both the standard-gain horn and the AUT are smaller than their

separation distance, implying that the chamber size is adequate to satisfy the far-field

requirement for gain measurements.

The vector network analyzer (VNA) is an RF instrument used for measuring the

power transmitted and received by the antennas. The individual powers Pr and Pt

are not measured directly. Instead, the power ratio Pr/Pt is provided by the square of

the magnitude of the forward transmission coefficient |S21|2. It is important to ensure

that the antennas are not separated too far. This is because the free-space path loss(

4πRλo

)2should not be large enough to bring |S21| below the noise floor of the VNA.

Rearranging Eq. A.1, the broadside gain of the AUT can be isolated:

Gr = |S21|2(

1

εGt

)(

4πR

λo

)2

. (A.2)

Since the polarization alignment between the standard-gain horn and the AUT

can be quite accurate, the impedance mismatches of the antennas are expected to

play a greater role in the efficiency factor ε dropping below the maximum value of 1.

Impedance mismatches lead to reflected powers at both antennas, which are captured

by the VNA as reflection coefficients |S11| and |S22| (|S11| for the standard-gain horn

and |S22| for the AUT if the horn is connected to port 1 of the VNA and the AUT to

174

port 2). Therefore, the efficiency factor is estimated as:

ε = (1− |S11|2)(1− |S22|2). (A.3)

Therefore, the broadside gain of the AUT is:

Gr =|S21|2

(1− |S11|2)(1− |S22|2)

(

1

Gt

)(

4πR

λo

)2

. (A.4)

The antenna co-polarization pattern is obtained by rotating the turntable on which

the AUT is positioned for one full cycle (360) and recording the power ratio |S21|2

at each angle (at increments of 1). The pattern is first normalized to the power

ratio at broadside (which is maximum), then the actual gain value at each direction

(−180 < θ ≤ 180) is obtained by multiplying the pattern with the broadside gain

value calculated from Eq. A.4. The broadside gain and the antenna pattern are

usually given in dBi instead of numeric value.

To obtain the cross-polarization pattern, the standard-gain horn needs to be ro-

tated by 90 before running the measurement with the turntable rotation. The power

ratios measured should be significantly lower than those in the co-polarization mea-

surement. The obtained pattern is normalized to the power ratio at broadside from

the co-polarization measurement and then multiplied with the broadside gain value

calculated from Eq. A.4.

A.2 Circular Polarization Measurements

This section describes how the results from two LP measurements are combined with

post-measurement processing to give the gain and patterns of the circularly polarized

175

AUT.

A.2.1 Derivation

The key behind this method is the fact that the two orthogonal CP unit vectors can

each be expressed in terms of two orthogonal linear components [8]:

R =1√2(X− jY) =

1√2(H− jV) (A.5)

L =1√2(X+ jY) =

1√2(H+ jV). (A.6)

Eq. A.5 and Eq. A.6 show that the linear components are not restricted to the

standard unit vectors X and Y in the Cartesian coordinates. As shown in Fig. A.1,

a general pair of orthogonal linear unit vectors, denoted as H and V, can be rotated

by any angle as long as they remain in the plane orthogonal to Z (the direction of

field propagation). Although the angle between H and X (or similarly, V and Y) can

be arbitrary, the order between H and V needs to be consistent with the standard

(positive) orientation of the Cartesian coordinate system (given a particular direction

of Z).

Any electric field vector can be expressed in terms of its linear or circular compo-

nents:

~E = EHH+ EV V

= ERR+ ELL. (A.7)

176

Figure A.1: Generalization of two orthogonal linear components.

The right-hand circular component of any electric field vector can be found using

[8]:

ER = ~E • R∗

= (EHH+ EV V) • 1√2(H− jV)∗

=1√2(EH + jEV ) (A.8)

where R∗represents the complex conjugate of R and EH , EV , ER, EL are all complex

magnitudes. Similarly, the left-hand circular component of an electric field vector in

terms of its linear components is:

EL = ~E • L∗

=1√2(EH − jEV ). (A.9)

From Eq. A.8 and Eq. A.9, it can be seen that the circular components can be

177

obtained by combining or subtracting the linear components phased-shifted by 90.

Recall from Section A.1 that the individual transmitted and received powers are

not measured directly. Therefore, the electric field components should be expressed

in terms of power ratios. Using Eq. A.8, the electric field ratio Er,R/Et — the ratio

of the right-hand circular component of the received electric field to the transmitted

electric field — can be found:

Er,R =1√2(EH + jEV )

Er,R

Et

=1√2

(

EH

Et

+ jEV

Et

)

S21,R =1√2(S21,H + jS21,V ) (A.10)

where S21,H and S21,V are both complex and represent the values directly measured

by the VNA. S21,H is measured when the standard-gain horn is oriented for maximum

polarization match with the EH component of the AUT while S21,V is measured when

the horn is oriented for maximum polarization match with the EV component of the

AUT. S21,L, similarly, is given as:

S21,L =Er,L

Et

=1√2(S21,H − jS21,V ). (A.11)

Given S21,R and S21,L, Eq. A.4 can then be employed to calculate the RHCP and

LHCP gains of any CP antenna.

The axial ratio (AR) can be calculated by [8]:

178

AR =

|Er,R|+|Er,L|

|Er,R|−|Er,L|=

|S21,R|+|S21,L|

|S21,R|−|S21,L|, for RHCP antenna

|Er,L|+|Er,R|

|Er,L|−|Er,R|=

|S21,L|+|S21,R|

|S21,L|−|S21,R|, for LHCP antenna.

(A.12)

A.2.2 Procedure

As described above, a pair of LP orthogonal field components, EH and EV , phase-

shifted apart by 90 can be combined or subtracted to produce the RHCP or LHCP

component, respectively. Since the signal sought is CP, the two LP components do

not need to be aligned with the Cartesian coordinate axes (i.e. X and Y) used for

referencing the plane of the AUT. In fact, given a perfect measurement environment,

the CP patterns obtained are theoretically independent of the angle of rotation of

the LP components as long as the two components remain orthogonal to each other.

In reality, the anechoic chamber at the University of Calgary is far from a perfect

measurement environment. Asymmetry of the chamber, various scatterers (broken

tips of the absorbing cones, the arm of the turntable, etc.), multipath signals, errors

in alignment between the standard gain horn and the AUT, and other factors all play

a role in affecting the quality of the measured results.

The polarization direction of the electric field of the standard gain horn indicated

the direction of the particular LP component of the AUT being measured. To measure

EH and EV (technically, S21,H and S21,V in Eq. A.10 and Eq. A.11 are the quantities

provided by the VNA), two measurements were needed with the horn remounted

orthogonally between measurements.

The procedure for measuring the circularly polarized AUT using LP antenna mea-

surements is given below. To scan the two principal planes (φ = 0 and φ = 90) of

179

the AUT, four LP measurements were required, as shown in Fig. A.2.

1. Mount the AUT on the turntable for scanning the φ = 0 plane of the AUT.

2. Mount the standard gain horn with the shorter edge parallel to the floor of the

chamber. Rotate the horn about its aperture centre to the desired angle of horn

rotation to measure EV of the AUT (Fig. A.2(a)).

3. Align the aperture centre of the horn with the centre of the AUT.

4. Perform antenna measurement with turntable rotation to scan the φ = 0 plane.

5. Remount the horn with the longer edge parallel to the floor of the chamber.

Rotate the horn about its aperture centre to the same angle to measure EH of

the AUT (Fig. A.2(b)).

6. Repeat steps 3 and 4.

7. Remount the AUT for scanning the φ = 90 plane. Keep the horn setup the

same for measuring EH of the AUT (Fig. A.2(d)).

8. Repeat step 3.

9. Perform antenna measurement with turntable rotation to scan the φ = 90

plane.

10. Repeat step 2 (Fig. A.2(c)).

11. Repeat steps 8 and 9.

180

x

y

EV

x

y

x

yEH

EV

EH

x

y

AUT AUT

AUTAUT

Horn Horn

Horn Horn

(a) (b)

(c) (d)

Figure A.2: Linearly polarized antenna measurements required for obtaining CP pat-terns of the AUT in φ = 0 (top diagrams) and φ = 90 (bottom diagrams) planes.A negative angle of horn rotation is shown.

181

Appendix B

Miniature Pumps

The system for creating the pressure gradient inside the channel is beyond the scope

of this thesis. However, some preliminary investigation was performed and miniature

pumps appear to be the most suitable candidate. The technology is well established

in the field of microfluidics, and miniature pumps of various sizes and capacities are

available. Some commercially available pumps are tabulated in Table B.1.

The flow rate indicated is the maximum free flow rate for water. The channels

designed in this thesis are generally less than 3.5 mm wide and 500 µm high. A

slug can be readily moved with the makeshift system of applying pressurized air

from a can of gas duster. The actual flow rate or pressure threshold required to

initiate movement in the slug depends on a number of factors, such as the size of

the slug, the number of slugs inside the channel, and material selection. The power

consumption is for the duration when the pump is on. Ideally the pump will only

be turned on when a change in state is required; therefore the power consumption of

the switching mechanism is minimal. This is unless a holding pressure is required to

keep the slug(s) in position after switching. Given the size of these pumps, especially

the larger ones, implementing an array of antennas in which one pump generates the

pressure gradient in multiple channels to different antenna elements would illustrate

a practical and efficient use of this technology of pneumatic control.

182

Table B.1: Specifications of some commercially available miniature pumps.

Pump Dim

ensions(m

m3)

Weight(g)

Flow

Rate

(mL/min)

PowerConsu

mption

(W)

Bartels mp51 3.5× 14× 14 0.8 5 0.2Bartels mp61 3.8× 15× 30 2 7 0.2TCS M2002 14× 14× 28 11 700 1.89TCS M4002 25× 26× 41 33 2800 6TCS M5002 30× 32× 62 70 4500 121 TCS Micropumps Ltd.www.micropumps.co.uk

2 Bartels Mikrotechnik GmbH.www.bartels-mikrotechnik.de

183

Appendix C

Photolithography Process of

SU8-2100

This appendix documents the detailed procedure for processing the SU8-2100 pho-

toresist in AMIF, which was developed and refined by the author during the course

of this project. The process flow is given in Fig. C.1.

Substrate

Pretreat

Rinse &

DryCoat Soft Bake Expose

Post

Exposure

Bake

Develop

Figure C.1: SU8-2100 process.

C.1 Initial Preparation

• Turn on nitrogen supply and vacuum pump.

• Set up programmable hot plate away from dump rinser but not beyond 3/4

point of wet deck lengthwise, and away from the back of wet deck by 13 cm.

Turn on hotplate with switch at the back.

184

• If necessary use an IPA soaked wipe and/or N2 gun to clean hot plate surface

and aluminum plate.

• Level hot plate on top of 4 adjustable legs with spirit level (legs and level on

top right of labware shelf).

• Place aluminum plate (diameter: 41/8”, thickness: 0.5 mm) on top of hot plate

as heat sink.

• Set programmable hot plate for soft bake (do NOT start) (program #4):

– Room temperature to 30C at 300C/hr

– 30C to 65C at 240C/hr

– Stay at 65C for 7 mins

– 65C to 95C at 240C/hr

– Stay at 95C for 40 mins

– 95C to room temperature at 240C/hr

• Set programmable hot plate for PEB (do NOT start) (program #5):

– Room temperature to 30C at 240C/hr

– 30C to 65C at 200C/hr

– Stay at 65C for 5 min

– 65C to 95C at 200C/hr

– Stay at 95C for 14 mins

– 95C to room temperature at 200C/hr

• Set spinner program (program #14):

– Spread1: 500rpm, 7s, 80rpm/s

– Spread2: 900rpm, 3s, 150rpm/s

– Spin: 1850rpm, 45s, 250rpm/s

185

– Slow: 0rpm, 8s, 200rpm/s

• Follow Mask Aligner document and set mask aligner parameters to:

– Exp. time: 95s

– Exp. type: Prox

– WEC type: Cont

– WEC: 50 um

– Al. gap: 120 um

– Exp. gap: 120 um

• Wear nitrile gloves. Clean mask with IPA and dry with nitrogen. Clean with

cold piranha solution if necessary (refer to piranha preparation document; wear

face shield, chemical apron, and trionic gloves).

• Insert mask in mask aligner.

C.2 Substrate Pretreat

• Piranha-clean wafer if necessary (refer to piranha preparation document).

• Rinse wafer with DI water and nitrogen-dry.

• For depositing resist directly on glass wafer, apply HMDS on wafer in oven.

Allow wafer to cool after. Skip next step.

• For depositing resist on copper-plated wafer, place wafer on hot plate for 5 min-

utes at 150C (see Initial Preparation). Allow wafer to cool to room temperature

on wipe-covered aluminum plate.

186

C.3 Coat

• Clean spinner chamber with acetone-soaked wipes.

• Run a dummy spin program using a dummy wafer.

• Place wafer on vacuum chuck and check centering.

• Select spinner program (see Initial Preparation).

• Open SU8 bottle; use a pipette to transfer some Remover PG onto a wipe and

carefully wipe the lip and top of SU8 bottle.

• Static-pour 5 mL of SU8 (try to avoid bubbles on wafer). CRITICAL STEP!

– Aim for a spot 2.5 cm off centre, have a small amount of resist out of

bottle first before SLOWLY moving across centre of wafer. Do not reverse

pouring motion (otherwise bubbles will appear).

– Twist bottle while reducing resist flow. Always better to have MORE

resist than less. Use wipe to cut off drip near edge of wafer.

– Ensure no film formed at the neck of the bottle prior to storage. Use

pipette to pop film.

• Close spinner lid; wait 2 minutes. The settled resist should cover an area of at

least 4 to 5 cm in diameter. Start program.

• Use a pipette to soak wipes with Remove PG and carefully wipe off streaks

of resist from edge and bottom of wafer. Use dry wipe to wipe away excess

Remove PG. May use cotton buds instead of wipes.

• Transfer wafer to programmable hot plate and place it on top of aluminum

plate. Wait 5min to allow resist to level (do NOT start program yet).

• Clean spinner chamber with Remover PG-soaked wipes thoroughly. Dispose

wipes and eventually, gloves in dedicated SU8 waste bin.

187

• Clean all surfaces (also vacuum wand and tweezers) with acetone-soaked wipes

to remove Remover PG.

C.4 Soft Bake

• Transfer wafer to programmable hot plate.

• Start program in hot plate (see Initial Preparation). “RUN” > choose program

> “ENTER”.

• Rotate wafer 90 every 15 to 20 minutes.

• Once 19C is reached, place wafer on wipe-covered aluminum heat sink. Wait

5 minutes (relaxation time).

C.5 Expose

• If necessary use an IPA soaked wipe to clean the MA6 chuck.

• Align dummy wafer with markings on wafer chuck before loading mask.

• Load mask and centre dummy wafer with respect to mask. Pull out tray and

remove dummy wafer.

• Place wafer on wafer chuck and align with markings. Do NOT adjust wafer’s

position after loading wafer if aligning with mask is not required.

• Start exposure (see Initial Preparation).

• Remove wafer. Reset Exp. type back to Soft, Al. gap back to 20 um, and WEC

back to 10 µm.

• Wipe MA6 chuck with an IPA soaked wipe to remove any SU8 residue.

188

C.6 Post Exposure Bake (PEB)

• Transfer wafer to programmable hot plate. Wait 10 minutes (relaxation time).

• Start program in hot plate (see Initial Preparation). A visible latent image

should appear in 5 to 15s.

• Rotate wafer 90 every 15 to 20 minutes.

• Once 19C is reached, place wafer on wipe-covered aluminum heat sink. Wait

5 minutes (relaxation time).

C.7 Develop

• Place stirrer onto wet deck. Place small hot plate (same height) next to it.

• Place square pyrex dish on stirrer, off-centre, using small hot plate for partial

support.

• Place 2” magnetic pellet in bath close to centre of hot plate (one half of bath)

and 1/8”-thick Teflon plate next to pellet (other half of bath).

• Pour SU8 developer in square pyrex dish.

• Place wafer gently in bath on top of Teflon plate.

• Turn stir setting to 6 and keep stirring on for 20 minutes. Rotate wafer 90

every 4 minutes during stirring.

• Prepare dish for IPA rinsing.

• Turn off stirring and remove wafer GENTLY.

• Rinse wafer GENTLY with IPA. If white film appears, return wafer to SU8

developer bath and turn on stirring for 10s. Repeat multiple times if necessary.

Nitrogen-dry GENTLY.

189

Appendix D

Antenna Radiation Efficiency

Measurements

The improved Wheeler Cap method [63],[64] was chosen for measuring the antenna

radiation efficiency because of its simple procedure, basic materials required, and rea-

sonable estimates. A Wheeler Cap is a metal enclosure (of an arbitrary shape, gen-

erally) that is large enough to contain the antenna under test (AUT) while providing

sufficient clearance to minimize the disturbance to the AUT current distribution. A

picture of the Wheeler Caps used for this project is shown in Fig. D.1.

The original Wheeler Cap method relies on two separate reflection coefficient

(S11) measurements of the AUT — in free space and inside the Wheeler Cap — to

determine the radiation resistance (Rrad) and the loss resistance (Rloss) of the AUT.

The radiation efficiency can be found (assuming a series RLC circuit model for the

radiation mechanism of the antenna) [63]:

η =Rrad

Rrad +Rloss

. (D.1)

When the AUT is in its normal operating environment, i.e. in free space, Rrad

represents the energy transferred to free space while Rloss accounts for the dissipative

losses of the AUT (metal and dielectric losses). The measured S11 provides the an-

190

Small

Wheeler Cap

Large

Wheeler Cap

AUT

150mm

305mm

VNA

Figure D.1: Wheeler Caps. The lid of the large Wheeler Cap was lifted to show theAUT mounted on the base. A coax connector connected the enclosed antenna to theVNA outside. The small Wheeler Cap is shown for size comparison.

tenna resistance that consists of both Rrad and Rloss. When the AUT is placed inside

the Wheeler Cap, no energy (ideally) is radiated into free space. Most of the energy

is reflected back to the source with some amount lost as dissipation. In this case, S11

allows Rloss to be identified independently. Rrad can then be found with the S11 from

the free space measurement. Note that impedance mismatch under normal operating

condition, which quantifies the amount of energy reflected back to the source when

the AUT is in free space, is not considered a part of radiation inefficiency. Antenna

radiation efficiency only concerns with the dissipative losses of the AUT.

191

One major limitation of the original Wheeler Cap method is the requirement of

the AUT being at resonance. In practice, the operating frequency or the frequency

at which the most optimal impedance match is achieved does not necessarily coincide

with the resonant frequency of the AUT. Therefore, the original Wheeler Cap method

does not account for the presence of any antenna reactance if one attempts to measure

the radiation efficiency at a frequency at which the AUT is not at resonance.

The improved Wheeler Cap method, on the other hand, accounts for the reactive

component of the antenna impedance at any frequency of measurement. The mea-

surement setup is almost identical to the original Wheeler Cap method. However,

more reflection coefficient measurements of the AUT inside the Cap are required.

These additional measurements provide S11 data points that describe a circle on the

Smith Chart, which is a manifestation of the phase variation of the reflected wave

within the Cap [63]. This circle is referred to as the reflection circle. The location

of the circle with respect to the free-space S11 at the measured frequency — Sfs11 —

offers the information needed to determine the radiation efficiency. The procedure is

outlined below.

For high-efficiency antennas, the reflection circle is large and quite close to the

edge circle of the Smith Chart. This is because for highly efficient antennas, the

dissipative losses are small. Hence, most of the energy is returned to the source when

these antennas are placed inside the Cap, with |S11| being close to 1. |S11| = 1 is the

edge circle of the Smith Chart.

There are generally two ways to obtain the phase variation needed to sketch the

reflection circle. One method is to change the effective internal size of the Cap by

incorporating a sliding metal plate. The second method, which was chosen for the

192

measurements performed in this project, is to perform a frequency sweep within the

frequency range of interest.

The reflection coefficient (S11) of the AUT, when placed inside the small Wheeler

Cap, is shown in Fig. D.2. It is obvious that the sweep hardly resembled a circle

as one would expect when most of the energy was supposed to be reflected back

to the source with the AUT inside the Cap. This is because the size of the Cap

was large enough to allow for multimode resonances. These multimode resonances

introduced notches of high loss [64], which are observed in the |S11| plots in Fig. D.3.

These notches represent the portions of the S11 curve that approach the centre of the

Smith Chart in Fig. D.2. The loss from these resonances would artificially lower the

measured radiation efficiency.

The ring slot antenna being measured was not an electrically small antenna, so

the Wheeler Cap could not be made any smaller to reduce the multimode resonances.

Analysis methods were demonstrated to numerically reduce the influence of these

resonances on the measured efficiency [69]. A simpler approach is to manually select

a reflection circle that contains most of the frequency-swept curve on the Smith Chart,

as it provides a reasonable representation of the reflection coefficient of the AUT if a

lot of the multimode resonances are absent. It is important to ensure that the selected

circle is not oversized, which will otherwise lead to an overestimation of the radiation

efficiency. The rule of thumb used here is that the circle must at least overlap part

of the S11 curve or be within the curve. This can be observed in the reflection circle

chosen for the S11 curve in Fig. D.4. Here ∆s,max and ∆s,min represent the longest

and the shortest distances between the selected reflection circle and the location of

Sfs11, respectively.

193

With the reflection circle chosen and relevant information obtained from the Smith

Chart, the radiation efficiency is determined using the following equation [63]:

η =2

(∆s,max)−1 + (∆s,min)−1· 1

1− |Sfs11|2

. (D.2)

By using the small Wheeler Cap shown in Fig. D.1, which was just large enough

to contain the AUT, the measured radiation efficiency is 92.1% at 2.4 GHz. This

compares very well with the simulated value of 93.7%.

To further explore the validity of this method of manually choosing the reflection

circle, the large Cap in Fig. D.1 was used. There were significantly more multimode

resonances, as expected. To avoid overcrowding the Smith Chart, the frequency sweep

was reduced to generate an |S11| plot (Fig. D.3(b)) and a curve on the Smith Chart

(Fig. D.5) similar to those observed with the small Cap. A radiation efficiency of

90.3% was obtained from the large Wheeler Cap measurement. As expected, it was

lower than the measured value using the small Cap. This was mainly attributed to the

higher metal loss from the cavity walls associated with greater number of multimode

resonances inside the large Wheeler Cap.

194

0.2

0.5

1.0

2.0

5.0

+j0.2

−j0.2

+j0.5

−j0.5

+j1.0

−j1.0

+j2.0

−j2.0

+j5.0

−j5.0

0.0 ∞

X: −0.1551Y: 0.2144

S11fs

Figure D.2: S11 (dotted black curve) of the AUT inside the small Wheeler Cap, shownon the Smith Chart, with a frequency sweep from 2.1 to 2.7 GHz. Sfs

11 represents thereflection coefficient (a complex value) of the AUT in free space at 2.4 GHz.

195

2.1 2.2 2.3 2.4 2.5 2.6 2.7−20

−15

−10

−5

0(a) Small Wheeler Cap Measurement

Freq (GHz)

|S11

| (dB

)

2.35 2.4 2.45−20

−15

−10

−5

0(b) Large Wheeler Cap Measurement

Freq (GHz)

|S11

| (dB

)

Figure D.3: |S11| of the AUT inside the Wheeler Caps.

196

0.2

0.5

1.0

2.0

5.0

+j0.2

−j0.2

+j0.5

−j0.5

+j1.0

−j1.0

+j2.0

−j2.0

+j5.0

−j5.0

0.0 ∞

X: −0.1551Y: 0.2144

X: −0.5065Y: 0.7979

X: 0.4424Y: −0.7708

S11fs

∆s,min

∆s,max

Figure D.4: Determining the reflection circle (dashed grey circle) from the measuredS11 (dotted black curve) on the Smith Chart, with a frequency sweep from 2.1 to2.7 GHz and the AUT inside the small Wheeler Cap. Each data point (X,Y) representsthe real and imaginary values of the reflection coefficient at that specific location onthe Smith Chart.

197

0.2

0.5

1.0

2.0

5.0

+j0.2

−j0.2

+j0.5

−j0.5

+j1.0

−j1.0

+j2.0

−j2.0

+j5.0

−j5.0

0.0 ∞

X: −0.1551Y: 0.2144

X: 0.5902Y: −0.601

X: −0.6101Y: 0.7161

∆s,min

∆s,max

S11fs

Figure D.5: Determining the reflection circle (dashed grey circle) from the measuredS11 (dotted black curve) on the Smith Chart, with a frequency sweep from 2.35 to2.45 GHz and the AUT inside the large Wheeler Cap.