DesignCon 2014

Quantitative EMI Analysis of

Electrical connectors Using

Simulation Models

Michael Rowlands, Molex

michael.rowlands@molex.com

Alpesh Bhobe, Cisco Systems

abhobe@cisco.com

Patrick Casher, Molex

patrick.casher@molex.com

Xiao Li, Cisco Systems

xiaoli3@cisco.com

Abstract

Radiated energy is a problem that grows in complexity as data rates increase. In

addition, EMI problems often show up late in the system verification process, near to

system product shipping deadlines. The solutions to these EMI problems are very costly

and difficult to implement. Hence it is always good to capture the potential EMI

problems via simulation and analysis during the product design phase, rather than during

EMC regulatory measurements towards the end of the product development. In addition,

simulation techniques for EMI are often complicated and time consuming, and also not

suited for broadband analysis.

This paper describes a method of using 3D field solver tools to analyze radiated

energy across a range of frequencies. A 3D field solver model is run and the s-

parameters are generated across a range of frequencies. The initial solve point is used to

generate quantitative results for radiated energy. Then just the initial solve is re-run at a

various frequencies, chosen based on interesting points from the s-parameter results. The

initial solve completes quickly so that multiple points can be used to generate radiated

energy results in a range of frequencies. The method is then used to analyze the EMI

performance from a few connector structures and compared to lab measurements.

Various features are then compared concerning their effect on EMI.

Author(s) Biography

Michael Rowlands is an Electrical Engineer in the Signal Integrity and Connector

Design group at Molex. He specializes in signal integrity at multi-gigahertz frequencies.

He received a Bachelor's and Master's degree in Electrical Engineering from MIT in

1998. Upon graduation, he worked four years as a signal integrity engineer at Teradyne

in Boston. He designed cable assemblies, circuit boards and interconnects for test

equipment up to 6 GHz. In 2002, he worked at a startup company in Illinois. The

company designed dispersion compensation microchips at 12.5 Gbps for fiber-optic

communications. He designed circuit boards to demonstrate and verify 12.5Gbps

performance and made algorithm improvements based on system modeling. He has

authored or co-authored and presented technical papers at ECTC, DesignCon, IMAPS,

IPC-APEX and PCB East. In 2005, as part of the Research and Development at Endicott

Interconnect Technologies years he designed and analyzed circuit boards, chip packages

and custom computing systems. Since 2009, he's worked at Molex designing next-

generation 25-40Gbps I/O and board-to-board connectors.

Alpesh U. Bhobe received his Ph.D. in Electrical Engineering from the University of

Colorado at Boulder, Colorado in 2003. He was a Post-Doc at NIST in Boulder, Colorado

from 2003-2005. While at the University of Colorado and at NIST his research interest

included the development of FDTD and FEM code for EM and Microwave applications.

Currently, he is working as a Manager in EMC Design Cisco Systems, San Jose, CA.

Patrick Casher received his BSEE and MSEE in Electrical Engineering from Illinois

Institute of Technology in 1991 and 2006 respectively. The concentration of study for

his graduate work was in the areas of Numerical Methods, Electromagnetics and High-

Speed Circuit design. He joined Molex Incorporated in 2001 and he is currently the

Signal Integrity Engineering Manager. His team is responsible for the development of

high-speed IO connectors and cages from a signal integrity and shielding effectiveness

standpoint. He holds a least a dozen patents in the fields of high-speed connector and

shield design.

Xiao Li is a master student of EMC lab of Missouri University of Science and

Technology (MS&T) where his study focuses on the analysis and measurement for EMC

and SI problems. He received his first master degree from Beijing Jiaotong University,

China. Prior to moving to the US in 2005, he was working as EMC research at the EMC

lab of the National Institute of Metrology in Beijing, China. Currently he is working at

CISCO as an intern in EMC team focusing on EMC simulation, troubleshooting and

measurement tool development.

Section I

Introduction

Electromagnetic Interference/ Electromagnetic Compatibility (EMI/EMC) is a

concern for any electronic system, especially at multi-Gbps data rates. EMI/EMC

problems are often found late in the build phase of a project which makes problems

expensive and difficult to fix. The goal of this study is to establish a manageable

simulation method for analyzing broadband EMI effects and to use it for quick EMI

analysis , with various sources and structures.

EMI from backplane connectors have been studied via simulations and

measurements before. However, a comprehensive method to analyze radiation up to 40

GHz does not exist it literature. In [1] the study focuses on radiation mechanism and

radiation affecting factors for a generic backplane connector up to 10 GHz. The paper

also discusses techniques to reduce radiated emissions. A common-mode current and a

near-field probe measurement technique were introduced in [2]. The method was applied

to evaluate the EM1 performance of an open-pin-field module-on-backplane connector

up to 3 GHz.

The outline of this paper is as follows. In Section II we introduce the simulation

methodology to analyze the EMI performance of backplane connections from 1 – 18

GHz. Section III explains in details the measurement setup used to analyze the Total

Radiated Power from connectors. The section also discusses the methodology used to

correlate simulation results to measurements. In section IV we present EMI radiation

analysis from different connectors types based on the proposed simulation method.

Section II

Simulation Method The electromagnetic simulation method to analyze radiation from PCB connectors

must be sufficiently accurate to replicate real life EMI issues. Additionally, in order to be

practical, the EMI simulation method must generate results in the order of hours or a few

days. The simulation method introduced in this paper begins by using a 3D full wave

solver to calculate EM fields & Total Radiated Power in the frequency range of 1 – 18

GHz. Then, using equation 1, the solver frequency sweep generates s-parameters that are

used to calculate all energy that doesn’t come out of simulation ports. The field solver

antenna parameters are then used to generate a ratio of dissipated- to-radiated energy.

This ratio is used along with the s-parameter data to calculate radiated energy and

dissipated energy at the field-solve frequency. The s-parameter data over frequency

shows resonance peaks and anti-peaks. These frequency points are noted. The initial

field solve is re-run at various frequencies, without sweeping frequencies. Specific

frequencies are chosen to include the peak resonance and anti-resonances noted in the

broadband s-parameter energy. Additional frequencies to solve fields are spaced out

through the relevant band, by choosing a near DC frequency, and then choosing regular

intervals. At each discrete frequency, the fields are solved and the antenna parameters

calculated. The final result is a set of curves vs. frequency for energy lost in the

structure, energy dissipated and energy radiated. The final results curves give a

quantitative look at energy radiated vs. energy dissipated. In a structure, using a Network

Analyzer, the energy going in and coming out can be readily measured by s-parameters.

The energy that doesn’t transmit, where does it go? Knowing the radiated vs. dissipated

energy ratio tells exactly where the energy goes.

In order to establish this broadband EMI simulation method, the simulation results must

be correlated to measured data. In this study, measurements are made using a

Reverberation chamber, a single-ended co-axial input, a horn probe and a loop probe as

shown in Fig. 1. In order to correlate simulation and measurement results, efforts are

made to change the measurement geometry to be more like the simulation, and vice

versa. The structures analyzed include typical connector configurations and typical circuit

board configurations. When energy goes into a structure, and doesn’t come out, it often

assumed the radiated energy is the lost energy, minus the typical differential insertion

loss of the structure. The sources analyzed include a pure differential signal, pure

common mode, skew sources which include common and differential energy, plus a

ground path with energy on the entire path. These experiments will compare radiated

energy in various structures, each with a range of sources. Conclusions are drawn from

the data, to identify what structures and sources cause the biggest radiated energy. Also,

the results are compared to the s-parameter data to see what, if any, correlation there

exists between s-parameters and radiated energy. The s-parameter to radiated energy

relation is analyzed to see if s-parameters are sufficient for understanding EMI, or if

detailed information on the 3D structure is needed for accurate EMI analysis.

The method to simulate radiated power, using the HFSS 3D field-solver, begins with the

mesh settings. The mesh on the rectangular radiation boundary box is set to a length

based limit of wavelength/12 for 40GHz in air. This forces sufficient mesh on the

boundary box to get accurate radiated power results. The first solution is done at a high

frequency and then the frequency sweep is completed to get broadband s-parameter

results. The s-parameter results, especially, return and insertion loss, are examined to

find any resonances in the structure. These resonance frequencies are noted so that

subsequent radiated power solution will include may frequency points near the

resonances.

After the simulation is complete, the antenna parameters are calculated and the accepted

and radiated power is recorded at the mesh solution frequency. A convenient way to

generate radiated power results, for multiple frequencies, is to use dependent solves.

After the first solution is complete, dependent solution points are generated at every

frequency of interest. The dependent solution points re-use the mesh from the first

solution, and are set to run one more adaptive pass. Analyzing all the dependent solves

overnight, the mesh solution antenna parameters are generated and plot to make a curve

of radiated power vs. frequency.

A check is done on the simulation results are generating using a 3D electro-magnetic

field solver. Here are the settings used in HFSS to produce accurate results:

HFSS 3D field solver settings to simulate radiated power:

- In order to generate sufficient accuracy, set the maximum mesh length on the

radiation boundary, to 1/12 of the wavelength at the mesh solution frequency.

- Run a full solve at a multi-GHz solution frequency, such as 40GHz, and complete

the simulation sweep. Look at differential and common mode loss and note any

resonance frequencies.

- Edit the waveport sources. A single-ended 1 mW source, (0.321623V across 50

ohms), with all other ports terminated to 50 ohms is convenient because it

matches the lab setup. In HFSS, “Incident voltage” and “Include Post Processing

Effects” must be turned on to match exactly with the lab 1 mW source.

- At the solution frequency, generate the accepted power and radiated power

antenna parameters using a far-field, infinite sphere setting.

- One way to get a curve of radiated power vs. frequency after solving the model

for the first time, is to disable the frequency sweep and add dependent solves at

each frequency of interest, which point to the initial solved mesh and run one

more adaptive mesh. Solving all these setups takes many hours and is typically

run overnight.

In order to verify the accuracy of the antenna parameters and understand the radiated

power calculation, a field calculator was used on the solved 3D model. The accepted

power from the far-field antenna parameters represents all the power that does not come

out of the ports of the model. Therefore, the accepted power equals the input source

power times (1 minus energy from all ports):

����������� = ���������� × [| ��|� +| ��|

� +⋯+| ��|�] 1

The radiated power equals all the power passing through the radiation boundary surface.

The field calculator is used to generate radiated power by integrating the pointing vector

on the surface of the radiation boundary.

���� =∮� ×�∗ ∙ � 2

The accepted power minus the radiated power equals all the power dissipated in the

model and not coming out of the ports of the model. This quantity is generated in the

field calculator by integrating the conductor loss on all the conductive surfaces of the

model, then adding all the dielectric loss in all the non-conductive objects in the model.

In [4-5] the antenna parameters are compared to the calculated results. These results

show good agreement between the antenna parameters and the results generated manually

with the field calculator. Henceforth, the antenna parameters will be used to quickly

generate radiated power and dissipated power in a model.

Table 1 Antenna Parameters Compared to Field Calculator Results

Frequency 30 GHz

Source Voltage 0.316228 V

Zo 50 ohms

Source power 0.001 Watts

Antenna incident power 0.001 Watts

1 – {power out of all ports} 0.0002922 Watts

Antenna accepted pwr 0.000298 Watts

Field calculator radiated rower 0.000223 Watts

Antenna radiated power 0.000221 Watts

Field calculator conductive loss 5.34E-05 Watts

Field calculator dielectric loss 2.65E-05 Watts

Field calculator total dissipated power 7.99E-05 Watts

Antenna dissipated power (acc – rad) 7.70E-05 Watts

A ground structure driven with a small source is used to simulate a system with some

noise of the ground path. The source impedance of that ground structure is set to a low

number, such as 0.1ohms and the source voltage is set to 14mV to create a source power

of 1mW/ 0dBm.

Section III

III. 1: Measurement Procedure In order to correlate simulation and measurement, both the simulation and the lab

measurement must be setup carefully, with the environments matching as closely as

possible. This is done with a simulation-measurement-simulation method. The

simulation is done first, noting the various settings of the simulation and getting familiar

with the performance results. Then the measurement is done, with an effort to replicate

the simulation settings. Inevitably, the measurement equipment cannot do exactly the

same things as the simulation. The exact measure settings are noted and then brought

back into the simulation. The final simulation is done to reproduce the measurement

results as closely as possible.

The measurements of radiated power are done using a reverberation chamber as shown in

Figure 1. The equipment used is ETS-Lundgren SmarTIMM. The 1-18GHz horn is used

as the receiver is the ~2 meters x 1 meter x 1 meter chamber. The network analyzer

makes the measurements, which are done by driving the device with a signal, and

measuring the radiated energy with the receiver antenna. Cables connect the network

analyzer to the device, through an amplifier to boost the signal to a level that’s easier to

measure. There’s a metal rod with fines in the chamber which stirs and reflects the

electromagnetic waves into numerous angles. The data recorded is the average and

maximum energy measured over all the stirring. The horn antenna receives energy

radiated from the device and cables connect that signal back to the network analyzer

which measures the received energy.

Figure 1: Reverberation Chamber Measurement Setup

In order to correlate measured data to simulated results, the different components

of the measurement path like the cables, the chamber, etc. must be calibrated. The cable

loss is measured by simply bypassing the amplifier and the chamber, thus connecting the

cables in a loop. The amplifier gain is generated by measuring the cable path with the

amplifier and subtracting the cable loss. The chamber calibration is show in Figure 2.

The horn-to-horn measurement is done in the chamber, and the cable loss is subtracted to

get the chamber calibration factor. The last piece of the puzzle is the device loss. For

instance, in a connector test board, there are a few centimeters of trace before the

connector, which adds a little bit of loss. The cable, amplifier, chamber calibration and

test board loss are added together to get the total calibration factor.

Figure 2: Reverberation Chamber Calibration Setup

Calibrating the test setup is critical to correlating measurement results to

simulation results. For the cable and amplifier, this is straightforward by connecting

cables in the right path and yields predictable results. The chamber calibration requires

careful setup of the antenna horns and calculations with the cable loss.

(a) (b)

Figure 3 (a) and (b) show the reverberation chamber calibration setup.

(a) (b)

Figure 3 (a) Cable feed-through into EMI reverberation chamber & (b) cable feed

through to horn antenna

The total calibration factor is added to the raw measurement, in order to correlate

simulation results to measurements. The simulated result is equal to a known source,

right at the device, with all the far-field radiation integrated at the radiation boundary of

the simulation. Thus the measurement + total calibration factor = simulated radiation

power.

III. 2: Results of Measurement to Correlation Effort

The calibration results are shown in Figure 4 includes cable loss, amplifier gain,

chamber calibration and total calibration curve. The gain curve is lower at low

frequencies because the amplifier is at its max output for most of the frequency range.

The amplifier nominal gain is 23dB, and has a maximum output power of about 12dBm.

Note that the gain curve and the cable loss mostly cancel each other, so the chamber

calibration is similar to the total calibration.

Figure 4: Measurement Calibration Data

During the measurements, there is access to two reverberation chambers. They

are identical in geometry and similar in antenna horns. The chamber calibration factor is

similar between the two reverberation chambers. There are some noticeable differences

below 6GHz. This result substantiates the concept that the chamber effects are linear,

passive, and repeatable across multiple chambers of the same geometry.

Combining the total calibration factor with a measurement, allows the direct

comparison of simulation results to measurement results. The correlation, between

simulation and measurement, is done with a surface mount connector, on a test board,

with edgecard mating. The measurement equipment is set to a 1 mW, single-ended

source. The simulation is also set to a 1 mW, single-ended source. Figure 6 shows the

simulation and measurement of radiated power, and shows good correlation overall. The

trend across the frequency range correlates well. The magnitude matches well at

frequencies greater than 10GHz. The measurement shows peaks much more sharply than

the simulation.

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Cable and Board Loss Amplifier Gain

Chamber Calibration Factor Total Calibration Factor

Figure 5: Dual Chamber Calibration Factors

Figure 6: Total Radiated Power, Simulation vs. Measurement

Since most radiated energy problems are at 10GHz and higher, the measurement-to-

simulation correlation is sufficient to compare connectors and geometries in simulation

and see what features tend to decrease radiated power.

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Section IV

Results of Simulation Experiments

Simulations are completed for various configurations. Multiple connectors are

analyzed, and each connector is analyzed with a set of multiple sources. The connectors

analyzed are a right angle surface mount connector, a right-angle press-fit connector; a

mezzanine connector and a vertical surface mount connector. This is summarized in

Table 2 and Figure 7.

Table 2 Connector types simulated to generate total radiated power

Connector Type Ground structure

Right-angle surface mount Ground pins same as signal pins

Right-angle press-fit Ground pins wider than signal pins

Mezzanine press-fit Large ground pins, like a shield

Vertical surface mount Large ground pins, like a shield

Figure 7: Connector types simulated to generate total radiated power

The energy sources are in a differential configuration, common mode

configuration, single-ended, ground noise and “super common mode”. The ground noise

configuration is a voltage source driving the entire ground structure of the model. The

super common mode configuration is a voltage source that drives all metal in the model,

including the ground structure and the signals. In all the source configurations, the 3D

field solver model uses open waveports, which means that all metal, including ground is

driven as a terminal and solved as a signal. In the edit source setup, the grounds are set to

0V for the single-ended, differential and common mode configurations. For the ground

noise configuration, the signals are set to 0V and the ground structure is driven with a

voltage. In the super common mode configuration, both signal pins and the ground

structure are driven in phase with 1/3 of 1 mW each. The voltages are set so that the

source is 1 mW. The ground structure termination is set to 0.1 ohms to ground to

represent a low impedance path to ground. The signal pins are terminated with 50 ohms

to ground. The 1 mW normalized source configurations are noted in Table 3.

Table 3: Source configurations normalized to 1 mW

Source

Configuration

Signal 1 (V) Signal 1

phase

Signal 2 (V) Signal 2

phase

Ground (V)

Single-ended 0.316228 0 0 0 0

Differential 0.223607 0 0.223607 180 0

Common Mode 0.223607 0 0.223607 0 0

Ground noise 0 0 0 0 0.014142

Super Common

mode

0.181659 0 0.181659 0 0.008124

The purpose, of the multiple-source configurations, is to understand radiated

power as a function of sources in a system. A typical concern for EMI is noise on a

ground structure. Another typical EMI concern is common mode energy on a differential

path. Figure 8 shows total radiated power (TRP) results for the prototype right-angle

surface mount connector 7. The results in Figure 8 clearly show that given a constant

source power sinewave, common mode energy radiates more at frequencies below 8

GHz. Above 8 GHz, differential energy and common mode energy are similar. The

radiated power from a differential signal has a resonance peak at 10 GHz, while the

common mode radiated power is lower at that frequency. Results clearly show that

ground noise doesn’t radiate very efficiently per mW, compared to the signal structure.

With a 14 mV amplitude sinewave, on the ground structure at 10 GHz, the structure

radiates power at almost 2 orders of magnitude lower than a 0.316 V sinewave driven on

a signal pin.

In order to compare the radiated power of these various structures at a constant

voltage, a source of 0.1V amplitude is used in all configurations. The results are plotted

in Figure 9. Given the same source power, the differential mode radiated power is the

highest of all the configurations at 10 GHz. The common mode radiated power is highest

from 0 to 9 GHz. Above 11 GHz, the common mode and differential mode radiated

power is about the same.

Figure 8: Total Radiated Power vs. Frequency, All sources normalized to 1 mW,

prototype right-angle surface mount connector

Figure 9: Total Radiated Power vs. Frequency, All sources normalized to 0.1V,

prototype right-angle surface mount connector

When the configurations are driven with the same voltage, the relative merit of

the configurations changes significantly compared to constant source power. By voltage,

the ground structure radiates the most power. Due to low impedance, the ground

structure radiates much less than differential and common mode. The super common

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Frequency (GHz)Single-ended 1mW Differential 1mWCommon Mode 1mW Ground noise 1mWSuper common mode 1mW

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Single-ended 0.1V Differential 0.1V Common Mode 0.1V

Ground noise 0.1V Super comm 0.1V

mode configuration is just a super-position of the ground noise and common mode

configurations, so ground noise and common mode will be the focus of subsequent

analysis.

Let’s examine the radiated power results compared to s-parameter values. S-

parameter files for electronic structures are generally more available than radiated power

data. In Figure 10, it can be seen that taking 1 minus all energy coming out of the ports,

yields a curve that has the same shape as the radiated power of a single-ended source.

This makes sense because the s-parameter file contains single port data. Also, it makes

sense because all energy into the structure must go somewhere. Most of it goes out of the

ports, while the remainder (1 – all ports) is either radiated or dissipated. In the antenna

parameters, this remainder is the accepted power. The dissipated power is the dielectric

and conductor loss. The accepted power reduced by the dissipated power generates the

radiated power curve. Therefore, the radiated power from a single-ended source is 1-

{energy from all ports} shifted lower by the loss in the structure. Both the differential

source and radiation results and the common mode source and radiation results roughly

correlate by shape. The loss dips correlate to radiated energy peaks, but magnitudes do

not correlate well. S-parameters are good to estimate the trends of radiated power vs.

frequency, but are not good enough to estimate absolute magnitudes or whether a

component will pass or fail EMI tests.

Figure 10: Prototype Right-angle SMT Connector: Radiated Power vs. S-parameter

In a typical electronic system, the sources will not be 1 mW for all configurations,

nor will they be 0.1V on each metal pin. In addition, the sources will not be a single

frequency sinewave. For the purposes of a system analysis, the sinewaves are weighted

by a typical frequency spectrum for binary data. Differential, common mode and ground

noise uses a 25Gbps binary waveform spectrum. The single-ended source uses a binary

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Frequency (GHz)SDD21 Differential 1mWCommon Mode 1mW 1 - allportsSCC21 Single-ended 1mW, rt angle SMTSingle-ended dissipation

waveform spectrum at a data rate of 8Gbps. Differential signals are driven with

waveforms typically from 0.3V to 1V. The common mode signals are designed to be

small, for instance the IEEE 802.3ba specifies AC common mode output to 30mV max.

Ground noise and super common mode are designed to be very small in a system,

hopefully 10mV or less. Single-ended pins may be driven to 0.3V to 1.0V, similar to the

differential signals. However, they are typically lower data rates and slower rise time

than differential signals. Assume 0.6V amplitude, with 6dB loss on the differential path,

the result is 0.3V on the differential signal. Assume a 0.5V sinewave on the single-ended

path with a frequency range of 1 to 8GHz. Assume 30 mV common mode, plus -15dB

mode conversion of the 0.6V differential signal, which adds another 100mV for a

maximum common mode total of 0.13 V. While lab measurements occasionally show

large ground noise spikes of 100 mV peak amplitude or more, the high frequency

sinewave component is typically much smaller. Let’s use 25mV as the ground noise

sinewave amplitude. Figure 11 adjusts the source voltages to these estimates of typical

system values.

Figure 11: Analysis of radiated power from a prototype right-angle SMT connector in a

system, weighted by the frequency spectrum of binary data waveforms

For this estimate of signals in a system, it’s clear that large amplitude single-ended

sources generate the most radiated power at lower frequencies, up to about 4GHz,

assuming a fast, 8Gbps single-ended source. Differential signals dominate above 8GHz.

In no frequency range, does the 25mV ground noise generate the most radiated power.

However, it must be noted that all plots shown thus far are in an unshielded environment.

Most systems have a grounded shield or cage, around some components, to reduce

radiated power. The next experiment is to simulate a shield around a connector and

examine what happens to radiated power when voltage is driven onto that shield. The

signal pins inside the cage certainly radiate more per mW than the ground structure, but

they are shielded by the cage. In Figure 12, a shielded differential signal is analyzed and

compared with a noise source on the shield structure.

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Single-ended 0.5V, 8Gbps Differential 0.3V, 25Gbps

Common Mode 0.13V, 25Gbps Ground noise 0.025V, 25Gbps

Figure 12: SHIELDED SYSTEM: Analysis of radiated power in a system with shielding

around the connector, weighted by the frequency spectrum of binary data

It’s clear that for an unshielded case, a 25mV source on the ground structure is

much less significant than the driven signals, be they common mode, differential or

single-ended. However, with a shielded structure, all the driven signals are sufficiently

shielded that they have radiated energy that’s insignificant compared to a noisy shield.

The right-angle surface mount connector analyzed has a noticeable differential

resonance around 10GHz. This connector is a prototype and the final connector design

has significant resonance reduction, compared to the prototype. The final right-angle

SMT design with resonance improvement is analyzed and compared to the prototype

design (Figure 13). Notice that there’s a big improvement at 10GHz, since the resonance

is reduced.

Figure 14 compares the TRP for the 4 different connectors when each is driven

with a purely differential signal. At low frequency, there’s up to 20dB difference in the

radiation from one connector to another. At frequencies of 10GHz and above, the

differences are less pronounced.

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Single-ended 0.5V, 8Gbps Differential 0.3V, 25Gbps

Common Mode 0.13V, 25Gbps Ground noise 0.025V, 25Gbps

Figure 13: Total Radiated Power vs. frequency, estimated system sources, for prototype

vs. final design of right-angle SMT connector

Figure 14: Total Radiated power vs. Frequency among 4 connector types, with

differential 25Gbps sources

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Differential 0.3V, 25Gbps final Common Mode 0.13V, 25Gbps final

Differential 0.3V, 25Gbps proto Common Mode 0.13V, 25Gbps proto

final design SDD21 prototype SDD21

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Mezz, Diff 0.3V, 25Gbps Right-angle press-fit, Diff, 0.3V, 25Gbps

Vert SMT, Diff 0.3V, 25Gbps right-angle SMT, Diff 0.3V, 25Gbps

Figure 15 shows the TRP results for different connectors when driven with a

common mode signal. At low frequency, there’s up to 15dB difference in the radiation

from one connector to another. At frequencies of 10GHz and above, the differences are

less pronounced.

Figure 15: Total Radiated power vs. Frequency among 4 connector types, with common

mode 25Gbps sources

Using the selection of connector results, these results are compared to EMI spec

to estimate what radiated energy performance is good. With the datasets of total radiated

power for multiple connectors, in various configurations, the question remains; what will

pass EMI tests? This is a question that’s best answered by measurement to the spec, or

extensive analysis. Some estimates can be made to get a rough idea of EMI. There is an

FCC spec for EMI for enterprise (business/commercial) environments, which has a limit

of 54dBuV/m for electrical field, at 3 meters from a component, above 1GHz. Using

total radiated power, over the surface of a sphere with 3 meter radius, the average electric

field is calculated. In order to meet the 54dBuV/m spec, the estimate is that the total

radiated power (TRP) must be less than or equal to -24dBm. In Figure 16, the estimated

TRP target is plotted compared to the four analyzed connectors, in the unshielded case. It

is clear that at 25Gbps no unshielded connector meets the target.

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Mezz, Comm 0.13V, 25Gbps Right-angle pfit, Comm, 0.13V, 25Gbps

Vert SMT, Comm 0.13V, 25Gbps Right-angle SMT, Comm 0.13V, 25Gbps

Figure 16: Total Radiated Power, Estimated target to meet 54dBuV/m FCC spec vs.

Connector type @ 25Gbps

In the case with 12.5Gbps, the radiated energy (Fig. 17) is much closer to meeting

the -24dBm target. In fact, the Mezzanine connector meets the TRP target with both

common mode and differential mode energy sources.

Multiple drivers are also investigated. The cases analyzed include differential and

common mode sources, driven on all differential pairs in the model. The right-angle

press-fit connector is analyzed in this configuration, with all 8 pairs (16 signal pins)

driven with a source. The results are shown in Figure 18.

These results show that multi-pair sources add up to higher radiated power than a

single pair, as expected. More power in equals more power out. However, it seems that

the differential radiated power adds up more linearly than the common mode power. One

hypothesis is that the common mode sources generate some patterns that cancel out

radiated energy. Another hypothesis is that the pairs in the right angle pairs have various

lengths, each of which excites a different resonance. These differential resonances may

add up to create smooth, wide peaks in the radiated energy. The multi-pair source

configurations warrant further study.

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Mezz, Diff 0.3V, 25Gbps Right-angle press-fit, Diff, 0.3V, 25Gbps

Vert SMT, Diff 0.3V, 25Gbps Right-angle SMT, Diff 0.3V, 25Gbps

Mezz, Comm 0.13V, 25Gbps Right-angle pfit, Comm, 0.13V, 25Gbps

Vert SMT, Comm 0.13V, 25Gbps Right-angle SMT, Comm 0.13V, 25Gbps

TRP target for 54dBuV/m average E-field

Figure 17: Total Radiated Power, Estimated target to meet 54dBuV/m FCC spec vs.

Connector type, 12.5Gbps

Figure 18: Total Radiated Power vs. Frequency for Multi-pair sources vs. single-pair source

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Mezz, Diff 0.3V, 12.5Gbps Right-angle press-fit, Diff, 0.3V, 12.5Gbps

Vert SMT, Diff 0.3V, 12.5Gbps Right-angle SMT, Diff 0.3V, 12.5Gbps

Mezz, Comm 0.13V, 12.5Gbps Right-angle pfit, Comm, 0.13V, 12.5Gbps

Vert SMT, Comm 0.13V, 12.5Gbps Right-angle SMT, Comm 0.13V, 25Gbps

TRP target for 54dBuV/m average E-field

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Differential 8 pairs, 0.3V 25Gbps Differential 0.3V, 25Gbps

Common Mode 0.13V, 25Gbps Common Mode 8 pairs, 0.3V 25Gbps

Single-ended connector paths are analyzed surrounded by ground contacts on both

sides, compared to ground contacts on one side. (Figure 49) The data shows clearly that

the signal with grounds on both sides radiates less energy than with ground on one side.

This suggests that connects that more ground pins, even small structures, help confine the

radiated fields.

Figure 49: Snapshot and plot of radiated energy vs. ground pin configuration

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Single-ended 0.5V, 8Gbps grounds on both sides

Single-ended 0.5V, 8Gbps grounds on one side

SECTION V

Conclusion

Simulations are shown to produce accurate radiated energy results. Simulation

results correlate well with measurements if the both the simulation and the measurement

are setup accurately. In this investigation, the key factors for correlation are:

1. Match the simulation source to the measurement source

2. Generate sufficient mesh on the radiation boundary of the simulation

3. Calibrate the measurement carefully, including cable, horn, chamber and board

effects. The chamber loss is found to be the trickiest part of the calibration.

Simulation results across various connectors and multiple source configurations are

completed. These results show some differences in connector radiation at frequencies

below 10GHz, and at strong resonance frequencies. Above 10GHz, all connectors look

similar. At resonance frequencies, the connectors show higher a radiated energy peak,

corresponding to the signal source. For instance, a connector structure with a common

mode resonance shows a radiated energy peak when driven by a common mode signal,

but not when driven by a differential signal. At low frequencies, the connector with a

large, simple ground structure has the lowest radiated energy, while the connector with a

large, multi-pin, ground structure has the higher radiated energy. With sources at

25Gbps, the data suggest that all connectors analyzed probably need a shield to meet

FCC EMI specs. At 12.5Gbps, one connector has radiated energy that meets the

estimated TRP target. With many sources at multi-Gbps data rates, the simulation data

suggests it is not likely that an unshielded, open air connector will meet FCC EMI specs.

Using readily available S-parameters, one can generate an estimate of radiated power

by calculated all the energy that doesn’t exit the ports of the model. Then, by subtracting

a typical smooth curve of conductive and dissipative loss, the radiated power can be

estimated. This radiated power estimate is reasonable for sources that match the s-

parameter source, which in most cases is single-ended. It is not accurate to apply this

method to generate estimates of differential or common mode radiated energy from a

single-ended s-parameter file. Further study is needed to evaluate whether a single-ended

s-parameter file can be converted to a differential/common s-parameter file and then used

to generate reasonable estimates of differential and common mode radiated energy.

Results show that single-ended sources, at relatively high voltage amplitudes and at

data rate of a few Gbps, can be the dominant radiated energy source up to about 5GHz.

Differential and common mode sources can have similar radiated energy or significantly

different radiated energy, depending on the connector geometry. Simulations confirm

that tens of millivolts, on the outer shield structure, are likely to be the dominant radiated

energy sources in a system.

References

[1] Zhang Li ; Pingfang Yu ; Chua Chee-Parng, “The EMI characteristics of High

Speed backplane connector,”

[2] Xiaoning Ye, James L. Drewniak, Jim Nadolny and David M. Hakanson, “High-

Performance Inter-PCB Connectors: Analysis of EMI Characteristics,” IEEE

Trans. Electrogmagnetic Compatibility, vol. 44, pp. 165–174, 2002.

[3] Xiaoning Ye, Jim Nadolny, James L. Drewniak, Richard E. DuBroff, Thomas P.

VanDoren and Todd H. Hubing, “Experimental and FDTD study of the EMI

performance of an open-pin-field connector for modules-on-backplanes”, IEEE

Trans. Electromagnetic Compatibility, vol. 2, pp. 789-794, 2000.

[4] Xiaoxia Zhao, Angela Li, Hongmei Fan, Alpesh Bhobe, Kam Taunk, Jinghan Yu,

Philippe Sochoux, et al, “Validating EMC Simulation by Measurement in

Reverberation Chamber”, Proc. DesignCon, 2013

[5] Alpesh Bhobe, Mike Fogg, Steve Dunwoody, Rich Long, “Application of Full-

wave 3D Field Solvers to Predict EMI Behavior in SFP Cages”, Proc.

DesignCon, 2010