PRINCIPLES OF PHASE LOCKED LOOPS (PLL)

(TUTORIAL)

VĚNCESLAV F. KROUPA

INSTITUTE OF RADIO INGINEERING AND ELECTRONICS

ACADEMY OF SCIENCES OF THE

CZECH REPUBLIC

1

Introduction

In recent years, personal communications in high Megahertz and low Gigahertzfrequency ranges are booming. Behind this achievements was the technological

progress in integrated circuitry on one hand and application of frequencysynthesis on the other hand.

I. Principles

The task of the phase locked loops is to maintain coherence between input(reference) signal frequency, fi, and the respective output frequency, fo, via

phase comparison. The theory is explained in many textbooks [e.g., 1, 2] andpractically in all books on frequency synthesis. [3 through 10]. Here, we shall

repeat, in short, all major features with some new achievements.

A/ Basic equations

Each PLL loop works as a feedback system shown in Fig. 1.

Fig. 1 Basic feedback network of PLL

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To get more insight into the PLL properties, we shall simplify,without any loss of generality, the block diagram to that shown in Fig. 2.and introduce the Laplace transfer functions of the individual buildingcircuits - suitable for investigation of small signal properties.

Fig. 2 Simplified block diagram of the PLL with individual transferfunctions

Investigation of the above figure reveals that the input phase ŒŒŒŒi(t) iscompared with the output phase ŒŒŒŒo(t) in phase detector (ring modulator,

sampling circuit, etc.).

the proportionality factor, Kd [volt/2____], is called the "phase detectorgain."

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Next, vd(t) passes the loop filter, F(s)

where hf(t) is the time response of the loop filter. After applying v2(t) onthe frequency control element of the voltage controlled oscillator (VCO) weget the output phase

the proportionality factor, Ko [2____ Hz/volt], is the oscillator gain.

Since, in most cases, Kd and Ko are voltage dependent the generalmathematical model of a PLL is a nonl+inear differential equation. Itslinearization, justified in small signal cases ("steady state" working modes),provides a good insight into the problem.

the relation between input and output phase in the Laplace transform

The ratio, jjjjo(s)/jjjji(s), the PLL transfer function, is given by

where we have introduced the forward loop gain K =KdKo and the open loop gain G(s)

B/ Order of PLL

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In the simplest case there are no filters both in forward and feedback paths.

This phase lock loop is designated as the first order loopGenerally the denominator in H(s) is of a higher order in s and we speak

about PLL of the second, third, ect

C/ Type of PLL

The number of poles in the transfer function G(s), i.e. the number ofintegrators in the loop define the

type of the loop

D/ Phase error at the output of the phase detector (PD)

where

After elimination of jjjjo(s)

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By assuming the gain, G(s), as a ratio of two polynomials

where n is number of integrators in PLLwe get for the phase error

E/ Transient and steady state errors

Due to input phase steps, frequency steps, and steady frequency changes

After introducing any of the respective steps into (10 or 12) and performing the inverse Laplce transform we find the respective transients

With the assistance of the Laplace limit theorem we get for the final valueof the phase error

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F/ Block diagram algebra

Actual PLLs are often much more complicated than block diagrams in Fig. 1 or 2

For arriving at transfer functions,

|H(s)|2 and |1 - H(s)|2

we can apply the rules of the Block diagram algebra.

Investigation of the relation (5) reveals that the feedback blockcan be put outside of the basic loop.

In this way we arrive at the effective transfer functions,

|H’(s)|2 and |1 - H’(s)|2,

or

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Fig. 3 Simplification of the block diagrams of PLL: a/ series connection, b/ parallelconnection, c/ and d/ feedback arrangement, e/ more complicated system.

II. Phase locked loops of the 1st and 2nd order

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The most common PLLs are those of the 2nd order. Their advantage is theabsolute stability and simple theoretical and practical design.

A/ PLL of the 1st order.

their open loop gain is

with transfer functions

Note that DC gain KA can be used for changing the corner frequency, of this simplePLL, to any desired value - Fig. 4.

Fig. 4. The block diagram of the 1st order PLL

Since the open loop gain K has dimension of the 2____Hz normalization of the

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input or reference frequency in respect to it provides nearly all informationabout the behaviour of the PLL.

The transfer function H(jx) behaves as a low pass filter inrespect to the noise and spurious signals accompanying thereference signal whereas 1 - H(jx) as a high pass filter inrespect to the noise and spurious of the VCO.- see Fig. 5.

Fig. 5. Transfer functions Hi(jx) = 20log(|H(jx)|) and Ho(jx) = 20log(|1-H(jx)|)

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B/ PLLs of the 2nd order.

1st order PLL has only one degree of freedom, namely the DC gain

K=KdKoKA

Other difficulties are rather modest attenuation in the respective stopbands

only 20 dB/ decade.

This last problem can be removed with introduction of a suitable low passfilter into the forward path.

(1) A simple RC filter

In instances where we need to increase attenuation of the PLL for highfrequencies application of the simple RC low pas filter, provides the

desired effect. Note that the filter time constant T1

presents an additional degree of freedom

for the design of PLL properties.

Fig. 6. 2nd order PLL loop filters: a simple RC filter.

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The open loop gain is

the transfer function H(s) of the PLL

After introduction of the natural frequency qqqqn and the damping factor KKKK

we can rearrange the open loop gain into

and the PLL transfer function into its “characteristic form”

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After normalization of the frequency qqqq in respect to the natural frequency

we get for the open loop transfer function

and for the PLL transfer functions

.

Fig.7(a) Transfer functions Hi(jx) = 20log(|H(jx)|) and Ho(jx) = 20log(|1-H(jx)|), (b) phase characteristic of the open loop gain G(jx) of the 2nd order PLL loop

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with a simple RC filter.(2) Phase lag-lead or RRC filter (Fig. 8)

Fig. 8. 2nd order PLL loop filters: phase lag-lead or proportional - integral networks.

Transfer function of the RRC filter

provides a further degree of freedom. The open loop gain is

the respective transfer function

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We can again introduce the natural frequency and the damping factor

and arrive to the characteristic form the transfer functions

and to

Note that the freedom for independent choice of qqqqn and KKKK resulted in reduced slope ofthe stop band of H(jx) on one hand and in a reduced phase margin on the other hand

Fig. 9(a)Transfer functions Hi(jx) = 20log(|H(jx)|) and Ho(jx) = 20log(|1-H(jx)|), (b)phase characteristic of the open loop gain G(jx) of the 2nd order PLL loop

with an RRC filter.

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C/ PLLs of the 2nd order of the type 2.

The loop contains two integrators, the second one in the loop filter

Fig. 10 2nd order PLL loop filters: active phase-lag lead network ( dashed is one of the 3rd order loop configuration).

Its transfer function is

For operation amplifier (A>>1) the time constants are

and the open loop gain

Effective loop gain K=KdKAKo, however, for DC the gain is KDC=KdKAKoF(0)

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=KdKAKoA The PLL transfer function

the natural frequency qqqqn and damping KKKK

from which

Introduction of qqqqn and damping KKKK leads to the PLL transfer functions

After plotting the transfer functions Hi(x) and Ho(x) we find out that theycoincide with those plotted in Fig. 9 for the PLL with the gain K (high gainloops). However, we find a substantial difference with the phasecharacteristic which starts, due to the two integrators in G(s), at nearly -180 degrees. This is very important in instances with unintentionallyintroduced poles or delays, due to the use of sampled phase detectors, intothe loop gain G(s) since the stability of the system deteriorates. Theproblem will be discussed in the next sections.

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Fig. 11(a) Transfer functions Hi(jx) = 20log(|H(jx)|) and Ho(jx) = 20log(|1-H(jx)|); of the 2nd order PLL loop of the type 2;(b) phase characteristic of

the open loop gain G(jx).

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III. Phase locked loops of the 3rd order type 2.

Investigation of Figs 7, 9, and 11 reveals PLL of the 2nd order with simpleRC filter exhibits the slope of the transfer function

Hi(jx) in the stop band of -40 dB/dec. But the high gain RRC loops have the slope

of 40 dB/dec. in the stop band of the Ho(jx) transfer function.

The problem will be solved with introduction of an independent RC sectionin the loop filter F(s) in the type 2 systems

Note that even this 3rd order loop is unconditionally stable since G(s)exhibits a positive phase margin.

After introduction of the natural frequency qqqqn and the damping factor KKKK we get for the transfer function

The transfer functions together with the phase margin are plotted in Fig. 12

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Gom

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Fig. 12 Transfer functions Hi(jx) = 20log(|H(jx)|), Ho(jx) = 20log(|1-H(jx)|), andopen loop gain Go(jx) of the 3rd order PLL loop of the type 2; SSSS=.3 and KKKK=1.5.

Fig. 13 Properties of the 3rd order PLL for different damping constants ofthe original 2nd order loop and for different SSSS of the additional RC section:

a) phase of the open loop gain; b) magnitude of the overshoot Mp of thetransfer function 20log(|H(jx)|2).

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IV. Time delays in PLL.

A/ Simple time delaySimple time delay, gggg, is respected by multiplying the open loop gain by the factor

Evidently it only changes the phase margin. From Fig. 14 we see that its influencemight be considerable [11].

Fig. 14 Phase shift introduced by a simple normalized time delay qgqgqgqg.

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B/ Sampling

In modern technology many analog processes are replaced with digital processing

.This is also true for PLLs

The proper approach would be the investigation with the assistance of the z-transform .

The other possibility is to modify the original Laplace transform of G(s)

where

and

Evidently

and

The situation with the sampled PLL is illustrated in Fig. 15

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Hem

Hfm

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Fig. 15 a) block diagram of the PLL with sampling phase detector; b) the simulating analog system

Finally we arrive at the often suggested approximation of the sampling process,with the assistance of an additional RC section.

Fig.16Properties of the transfer function Hom= 20log(|Fh(s)|) compared with thatof a simple RC section Hfm = 20log[|1/(1+jqqqqT/2)].

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Fig. 17 Transfer functions Hi(jx) = 20log(|H(jx)|), Ho(jx) = 20log(|1-H(jx)|)and 20log(|G(jx)|)

of the sampled 3rd order PLL loop of the type 2 as in Fig. 12.

Note the reduced phase margin for the case where the ratio of the natural frequency qqqqn to sampling frequency qqqqs is

1:10

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V. Responses of PLL to the step and periodic phase and frequency changes.

The respective changes can be divided into three major groups:

1) Phase or frequency steps2) Periodic changes (spurious phase or frequency modulations,discrete spurious signals, etc.)3) Noises accompanying both reference and VCO signals

The information provides the phase difference at the output of the phasedetector jjjje(s) or more exactly ŒŒŒŒe(t). Since jjjje(s)/jjjji(s) = 1-H(s) we mustinvestigate the following relation

A) Step changes

(1) Phase step FkFkFkFki at the input of the phase detector (jjjji(s)=FFFF jjjji/s).

In the normalized form we have

Solution of the quadratic equation in the denominator reveals

After application of the Laplace transform tables (e.g. [12]) and the above roots we get

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Fig. 18 Normalized transients FkFkFkFke1(t)/FkFkFkFki due to the phase stepFkFkFkFki for different damping factors KKKK ; a) for simple RC loop filter;

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b) for high gain loop with lag lead RC filter.

(2)Frequency step FqFqFqFqi at the input of the phase detector (qqqqi(s)=FFFF qqqqi/s2).

After a step change of the division ratio N in the feedback path by FFFFN theeffective change of the “feedback reference frequency “ is FFFFfr = fr FFFFN/N. Theconsequence is the transient in the output phase kkkke(t).

Application of the roots from (52) and of the Laplace transform tables gives

Which simplifies for very high gain and the type 2 loops

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Fig. 19 Normalized transie

nts FkFkFkFke2(t)/(FqFqFqFqi/qqqqn) due to the frequency step FqFqFqFqi for different dampingfactors KKKK for high gain loop with lag lead RC filter; a) for simple RC loop

filter; b) for high gain loop with lag lead RC filter.

(3) A step of acceleration (frequency ramp) FFFF i (radians /s^2)

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In this case we get

After performing the inverse Laplace transform we arrive at

Fig.20

Normaliz

ed transients FkFkFkFke3(t)/(FFFF i/qqqqn2) due to the frequency ramp FFFF i for

different damping factors KKKK for high gain loop (DC phase error is retained).

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2) Periodic changes.

In these instances we are interested in settled or steady states

(A) Phase modulation of the input signal.

For simplicity we shall consider modulation with a single sine wave

The output modulation would remain sinusoidal,

however, shifted by the transfer function

In instances where PLL should be used as phase detector then the desiredinformation must be recovered at the output of the loop detector, however,

only for frequencies outside of the pass band ,i.e. for }}}}>qqqqn.

(B) Frequency modulation of the input signal.

By starting again with the sinusoidal modulation

which remains unaltered for modulation frequencies }}}}<qqqqn, however, onlyfor PLLs of the type 1.

The amplitude of the normalized phase at the output of the loop detector inthe instances of the PLLs of the type 2 is peaking for KKKK <1

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VI. Stability of PLL

Since PLL are feedback systems with the feedback transfer function G(s)they will oscillate whenever the gain G(s) is equal to minus 1, i.e.

This condition is met in instances where

i.e. for

and

Investigation of the 1st and 2nd order loops reveals unconditionally stabile.

However, this need not be the case with higher order loops.

By taking into account that

condition (1) depends on character of the polynomial P(s)

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A) Hurwitz criterion of stability

We shall write a determinant FFFFn from the coefficients of thepolynomial Pn(s) in accordance with the following rules:1) We start the first column with an-1 and proceeds with an-3, etc. in rows

below2) We start the second column with an and proceeds with an-2, etc. in rows

below3) We start the 3rd and 4th column with zeros but further apply the 1st and

2nd columns4) We start the 5th and 6th column with two zeros but further apply the 1st

and 2nd columns, etc5) We finish as soon as the determinant has n columns and n rows.

We evaluate all principle minor subdeterminants (minors) FFFFi; if theyall are larger than zero the feedback system is a stable one.

B) Computation of the roots of the polynomial P(s).

If real parts of all roots are negative the loop is stable.

C) Expansion of the function 1/[1+G(s)] into a sum of simple fractions

Investigation of the function 1/[1+G(s)] reveals that that it is equal to theratio of two polynomials R(s)/S(s)

where s1, s2, ...sn, are roots of the polynomial S(s).

Application of the tables with Laplace transform pairs provides solution inthe time domain. Another procedure is in changing the above relation into asum of simple fractions with constants in the nominators, i.e.

D/ The root-locus method

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Root-locus method of the function 1+G(s) is intended to find locationof the respective roots in the complex plain.

At present computer solution of the polynomial of Pn(s), with thechanging parameter K or any other, provides us with a set of roots whichcan be thereafter plotted in the complex plain.

Example: We will plot roots of the 2nd order PLL with the open loop gain

The polynomial for computation of roots is of the 2nd order

The above equation is that of the circle with the center, -1/T2, 0, and theradius r2 = 1/T2

2 - 1/T1T2. After introducing the loop parameters qqqqn and KKKKthe root locus is their function.

Fig. 25 Root locus of 1+G(s)for the 2nd order PLL type 2 with the RRC loop filter

E/ Frequency analysis of the transfer functions - Bode plots

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Transfer functions of individual PLL blocks provide information about allimportant properties of

Phase-lock loops enclosing stability.

1/ Frequency independent gain K=KdKaKo

2/ Factor with one zero in the origin jqqqq

3/ Factor with one pole in the origin 1/jqqqq

4/ Factor with one zero 1+jqqqqTo

5/ Factor with one pole 1/(1+jqqqqTo)

6/ Time delay exp(-jqtqtqtqt)

7/ A quadratic transfer function which can be encountered both in thenominator and denominator

[(jqqqq)2 + 2jKqKqKqKqn +qqqqn2 ]±1

In the earlier and often in the contemporary literature stability of thePLL systems is investigated with the simple Bode plots in accordance withthe old tradition of servo systems.

However, application of modern computers provides more insight and moreprecision solutions.

Nevertheless, for the sake of completeness we shall repeat here somebasic rules for construction of the Bode plots. After computing logarithm ofthe open loop gain we get

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Fig. 26 Bode plots for 1/ Frequency independent gain K=KdKAKo ,2/ Factor with one zero in the origin jqqqq, 3/ Factor with one pole in the origin

1/jqqqq: A/ Decibel gain, B/ phase

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Fiug. 27 Bode plot of the function 1+jqqqqTo

Fiug. 28 Bode plot of the function 1/(1+jqqqqTo)

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In Fig. 29 we compare an old Bode plot construction with

the computer drawing.

a)

Fig. 29 Bode plot of the 3rd order type 2 PLL

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VII. Phase locked loops of the 4th and higher orders.

We have seen that the 2nd and 3rd order loops were unconditionally stable

However, we often introduce intentionally additional filtering sections toimprove properties of PLL’s but the stability is endangered.

A/ Twin-T RC filter

In instances where we need large attenuation at a specific frequencyaddition of the Twin-T RC filter, shown in Fig. 30 may solve the problem.

Fig. 17 Twin-T RC filter

This network exhibits “infinite attenuation” for the following arrangement

After introducing following relations

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we get for the “resonant” frequency, qqqqrf,

For the input resistance Ri << R and the output resistance Rout >> R

The transfer function of the Twin-T is

Fig. 18 Transfer function and phase characteristic of the Twin-T filter

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Investigation of the properties of these PLL will be started with the normalized open loopgain of the second order loop-type two, G2(jx),

and thereafter by adding additional gains as that of Twin-T,

GT(jx), and sampling Ge(jx)

where we have introduced the “resonant” frequency frf and the sampling frequency fs

The overall open loop gain

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The transfer functions Hi(x) and Ho(x) are plotted in Fig. 32 togther withthe open loop gain G(jx) and the phase margin ||||(jx) for LLLL =.1 and TTTT = .05;

Note that the phase margin is small, 20 deg., only. In addition both transferfunctions have peaks of about 10 dB which indicates under damping.

Fig. 32 Transfer functions Hi(x) = 20log(|H(jx)|), Ho(jx) = 20log(|1-H(jx)|)and20log(|G(jx)|) of the sampled 4rd order PLL loop of the type 2 with

additional Twin-T filter with parameters LLLL =.1 and TTTT = .05

B/ Active 2nd order low pass filter

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From different configurations we shall investigated the only one shown

Fig. 33 Active 2nd order low pass filter

Its transfer function with a very large gain of the operation amplifier is

After introduction of the natural frequency qqqqnf

and damping d

we get for the transfer function (in the normalized form)

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which is plotted in Fig. 34 for different damping constants together with the respectivephase characteristics.

Fig. 34 a/ Transfer functions of the active 2nd order low pass filter; b/ its phasecharacteristics.

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Fig. 35 Transfer functions Hi(x) = 20log(|H(jx)|), Ho(jx) = 20log(|1-H(jx)|)and 20log(|G(jx)|)of the sampled: a) 4th order PLL loop of the type 2 with additional 2nd order low pass

filter with parameters ???? =.1 and d = .6; b) of the 5th order PLL with parameters ???? =.1, d= .6 and SSSS=.2.

C/Phase lock loop of type 3

Loop of the type 3 are encountered rarely for special services only. For the sake of

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simplicity we will consider two active RRC filters (see Fig. 10) in series.

After introducing the natural loop frequency qqqqn and the damping factor KKKK we can changethe above relation into

A typical transfer functions with the respective phase characteristic are

Fig. 36 Transfer functions Hi(x) = 20log(|H(jx)|), Ho(jx) = 20log(|1-H(jx)|2)and20log(|G(jx)|) of the 3rd order PLL loop of the type 3 with two additional 2nd order low

pass filters and with parameters EEEE = .5 and KKKK =.7

VIII. Noise properties of PLLRandom fluctuations of phase and amplitudes (generally designated as noise) of frequency

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generators are often limiting factors for many applications even in PLL’s.

Due to the limiting processes we can consider only

where

A/ Basic frequency instability measures in the frequency domain

1) Phase measures

The autocorrelation of the random phase departures kkkk(t) is defined

and the respective Power Spectral Density (PSD) S’kkkk(qqqq) (primed

indicate two sided spectra) is

We often encounter another definition, i.e. ©©©©(f), definingration of the phase power at frequencies fo±f in the 1 Hz bandwidth(where f is the so called Fourier frequency) in respect to the wholepower of the investigated signal

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where Skkkk(f) is the so called one sided PSD.

2) Frequency measures

In contradistinction to the uncertainty about the first moment of phasefluctuations, the first moment of frequency fluctuations can be put to zero

However, this is not the case with the 2nd moment which can be defined as

A further simplification will be achieved by normalizing frequencyfluctuations in respect to the carrier frequency iiiio = fo

Relation between the Power Spectral Density (PSD) Sy(f) and S

kkkk(f)

B/ Basic frequency instability measures in the time domain

At very low frequencies direct evaluation of phase PSD is difficult. The

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problem is solved with sample variances which provide other and veryeffective frequency stability measures. Nevertheless, in actual practice weencounter the Allan variance ( two sample variance) defined as

or the modified Allan variance

where

Frequency stability defined in the frequency and time domain measures arerelated with the assistance of a transfer function

The difficulty is that we can evaluate the integral in (102), in the closed form,only for a very particular form of Sy(f), namely a piece-wise linearized

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Fig. 37 Piece- wiselinearized noise characteristic of a 5 MHz crystal

oscillator

Note two dB measures on the vertical axis: the on the r.h. side are values of Sk (f), however, that on the l.h. side retains slopes of the Sk (f), but it isinvariant in respect to the carrier frequency as Sy (f). Consequently we cancompare noise characteristics of different generators in one and the same figure.

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All important noise processes, generally encounteredby evaluating the frequency instability, are

the random walk of frequency with the noise constant h-2

the flicker frequency noise with the noise constant h-1

the white frequency noise with the noise constant ho

the flicker phase noise with the noise constant h1

the white phase noise with the noise constant h2Sy(f) cy

2(g) Mod cy2(g)

h-2/f2 (2_)2gh-2/6 K 5.4ngoh-2

h-1/f 2h-1ln(2 ho/2g) K 0.94h-1

ho ho/2g Kho/4ngo

h1f h1(2_g)-2[1.38+3ln(qHg)] K.084h1/(ngo)2 h2f2 3h2fH(2_g)-2 fH/n(2_ng)2

K.076goh2fH/(ngo)3

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C/ Noise in oscillators

1/ Crystal oscillatorsThe resonator circuit exhibits the flicker and white noise

where Pr is the dissipated power and ar the flicker noise constant.

The noise of the maintaining circuit is

Finally, we arrive at the PSD of the oscillator phase noise where we have introduced the unloaded QU by putting 2QLKKKKQU.

------------------------------------------------------------------------------------------------

The magnitude of ar can be appreciated from noise measurements performedon quartz resonators. Its value was found approximately to be arKKKK10-12.75.After introducing this value, together with the quartz material constant,

foQU KKKK 1.3*1013, we get

and the plateau in the Allan variance is approximately for all quartz crystalresonators (since generally ar > ae)

51

(109)

2/ LC oscillators

The relation (106) is also valid for LC oscillators. For the mean values wecan write

Note that the coefficients hi (see 103) are mean values form experimentalmeasurements. Actual noise coefficients can differ by -2 to +1 order

For a preliminary estimation of the oscillator noise both crystal and LC wecan use the following diagram

Fig. 37 Noise characteristic of oscillators with parameters QL and fo.

52

(110)

(111)

D/ Noise in digital frequeny dividers

We provide practical formulae for a preliminary estimation of the outputnoise of digital dividers.

For TTL and ECL divider family

For GaAs divider family the above relation requires only a small correctionin the first term

We expect that these formulae can be also used for appreciation of the noisequality of actual devices.

a) b)

Fig. 38 a) Flicker phase noise of TTL and ELC digital dividers b) white phase noise of TTL and ELC digital dividers.

53

(112)

(113)

E/ Noise in Phase detectors and amplifiers

For a preliminary estimation we can apply an experimentally found relation

F/ Noise in loop filters After comparing actual PLL PSD S

kkkk,L (in the white noise region) withmagnitudes added by dividers S

kkkk,D and Skkkk,PD we find out that its level is

orders higher; the reason is Johnson noise generated in the filter resistors. ConsequentlyExample : KKKK = .1, Kd = 5/2____, Cmax = 10-6, T1/T2 = 10: S

kkkk,L KKKK 10-13/fn

Fig. 39 PSD Skkkk,L of the additive noise of different PLL’s together with

practical (full line) and theoretical limits (RRRR).

54

G/ Noise in PLL

We shall start from a rather general PLL arrangement.

Fig. 40 Block diagram of a general PLL with additive noise sources.

55

(114)

(116)

(115)

(117)

By assuming a locked loop we can write with the assistance of the Laplacetransform for the linearized arrangement

where

Since most of the noise components are random by nature and uncorrelatedthe PSD of the PLL output phase is

All the additive noises, due to the phase detector, loop frequency dividers,loop amplifiers, and loop filters can be summarized into a PSD SL(f)

56

Phase locked loop fi 9.976 106.= fo 9.491 107.= fn 861.078=

ζ .7 Kd 1.92 π.

Ko fo10 Qo.

M 0 δ .1 κ .2 Kd 0.302=

Ko 1.898 104.=r 2 Dr a4 r. fr fi

DrN fo

fr Dr 4=

α 0.15 d 0.6 fr 2.494 106.=

xm

fm

fnGm

j xm. 2. ζ. 1

j xm. 2

Gam1

1 j xm. α. 2. d. j xm

. α. 2 Gem 1 ej xm

. δ..

g3m1

1 2 j. ζ. xm. κ.

G5m Gm Gam. Gem

. g3m. H5m

G5m

1 G5m

Sφm10 13

fm

10 9

fn Hφm 10 log SφmNfo

2..

hvm him M NDr

2. fi

fo

2. Sφm

Nfo

2. 1. H5m

2. hom 1 H5m2. 1.

hout m 10 log hvm.

Soutm hout m 10 log fo2.

0.01 0.1 1 10 100 1 .103 1 .104 1 .105 1 .106 1 .107170

150

130

110

90

70

50

30

10

10

Soutm

Sim

Som

10 log Sφm.

20 log H5m.

20 log 1 H5m.

fm

Fig. 41 Output noise of 100MHz VCO locked to a 10 MHz crystal oscillatorvia a 5th order loop investigated in Fig. 35.

57

(115)

(116)

(117)

(118)

(119)

IX. Acquisition

Working ranges of PLL

1) Hold–in range FqFqFqFqH

2) Pull-in range FqFqFqFqP

Let us assume that the difference between the reference frequency qqqqi andthe free running VCO frequency qqqqc is larger than FqFqFqFqP the result is a beat

evidently

After taking into account the feedback properties of PLL and the principleof the harmonic balance we get for the 2nd order type 2 loops

Minimum of the above relation reveals

and for the pull-in range we get

58

(121)

(122)

(123)

(120)

a) 2nd order simple DC filter

b) Lag lead (RRC) filter ( PLL type 1)

c) Lag lead (RRC) filter ( PLL type 2)

d) Lag lead (RRC) filter ( PLL type 2) with time delay

Note that it exists certain delay for which the pull-in range is zero. This isillustrated with Fig. 42 and 43. The oscillating branch in Fig. 42 indicates the possibility of false locks.

59

Fig. 42 Normalized detuning xc=iiiic/qqqqn as function of x=iiii/qqqqn for PLL ofthe 2nd order type 2 for two amplifier gains and different delays

60

(125)

(126)

(127)

(128)

(129)

Fig. 43 Normalized pull-im range xP = FqFqFqFqP/qqqqn for PLL of the 2nd order type 2 for two amplifier gains and normalized delay

(points were found by computers).

3) Lock-in range FqFqFqFqL

Acquisition is expected without cycle slipping. This condition is met withzero beat note at the output of the PD

a) PLL of the 1st order

b) PLL of the 2nd order with RC filter

c) PLL of the 2nd order with RRC filter (high gain loops)

4) Pull-out frequency FqFqFqFqPO

From investigation of the transients due to the frquency step we get for itsmaximum

61

(131)

(130)

and finally

5) False locks

In some instances the pull-in process may result in The principle can be explained with the assistance of the following figure

Fig. 44 Block diagram of the PLL with the beat note a bit smaller thanFqFqFqFqP

For the slowly varying detuning Fq we have

Additional filtering or time delays may cause nnnn >____ /2 which will changethe sign of the slowly varying tuning voltage u2(t) and starts to push theloop out of lock and in some instances lock the VCO on a false frequency cf.Fig.45

62

(132)

Fig. 45 The DC component FqFqFqFq/K in the pull-in process: a)PLL of the 4th order Type 2 with two additional sections RC;

b) PLL of the 3rd order Type 2 with additional time delay (KKKK=.7, SSSS=.3).

6) Pull-in time

Solution will start with the simplified block diagram in Fig. 44. Note thatthe AC path is responsible for the magnitude of the beat note YYYYc.Furthermore we will assume the 2nd order loop with RRC filter with thereduced gain

63

(133)

Finally we arrive at an approximate pull-in time for the sine wave PD

with slightly different values for othertypes ofphase detectors - seeFig. 46

64

Fig. 46 Asymptotic approximations of the pull-in time for PLL of the 2nd order : a) for a simple phase detector; b) for a phase-frequency

detector for two different damping constants

References:[1] F.M. Gardner: Phaselock Techniques. New York: J. Wiley, 1966, 2nd

ed 1979[2] W.F. Egan, Frequency Synthesis by Phase Lok, New York: J. Wiley,

1998, 2nd ed. 2000.[3] V. F. Kroupa, Frequency Synthesis: Theory, Design et Applications.

London: Ch. Griffin, 1973; New York: J. Wiley, 1973.[4] V. Manassewitsch, Frequency Synthesizers, Theory and Design. New

York: Wiley, 1976, 1980.[5] W.F. Egan, Frequency Synthesis by Phase Lock. New York: Wiley 1981.[6] U.L. Rohde, Digital PLL Frequency Synthesizers, Theory and Design.

Englewood Clifs: Prentice Hall, 1983.[7] J.A. Crawford, Frequency Synthesizer Design Handbook.

Boston/London: Artech House, 1994.[8] Bar-Giora Goldberg, Digital Techniques in Frequency synthesis. New

York: MacGraw-Hill, 1996.[9] U.L. Rohde, Microwave and Wireless Synthesizers, Theory and Design.

John Wiley 1997. [10] V.F. Kroupa , ed. Direct Digital Frequency Synthesizers. IEEE Press 1999.

[11] V.F. Kroupa, Theory of Phase-Locked Loops and Their Applicationsin Eectronics, Praha: Academia 1995 (in Czech).

[12] G.A. Korn and T.M. Korn: Mathematical Handbook for Scientistsand Engineers. New York : McGraw-Hill, 1961.

[13] E.J. Angelo, “A Tutorial Introduction to Digital Filtering,” The BellSystem Technical Journal, Vol. 60, No. 7, September 1981.

Acknowledgment.

This work has been supported by the Grant Agency of the Czech republic

65

under the contract No. 102/00/0958.

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