PrinciplefortheRealizationofDual-OrthogonalLinearly...

Hindawi Publishing CorporationInternational Journal of Antennas and PropagationVolume 2011, Article ID 231893, 11 pagesdoi:10.1155/2011/231893

Research Article

Principle for the Realization of Dual-Orthogonal LinearlyPolarized Antennas for UWB Technique

Grzegorz Adamiuk, Mario Pauli, and Thomas Zwick

Institute of High Frequency Techniques and Electronics (IHE), Karlsruhe Institute of Technology (KIT), Kaiserstraße 12,76131 Karlsruhe, Germany

Correspondence should be addressed to Grzegorz Adamiuk, [email protected]

Received 20 May 2011; Revised 26 July 2011; Accepted 27 July 2011

Academic Editor: Zhongxiang Q. Shen

Copyright © 2011 Grzegorz Adamiuk et al. This is an open access article distributed under the Creative Commons AttributionLicense, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properlycited.

A concept of an array configuration for an ultrawideband suppression of the cross-polarization is presented. The method is ex-plained in detail, and a mathematical description of the principle is given. It is shown that the presented configuration is convenientfor the development of very broad band, dual-orthogonal, linearly polarized antennas with high polarization purity. The inves-tigated configuration shows a high decoupling of the orthogonal ports and is capable for antennas with a main beam direction per-pendicular to the substrate surface, that is, for a planar design. The phase center of the antenna configuration remains fixed at onesingle point over the complete desired frequency range, allowing a minimum dispersion of the radiated signal. The influence ofnonidealities in the feeding network on the polarization purity is investigated. The presented method introduces a superior pos-sibility of an extension of typical UWB technique to fully polarized systems, which improves significantly performance in, forexample, UWB-MIMO or UWB-Radar.

1. Introduction

The ultrawideband (UWB) mask as it is defined by the FCC[1] allows the licence-free usage of the spectrum between3.1 GHz and 10.6 GHz with the maximal power spectraldensity of −41.3 dBm/MHz. This bandwidth opens manypossibilities, not only in communication systems with poten-tially high data rates but also in localization and radarsystems, where the accuracy or resolution is directly propor-tional to the bandwidth. On the other hand, the large band-width is challenging regarding the antenna design. Thisis especially true for dual polarized antennas. The latestresearch in the ultrawideband technology shows an increas-ing demand on ultrawideband dual polarized antenna ele-ments [2]. For many applications, polarization diversity isof big importance. In case of single polarization, there couldbe a degradation of the performance if transmit and receiveantenna are cross-polarized, especially in a line-of-sight en-vironment. Polarization diversity can also enhance the per-formance of radar and imaging systems [3–5] and is nearlyindispensable in the MIMO technique.

In literature, only few dual-orthogonal polarized ultrawideband antenna concepts can be found. Most of them ra-diate the electromagnetic wave over a very widebandwidthusing the traveling wave principle like the well-knownquad-ridged horn antenna [6], the dielectric rod antenna[7] or the tapered slot (a.k.a. vivaldi) antenna [8]. How-ever, such antennas are generally bulky and difficult tointegrate into any device. Furthermore, only few antennasoffer a polarization purity of better than 20 dB over thebandwidth of several GHz. All antennas based on travelingwave principle need generally a 3D structure in order toperform correctly. Such design makes an integration of theradiators into any compact device difficult. For this reason,a planar solution with a high cross-polarization suppressionhas been investigated to assure a wideband radiation of twoorthogonal linear polarizations. An important demand onthe antenna is a stable phase center of radiation, whichlocation is constant over the very widebandwidth. Thisguarantees small distortion of the radiated pulse in pulse-based UWB systems. The phase centers should possessthe same position for both polarizations, what assures the

2 International Journal of Antennas and Propagation

same radiation conditions independent of the polarizationmode. For the integration of the radiators into a device,the radiation pattern which is perpendicular to the antennasurface is advantageous. The main beam direction should beconstant over the frequency range. Similar beamwidths inthe E-plane and in the H-plane allow for the illumination ofthe same space independent of the polarization. This paperpresents a principle of an array configuration for the real-ization of ultrawideband dual-orthogonal, linearly polarizedantennas meeting the above-mentioned requirements. Thearray configuration for suppression of the cross-polarizationin a certain angular region is described. The principleintroduces simultaneously a method for stabilization ofthe movement of the phase center over the frequency. Inthe following, a mathematical description of the radiationcharacteristics is given, and the performance of the arrayconfiguration is evaluated. In the last part of the paper theinfluence of nonidealities (amplitude and phase balance) onthe overall performance of the proposed configuration isinvestigated.

2. Array Configuration

First, a single, arbitrary, planar antenna element, marked inFigure 1(a) as a blue surface, is considered. The target polar-ization is linear and conforms with z-axis. The radiatedelectrical field vector E (solid arrow) is oriented arbitrar-ily. Firstly, the orientation in the yz-plane is considered.The radiated E-field vector is distributed into copolarized(dashed arrow) and cross-polarized (dotted arrow). Thedesired linear polarization is parallel to the z-axis and can beeasily achieved by a rotation of the antenna. However, thisis not applicable in practice since the polarization may turnwith frequency. Furthermore, the rotation of the antenna isnot a solution in the case of an unintended elliptical polar-ization. To overcome these problems, a second mirroredantenna is used, as shown in Figure 1(c). The configurationis symmetrical to the y-axis. If both antennas are fed by asignal which is identical in amplitude and phase, the radi-ated polarization (solid arrow) from both elements is alsosymmetrical with espect to the y-axis. This means that thecross-polarization components (dotted arrows) are identicalin amplitude and phase and the co-polarization componentsare identical in amplitude but oriented in the oppositedirection. This results in a cancelation of the desired co-polarization. To overcome this drawback, a differential feed-ing of the antennas is employed, which is shown schemati-cally in Figure 1(d). It can be noted that through the imple-mentation of the differential feeding the cross-polarizationvectors are oriented in opposite direction and thereforeinterfere destructively, whereas the co-polarization ones arein the same direction and interfere constructively. Such con-figuration results in the radiation of the desired linearlypolarized E-field parallel to the z-axis. It is evident thatthe polarization is not depending on the electromagneticcharacteristics of the single antenna but only on the mechan-ical arrangement of the single antenna elements within thearray configuration. Noticeable is also that for the mainbeam direction (x-axis) the principle of cancellation of

1

z

yx

(a) Single element

1

2

z

yx

(b) Standard antennaarray

z

yx

1

2

(c) Antenna array withmirrored elements

1

2

z

yx

(d) Antenna array withmirrored, differentiallyfed elements

1

2

34

(e) Dual-orthogonal, linear polarizedantenna array with mirrored, differentiallyfed elements

Radiatedpolarization

Co-pol

X-pol

(f) Legend

Figure 1: Various antenna array configurations and their influenceon the polarization of the radiated wave.

cross-polarization with simultaneous enhancement of co-polarization is independent of the wavelength. Hence, it issuitable for UWB technique.

To obtain two linear orthogonal polarizations, the an-tenna array has to be upgraded by two additional elementsrotated by 90◦ as shown in Figure 1(e). If both subarrays areoperated independently, two linear orthogonal polarizationscan be radiated independently.

Depending on the antenna type, the position of the phasecenter can be a function of the frequency, which results in anunwanted dispersion of the radiated pulse [9]. This variationof the phase center is schematically shown in Figure 2(a),where the phase center at a certain frequency fi is representedby the black dot. Within an antenna array with two elementsthe phase center is always located in the middle between the

International Journal of Antennas and Propagation 3

z

yx

f1

f2f3

(a) Single element

z

yx

f1

f1

f1

f2

f2

f2

f3

f3

f3

(b) Array with axis-symmetrical elements

Figure 2: Variation of the phase center (black dot) of the radiator atdifferent frequencies fi.

phase centers of the two elements (regarding the main beamdirection). Thus, in an antenna array with a symmetricalarrangement of the second radiator, the variation of thephase center is limited to the symmetry axis (here y-axis),what is shown in Figure 2(b) [10]. This results in a par-tial compensation of the dispersion characteristics of theantenna. This principle is valid as long as the antennas haveidentical electromagnetical properties and do not couplemutually. If the elements have a phase center that is onlyvarying along one axis, with the proposed arrangement thephase center can be fixed at one single point over a wide fre-quency range. If a second antenna pair which is rotatedaround this point is used, a dual-polarized radiator with anidentical, frequency-independent phase center for both po-larizations is obtained.

Similar configurations have been shown in the literature[11, 12]. The authors concentrated, however, on narrowband antennas. The independency of the principle of the fre-quency and the advantages regarding the stabilization of thephase center have not been fully revealed.

3. Mathematical Description of Radiation

In the following, a mathematical model for the radiationproperties of a mirrored, differential-fed antenna array isderived. The antenna and the corresponding coordinate sys-tem is shown in Figure 3(a). The antenna placed in the upperhemisphere (z > 0) is indexed as antenna “1” and the one inthe lower hemisphere (z < 0) as antenna “−1”. The feedingvoltages of both radiators are differential to each other andnormalized to the input voltage. Hence, the feeding voltagecan be described by

U1 =1√2

, U−1 =1√2e jπ . (1)

The transfer function HAr(θ,ψ) of the array configura-tion can be determined as follows

HAr

(θ,ψ

) =∑

i

Hi

(θ,ψ

) ·Φi

(θ,ψ

), (2)

z

y

x

θr P(r, θ,ψ)

ψ

−1

1

(a) Mirrored antenna array elements in the coordinatesystem

d

P

r0

r0 −r2

Antenna 1 θ

z

x

ri

Antenna-1

(b) Configuration for the determination of the arrayweighting factor Φi(θ,ψ)

Figure 3: Coordinate system and configuration for the determina-tion of the array transfer function HAr(θ,ψ).

where Hi(θ,ψ) is the complex transfer function of the singleelement. Φi(θ,ψ) is a weighting factor which is derived fromthe definition of a typical array factor and is defined in thefollowing formula:

Φi

(θ,ψ

) = Uie− jβ0(r0−ri), (3)

r0 − ri = id

2cos θ, (4)

β0 = 2πλ0

, (5)

where ri represents the distance between the observationpoint P and the base point of the coordinate system, d is thedistance between the two antennas (cf. Figure 3(b)), and Ui

is the amplitude of the feeding voltage. Combining (2)–(5),the transfer function of the array HAr(θ,ψ) yields

HAr

(θ,ψ

) = H−1

(θ,ψ

) ·U−1e− jβ0(−1)(d/2) cos θ

+ H1

(θ,ψ

) ·U1e− jβ0(1)(d/2) cos θ ,

(6)


where H1 and H−1 are complex transfer functions of thecorresponding antennas. After simplification, rearrangementwith the exploitation of the geometrical symmetry of H1

and H−1, (6) can be written as functions of the polarizationcomponents

HAr

(θ,ψ

) =⎡

⎣H1,θ

(−θ,ψ)

uθ

−H1,ψ

(−θ,ψ)

uψ

⎤

⎦ · 1√2e j(1/2)β0d cos θ

+

⎡

⎣H1,θ

(θ,ψ

)uθ

H1,ψ

(θ,ψ

)uψ

⎤

⎦ · 1√2e− j(1/2)β0d cos θ ,

(7)

where uθ represents in this case the co-polarization unit vec-tors, and uψ represents the cross-polarization unit vector. Itcan be observed that the copolarized components uθ have thesame leading sign, and the cross-polarized components uψhave opposite leading signs of the respective summands inthe transfer function H. This results in constructive anddestructive interference of co- and cross-polarization, re-spectively.

To investigate the influence of the antenna arrangementon the corresponding components HAr,θ(θ,ψ) (co-pol) andHAr,ψ(θ,ψ) (x-pol), in the following, isotropic radiators areconsidered for the calculation of the transfer functionHAr(θ,ψ). One receives the array factor AF, which repre-sents the radiation characteristics independent of the usedelements. Writing (7) without the complex transfer functionsof the elements H1 and H−1, the array factor AF is obtained

AF(θ,ψ

) =⎡

⎣uθ

−uψ

⎤

⎦ · 1√2e j(1/2)β0d cos θ

+

⎡

⎣uθ

uψ

⎤

⎦ · 1√2e− j(1/2)β0d cos θ.

(8)

In case of two differentially fed antennas arranged symmetri-cally to the y-axis, the array factors for co-polarization (AFθ)and cross-polarization (AFψ) direction can be written as

∣∣AFθ(θ,ψ

)∣∣ =∣∣∣∣√

2 cos(

12β0d cos θ

)∣∣∣∣, (9)

∣∣∣AFψ

(θ,ψ

)∣∣∣ =

∣∣∣∣√

2 sin(

12β0d cos θ

)∣∣∣∣. (10)

4. Evaluation of the Radiation Characteristic ofthe Array

In this section, the influence of the previously describedantenna arrangement on the polarimetric radiation charac-teristics is evaluated. The array is arranged symmetricallyto the y-axis, and the co- and cross-polarized radiationproperties are described by (9) and (10), respectively. It canbe seen in these equations that the array factors depend onthe frequency, the distance between the antenna elementsd and the angle θ, that is, the observation angle (cf.Figure 3(b)). This means that in some angular regions theco-polarization is enhanced, and the cross-polarization is

suppressed and vice versa. This has to be taken into accountin the antenna design. Noticeable is lack of the dependencyon the angle ψ. It means that the radiation properties of thedescribed array are projected with the same quality on thewhole ψ-plane for certain angle θ.

The array factors |AFθ| and |AFψ| are shown inFigure 4(a) for an electrical distance del = d/λ0 = 0.5. Inthe angular region between approximately 60◦ and 120◦, theθ-component is amplified and the ψ-component is sup-pressed effectively. The maximum gain is 3 dB at an angle ofθ = 90◦. At this angle, the cross-polarization is nearly com-pletely suppressed. The angular dependencies of the arrayfactors for electrical distances of del = 1 and del = 1.5 areshown in Figures 4(b) and 4(c), respectively. With increasingdistance between the elements, the beams become narrowerand grating lobes occur. In the main beam direction(θ = 90◦), the maximum of |AFθ| and the minimum of|AFψ| are independent of the electrical distance between theantenna elements. In this region, the θ-polarization is ampli-fied and the ψ-polarization is suppressed independent of thefrequency.

Figure 5 shows the mean array factor of the θ- and the ψ-component averaged over a bandwidth between 3.1 GHz and10.6 GHz. An effective suppression of the cross-polarization(|AFψ|) and an amplification of the co-polarization (|AFθ|)can be observed for the angles 75◦ < θ < 105◦. At 90◦, theamplification of the θ-component is 3 dB, and the suppres-sion of the ψ-component is maximal. The maximal ampli-fication of the cross-polarization is approximately 1.9 dB at60◦ and 120◦. It does not reach the value of 3 dB due to theangular movement of the maximum of the cross polarizedbeams over the frequency.

The diagrams in Figures 4 and 5 have been verified bythe authors in terms of simulations and measurements forthe monopolarized radiator. In the following section, ameasurement verification for the dual-polarized array ispresented.

5. Measurement Verification

The antenna for the verification is based on a well-knownelliptical antenna, also known as diamond-shaped antenna[10, 13]. The antenna is able to radiate over a very largebandwidth and possesses strong cross-polarization compo-nents in the H-plane. Hence, the properties of the radiatorare convenient for the verification of the principle. Theprototypes of the radiators presented in this paper arerelatively bulky and are chosen only for verification purposes,due to their radiation properties. For compact realizationswith stable radiation patterns over frequency and highpolarization decoupling, the reader is referred to other worksof the authors [14–17].

The prototype of the single radiator is presented inFigures 6(a) and 6(b). The radiating element consists of anelliptical opening in the ground plane in which an ellip-tical metallization is incorporated. The antenna is fed atthe slot between the inner ellipse and the ground plane.As a feeding network, a microstrip line with tapering isused.


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0 30 60 90 120 150 180

Arr

ayfa

ctor|A

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Arr

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|AFψ ||AFθ|

Angle θ (deg)

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Figure 4: Dependency of the array factors |AF| on the electrical distance del between the elements over θ.

The dual-polarized array consisting of two pairs (fourelements) of the previously presented antenna is shown inFigures 6(c) and 6(d). The elements are pairwise symmet-rical, and the symmetry points of both pairs are the same.As a feeding network also microstrip line is used. Since bothelements of the single pair must be excited at the sametime, a power divider must be applied. The outputs of thedividers are tapered and connected to the feeding points ofthe respective radiators. In order to be able to drive bothpolarizations independently, two feeding networks, each forone polarization, are applied.

The measured gain for the co-polarization in the H-planefor the single antenna and for the dual-polarized radiator(port 1) is shown in Figures 7(a) and 7(b), respectively.

In both cases, two beams at 0◦ and 180◦ are present,which yield an orientation perpendicular to the substrate.The similar widths of the beams are expected, since theaperture of the radiator is not extended in the H-plane.Noticeable is higher gain of the dual-polarized array, whichresults from the application of two elements, which extendthe aperture along the E-plane. For the comparison, themeasured results of the gain for the E-plane, co-polarizationare shown in Figures 7(c) and 7(d), for the single element,and array, respectively. For the single element a not sym-metrical radiation pattern with respect to 0◦ over frequencyis observed. This is a result of not symmetrical antennageometry along this plane. Due to extension of the radiatorto an array, a symmetrization of the current distribution


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0

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0 30 60 90 120 150 180

|AF m|in

dB

Angle θ (deg)

|AFψ,m||AFθ,m|

Figure 5: Mean array factor |AFm| in dependency of θ.

(a) Single antenna, model [10] (b) Single antenna, prototype

Port 1

Port 2

(c) Array, model (d) Array, prototype

Figure 6: Single element and dual-orthogonal, linearly polarized,differentially fed antenna array for the verification.

along the E-plane is achieved; hence, the radiation patternis more symmetric. Outside of the main beam direction(0◦), side lobes are present, which are result of the rela-tively high distance between the elements. Due to previ-ously mentioned extension of the aperture in the E-plane,a narrower beam is observed in the E-plane than in the H-plane. With the measurements, it has been shown that the

array is able to enhance the co-polarization. For the full ver-ification of the principle, a destructive influence of the con-figuration on the cross-polarization has to be included.

In Figures 8(a) and 8(b), the measured gains in the cross-polarization, H-plane are shown for single element and array,respectively (please note the scale). In the case of singleelement-four strong cross-polarized beams are present.These are extensively suppressed in the case of the array.Some stronger cross-polarized components in the angularregion |θ| > 90◦ are a result of radiation from the feedingnetwork of the array and can be significantly suppressed byshielding or compact realization as shown by the authors,for example, in [14, 15]. For the completeness in Figure 8,the measurement results of single antenna and array for theE-plane, cross-polarization are shown. No significant nega-tive effects considering the polarization purity are observed.

A further advantage of the presented antenna configu-ration is the high decoupling between the pairs of ports fororthogonal polarizations. This occurs due to the annihilationof the electric fields at the ports for orthogonal polarizations.This effect is described and verified by the measurements in[14, 15]. In these publications, an important issue of minia-turization of the antenna array basing on the introducedprinciple is also presented.

It can be concluded that in certain angular regions, thecross-polarization can be successfully suppressed indepen-dent of the frequency. The size of the useful region is largerwith decreasing frequency and distance between the ele-ments. Hence, for the given frequency range, the distancebetween the mirrored elements should be kept small in termsof wavelengths. The concepts for the compact design of theradiator basing on the described principle are described bythe authors in [14, 15]. Due to the same aperture size of thecited antennas in the E- and H-planes, the radiators possessnearly the same beamwidths in the respective planes over thefrequency range from approximately 3 GHz to 11 GHz. Thisintroduces a significant advantage for the dual-polarizedantennas, since the same space is illuminated independent onthe polarization. The examples from the literature [14, 15]show an example of practical implementation of the here-described principle in the compact dual-orthogonal, linearlypolarized UWB antennas.

The principle is applicable for nearly all types of radi-ators. However, it has to be kept in mind, that prior tothe design of the antenna array and the selection of thesymmetry plane an intensive investigation of the radiationcharacteristics of the single element has to be performed.This knowledge is indispensable for the proper arrangementof the elements and successful suppression of the cross-polarization.

6. Requirements on the Feeding Network

In the theoretical investigation, the differential feeding of theantennas was considered as ideal, that is, the signals feed-ing single elements were exactly equal in amplitude, and thephase shift was exactly 180◦. However, this condition may notbe met in reality due to the tolerances in the manufacturingprocess or a nonideal design of the feeding network. Hence,


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Figure 7: Measured gain of the single element (Figure 6(b)) and array (Figure 6(d)) in the co-polarization, E- and H-plane over the frequen-cy.

the influence of imperfect feeding signals on the antennaperformance is investigated. Both, the imbalance in ampli-tude and the variation in the phase of the feeding signal areconsidered.

6.1. Amplitude Imbalance. A variation of the impedances inthe differential power divider or resonances in the antennasystem could lead to different power levels at both antennaelements. This influence is investigated in this section. It isassumed that the phase difference is exactly 180◦. The am-plitude balance Δa between the feeding signals is defined as

Δa = U1|dB − U−1|dB. (11)

To investigate the influence of Δa on the array factor, anomnidirectional radiation pattern of the elements is

assumed. In this case, (8) can be written as

AF(θ,ψ

) =⎡

⎣uθ

−uψ

⎤

⎦ · Δa · 1√2e j(1/2)β0d cos θ

+

⎡

⎣uθ

uψ

⎤

⎦ · 1√2e− j(1/2)β0d cos θ.

(12)

The fixed electrical distance between the elements of del =0.5 is chosen for the investigations (cf. Figure 4(a)). Thecalculated array factors |AFθ,ψ| for different Δa are shown inFigure 9. The resulting characteristics for varying amplitudebalances of Δa = −3 dB, Δa = −1.93 dB, and Δa = −0.91 dBare presented in Figures 9(a), 9(b), and 9(c), respectively. Theforms of the curves remain essentially the same, howeverthe amplitude of the array factor changes. This is especiallyvisible in the main beam direction, that is, θ = 90◦ for |AFψ|.This happens due to nonideal compensation of the cross-polarization radiated from one element by the second one.


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Figure 8: Measured gain of the single element (Figure 6(b)) and array (Figure 6(d)) in the X-polarization, E- and H-plane over the frequency.

This is a direct influence of the amplitude differences at bothof the antennas.

In order to investigate the limits of the principle, the arrayfactor for the cross-polarization |AFψ| at θ = 90◦ is con-sidered. Figure 9(d) shows the dependency of the arrayfactor on the amplitude balance Δa at θ = 90◦. If fora proper functionality a value of −20 dB is assumed, themaximum amplitude balance has to be below Δa = −1.3 dB,which equals a linear value of 0.86. In this case, the co-polarization is amplified by 2.4 dB which is 0.6 dB less com-pared to the ideal case of identical amplitudes of the feedingvoltage.

6.2. Phase Imbalance. The phase of a signal is very sensitiveto, for example, coupling effects or resonances and can thusbe easily influenced. Furthermore, during the manufacturingprocess, slight changes in the length of the antenna feedinglines can occur. The influence of these effects is investigatedin this section under the assumption that the amplitudes ofboth signals are identical (Δa = 0). The phase balance Δφ,

that is, a deviation from the ideal phase difference of 180◦, isdefined as

Δφ = ∠(ΔU1)−∠(ΔU−1)− π. (13)

Equation (8) can be written with consideration of Δφ as

AF(θ,ψ

) =⎡

⎣uθ

−uψ

⎤

⎦ · 1√2e j(1/2)β0d cos θ · e jΔφ

+

⎡

⎣uθ

uψ

⎤

⎦ · 1√2e− j(1/2)β0d cos θ.

(14)

Figure 10 shows the array factors |AFθ,ψ| as a function of thephase balance Δφ. For a proper comparison, the electricaldistance between the elements is again set to del = 0.5. Thephase deviation is considered to be 30◦, 20◦, and 10◦. Thecalculated results are shown in Figures 10(a), 10(b), and10(c), respectively. The phase deviation between the feedingsignals causes an angular shift of the minima and maxima


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ctor|A

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dB

|AFθ||AFψ |

0 30 90 120 150 18060

Angle θ (deg)

(a) Δa = −3 dB (linear: 0.7)

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dB

|AFθ||AFψ |

0 30 90 120 150 18060

Angle θ (deg)

(b) Δa = −1.93 dB (linear: 0.8)

−25

−20

−15

−10

−5

0

5

Arr

ayfa

ctor|A

F|in

dB

|AFθ||AFψ |

0 30 90 120 150 18060

Angle θ (deg)

(c) Δa = −0.91 dB (linear: 0.9)

−6 −5 −4 −3 −2 −1 0

Amplitude balance Δa in dB

−25

−20

−15

−10

−5

0

5A

rray

fact

or|A

F|in

dB

|AFθ||AFψ |

(d) |AF(θ = 90◦)| in Abhngigkeit von Δa

Figure 9: Dependency of the array factor on the amplitude balance Δa and the angle θ.

of the array factors. The maximal absolute value of the lobesand minimal values of the local minima remain unaffected.At the same time, the patterns are deformed. With increasingphase deviation, grating lobes occur in |AFθ| in a regionaround 180◦ or 0◦, depending on the leading sign of thephase deviation. However, the ability to suppress cross-po-larization remains nearly unaffected and acts in the same in-tensity but at a different angle θ.

If the performance is evaluated in the main beam direc-tion (θ = 90◦), a dependency of the array factors |AFθ,ψ| onthe phase balance Δφ can be observed. The dependency isshown in Figure 10(d). If one allows a maximum degradationof suppression of the cross-polarization |AFψ| to the level of

−20 dB, the phase imbalance has to be smaller than 8◦. Forthe sake of completeness, it should be noted that a commonreason for phase deviations between the feeding, signals isdifferent lengths of the feeding lines. In this case, the phasedeviation remains not constant over the frequency butdepends linearly on the frequency and the length differenceΔl of the feeding lines

Δφ(f) = Δl

λr2π = Δl · f

cr2π, (15)

where λr is the wave length in the feeding line and cr thepropagation velocity.


−25

−20

−15

−10

−5

0

5

Arr

ayfa

ctor|A

F|in

dB

|AFθ||AFψ |

0 30 90 120 150 18060

Angle θ (deg)

(a) Δφ = 30◦

−25

−20

−15

−10

−5

0

5

Arr

ayfa

ctor|A

F|in

dB

|AFθ||AFψ |

0 30 90 120 150 18060

Angle θ (deg)

(b) Δφ = 20◦

−25

−20

−15

−10

−5

0

5

Arr

ayfa

ctor|A

F|in

dB

|AFθ||AFψ |

0 30 90 120 150 18060

Angle θ (deg)

(c) Δφ = 10◦

−25

−20

−15

−10

−5

0

5A

rray

fact

or|A

F|in

dB

|AFθ||AFψ |

Phase balance Δφ in deg

0 2 4 6 8 10 12 14 16 18 20

(d) |AF| depending on Δφ

Figure 10: Dependency of the array factor on the phase balance Δφ and the angle θ.

Another issue influencing the performance of the con-cept is coupling between the array elements. This has beenleft out of the scope of this paper due to generally lowcoupling in the presented prototype for verification. Thecoupling degrades the gain of the whole array but has rathersmall effect on the polarization suppression, as long as thestructure and current distribution in the antenna are keptsymmetrical in the desired way. The coupling level, whichmight be neglected for the proper functionality of the con-cept, is low enough for most of the planar antennas placednext to each other and possessing the radiation beamperpendicular to the antenna surface. Such antennas areprerequisite for the described applications, nevertheless thedesigner has to keep the effect of the coupling in mind.

7. Conclusions

In this paper, a concept for the realization of fully polarizedultrawideband radiators is introduced. The principle of op-eration is explained, and a mathematical model for thedesign of the array is presented. The performance of the con-figuration regarding the cross-polarization suppression isinvestigated. The correctness of the considerations has beenverified by the measurement of single element and dual-polarized array. The advantage of this configuration is a highpolarization purity over a wide angular and frequency range,a very good decoupling between the ports for orthogonalpolarizations and a fixed phase center for both polarizationsover the entire desired frequency range. The presented


principle can be applied to all antennas with a main beamdirection orthogonal to the substrate surface. The influenceof the amplitude and phase balance on the polarization pu-rity is investigated, and the corresponding dependencies arepresented.

The solution can be exploited in the design of compactantennas, as presented in [14, 15]. In these publications anadditional advantage is reached, namely, a similarity of thebeams in the E- and H-plane. The described solution allowsfor the realization of radiators for high-end radar/imagingsystems or high data rate communication devices based ona fusion of MIMO and UWB.

Acknowledgment

The authors would like to thank the German Research Foun-dation (DFG) for the financial support.

References

[1] Federal Communications Commission (FCC), “Revision ofpart 15 of the commissions rules regarding ultra-widebandtransmission systems,” First Report and Order, ET Docket 98-153, FCC 02-48, 2002.

[2] R. Zetik, J. Sachs, and R. S. Thoma, “UWB short-range radarsensing,” IEEE Instrumentation and Measurement Magazine,vol. 10, no. 2, pp. 39–45, 2007.

[3] H. Mott, Polarization in Antennas and Radar, Wiley-Intersci-ence, New York, NY, USA, 1986.

[4] J.-S. Lee and E. Pottier, “Polarimetric radar imaging: frombasics to applications,” in Optical Science and Engineering, vol.142, CRC Press, Boca Raton, Fla, USA, 2009.

[5] D. Giuli, “Polarization Diversity in Radars,” Proceedings of theIEEE, vol. 74, no. 2, pp. 245–269, 1986.

[6] ETS-Lindgren, 2010, http://www.ets-lindgren.com.[7] J. Y. Chung and C. C. Chen, “Two-layer dielectric rod anten-

na,” IEEE Transactions on Antennas and Propagation, vol. 56,no. 6, pp. 1541–1547, 2008.

[8] G. Adamiuk, T. Zwick, and W. Wiesbeck, “Dual-orthogonalpolarized vivaldi antenna for ultra wideband applications,” inProceedings of the 17th International Conference on Microwaves,Radar and Wireless Communications, vol. 2, pp. 282–285,Wroclaw, Poland, May 2008.

[9] W. Sorgel and W. Wiesbeck, “Influence of the antennas on theultra-wideband transmission,” EURASIP Journal on AppliedSignal Processing, vol. 2005, no. 3, pp. 296–305, 2005.

[10] G. Adamiuk, L. Zwirello, L. Reichardt, and T. Zwick, “UWBcross-polarization discrimination with differentially fed, mir-rorred antenna elements,” in Proceedings of the IEEE Interna-tional Conference on Ultra-Wideband, (ICUWB ’10), Nanjing,China, September 2010.

[11] P. S. Hall, “Dual polarization antenna arrays with sequentiallyrotated feeding,” IEE Proceedings Microwaves, Antennas andPropagation, vol. 139, no. 5, pp. 465–471, 1992.

[12] K. Woelders and J. Granholm, “Cross-polarization and side-lobe suppression in dual linear polarization antenna arrays,”IEEE Transactions on Antennas and Propagation, vol. 45, no.12, pp. 1727–1740, 1997.

[13] J.-Y. Tham, B. L. Ooi, and M. S. Leong, “Diamond-shapedbroadband slot antenna,” in Proceedings of the IEEE Interna-tional Workshop on Antenna Technology: Small Antennas andNovel Metamaterials, (IWAT ’05), pp. 431–434, March 2005.

[14] G. Adamiuk, W. Wiesbeck, and T. Zwick, “Differential feedingas a concept for the realization of broadband dual-polarizedantennas with very high polarization purity,” in Proceedings ofthe IEEE Antennas and Propagation Society International Sym-posium, (APSURSI ’09), pp. 1–4, Charleston, SC, USA, June2009.

[15] G. Adamiuk, S. Beer, W. Wiesbeck, and T. Zwick, “Dual-orthogonal polarized antenna for UWB-IR technology,” IEEEAntennas and Wireless Propagation Letters, vol. 8, Article ID5200399, pp. 981–984, 2009.

[16] G. Adamiuk, L. Zwirello, S. Beer, and T. Zwick, “Omnidi-rectional, dual-orthogonal, linearly polarized UWB Antenna,”in Proceedings of the European Microwave Conference, (EuMC’10), pp. 854–857, Paris, France, September 2010.

[17] G. Adamiuk, J. Timmermann, W. Wiesbeck, and T. Zwick, “Anovel concept of a dual-orthogonal polarized ultra widebandantenna for medical applications,” in Proceedings of the 3rdEuropean Conference on Antennas and Propagation, (EuCAP’09), pp. 1860–1863, Berlin, Germany, 2009.

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