ON-WAFER MEASUREMENTSWITH THE

HP 8510 NETWORK ANALYZERAND

CASCADE MICROTECH WAFER PROBES

Eric StridPresident and C.E.O.

Cascade Microtech, Inc.P.O. Box 2015

Beaverton, Oregon 97075

RF & MicrowaveMeasurementSymposiumandExhibition

Flin- HEWLETT~~ PACKARD

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Hewlett-Packard is pleased to have the opportunity to present this paper,written by Eric Strid of Cascade Microtech, Inc., at the RF & MicrowaveMeasurement Symposium and Exhibition.

Abstract:

This paper presents a system, consisting of the HP 8510 network analyzerand Cascade Microtech wafer probes, that is used to make RF measurementsof microwave devices and IC's (MMIC's) directly on-wafer. Theconfiguration of the system and the characteristics of the wafer probes aredescribed. Techniques for making the best possible on-wafer measurementswill also be addressed, and a range of on-wafer measurement applicationswill be examined.

Biography

Eric Strid received his BSEE degree at MIT in 1974 and MSEE degree fromUC Berkeley in 1975. He first worked on microwave MIC's at FarinonTransmission Systems, San Carlos, CA. In 1979, he joined the galliumarsenide research group at Tektronix, which has recently evolved intoTriQuint Semiconductor. In 1983, he co-founded Cascade Microtech, Inc.,where he is now President and CEO. Eric has published various papers onpower GaAs FET's, noise figure measurements, analog and digital GaAsIC's, and high-frequency wafer probing.

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In this talk, we'll be covering the basics andsome details of how to use the CascadeMicrotech Wafer Probe System with HP 8510Vector Network Analyzer.

ON-WAFER MEASUREMENTSWITH THE HP 8510 NETWORK ANALYZER

AND CASCADE MICROTECH WAFER PROBES

5315

OUTLINEI. Introduction

II. The Wafer Probe System

III. Measurement Considerations

IV. Typical Measurements and Applications

5316

WHY PROBE AT MICROWAVE FREQUENCIES?• Immediate Measurements on Just Completed Wafers

Without Dice and Bond

• Wafer Mapping of RF Parameters

• Practical Means to Gather a Statistical,ly SignificantQuantity of RF Data

• Further Wafer Processing After Microwave Measuremen

• Automated, Active, On-Wafer Trimming

5317

3

We'll start with an introduction of theproblem of wafer probing at very highfrequencies and a discussion of therequirements of a probe for microwavefrequencies. Then we'll talk about themechanical and electrical aspects of theCascade probe system and how it iscalibrated at the probe tips. We'll thenmove to error corrected measurements usinga Vector Network Analyzer and some of theerror considerations therein. Finally, sometypical measurements and applications willbe discussed.

There are a variety of uses for wafer probesat microwave frequencies. One of theobvious applications is that you can getimmediate measurements on wafers thathave just been completed without having tothin them, dice them out, and bond theminto fixtures. It also keeps the positionalinformation between the die intact so thatyou can do a wafer map of RF parametersand use this data to work on process yieldimprovement. Because you can automatedata-taking with an autoprober, it's a wayto gather a significant quantity of RF data.Finally, it's a way to do in-processmicrowave measurements. For example, aFET can be measured as soon as its gatemetal is down, before doing passive elementlithography on the wafer. We haven't seenany practical wafer trimming approachesyet, but the probes can provide a means ofdoing the RF measurements if you had anautomated trimming solution.

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TYPICAL RF PROBE USAGE

IN IC DEVELOPMENT

Prototype Testing• Manual RF Probe

Test Development• Full RF Autoprobe

Manufacturing Test• Sample Monitor Structures at RF• Sample RF Testing of Component• Full RF Testing/Binning of Component

TYPICAL RF PROBE USAGE IN HIGH SPEEDDEVICE PROCESS DEVELOPMENT

Research• Quick Turn Around on Test of Transistors

and Demonstration Devices

Development• Element Characterization• Statistical Data Base for the Process• Establish Design Limits

Manufacturing• Process Monitoring

PROBLEMS WITH CONVENTIONAL PROBING TECHNIQUESAT MICROWAVE FREQUENCIES

TestInput-

Bonding Pod(CircuitConnection)

'COleUnder Test

5318

5319

5320

4

When a high speed IC or MMIC is underdevelopment, the designer can get feedbackas soon as prototype wafers come out byusing RF testing at the wafer level. Thiscan be useful when there are several designson the wafer and some designers want itdiced up for RF measurements and otherdesigners don't. When the IC is in the testdevelopment phase, there is often a lot ofdata taken with an autoprober and a verycomplete RF test plan. The key parametersthat really sort between good and bad partscan be figured out, for purposes ofstreamlining the manufacturing test. You'lleither know that if the test FETS are goodat RF, the die will be good, or maybe you'llwant to sample some of the components forRF testing then do a DC sort for that wafer.Or perhaps you should just do a full RF testof all parts, possibly binning components aswell.

In high speed device process development, aquick-turnaround capability to test partsimmediately after wafers come out of thefabrication lab can provide feedback at allphases. In the research environment, moreexperiments can be performed per unit time.As the process gets more mature, designersare interested in passive and active elementcharacterization and then developing a largedata base of the averages and standarddeviations of the element's parameters andto develop design limits for the parameters.In manufacturing, the typical useage is tomonitor active elements for their RFcharacteristics.

At low frequencies, the basic approach towafer probing is to use carefully alignedsharp needles to touch each pad on the ICunder test. These present some basicproblems at microwave frequencies becausethere is significant self-inductance, mutual­inductance, and mutual-capacitancesbetween the needles and between theneedles and the wafer itself. Result is that,electromagnetically, the fields are alltalking to each other from probe to probe athigh frequencies. If you were to look at aneedle probe on a network analyzer, itwould typically have a terrible return loss,look like an antenna, and have non­repeatable S-parameters at microwavefrequencies.

iIo...- _

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PROBE EQUIVALENT CIRCUIT

Zo Z1Input ( ~

9Zu#1

Zg12,....

4-

QZL2Zo Z2Input ( ~

#2

5321

..

This is a simple first order equivalentcircuit for a few probes. We assume thatthere is a 50 ohm line connecting to two ofthese probes and a common groundconnection which is connected through aground lead ZG12. There is someinductance Zl ana Z2 in the signal probesthemselves before contacting the device on

5

the wafer (ZLI and ZL2 here). We show thefirst order parasitics as inductors becauseusually the first problem we run into isinductance of the probes instead of thecapacitances. Usually the common leadinductances, ZG12 in this case, are moreproblematic than self-inductances Z 1 andZ2.

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CROSSTALK DUE TO COMMON LEAD INDUCTANCE

-10r----:-------------:;------------,

(All Contacts Shorted)

-40 l--...l---l----I.._l--...l---l---L.-I_..L--L---L.-Il--..L--L---L.-Il--..L--L----J

1 3 5 7 9 11 13 15 17 19

FREQUENCY (GHz)

This is the calculated worst-case crosstalk from the circuit of the previousslide wherein all the probe tips are shorted together. A common leadinductance of only 50 pH provides about 12 dB of isolation at 20 GHz.Since a typical bond wire has several hundred pH, this is a very significantproblem when compared to small needle probes.

6

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5322

ELECTRICAL REQUIREMENTSFOR MICROWAVE PROBES

• Good Return Loss (Better Than 10dB)

• Low Insertion Loss «0.2dB)

• Repeatable Probe S-parameters«0.01 Vector on Smith Chart),Implies Low Radiation

THE WAFER PROBE SYSTEM• Mechanical Aspects

• The Probe Head

• Measurement System and Calibration

The requirements for a microwave waferprobe are first that it have reasonably lowreflections. If there is a return loss ofbetter than 10 dB, typically a correctedvector network analyzer can then readthrough these reflections and get prettyclean data. Secondly, it should havereasonably low insertion loss so that thenetwork analyzer doesn't have to correctthrough a large loss between it and theprobe tips. Thirdly, the probe's S­Parameters should be very repeatable. Whenyou do an error correction on the 8510, youare assuming that the S-parameters betweenthe analyzer and the device under test arestable. We measure the repeatability bylooking at the vector difference on a Smith

5323 Chart which would be typically less than.01.

Next, we'll introduce the Cascade Microtechprobe system. We'll first talk about somemechanical aspects and how the probe headsare used in contacting the wafer, and then abasic approach in calibrating the NetworkAnalyzer for measurements.

CASCADE MODEL 42 PROBE STATION

5324

5325

7

Here is a photo of a Model 42 manual probestation. Its capabilities are a superset of atypical DC probe station in that DC probeneedles can be mounted on the top plate,but RF probe heads can also be added on upto four sides of the device under test. Allof the probes can then be raised andlowered in mass and the wafer movedunderneath them manually. A key featureof this probe station is that the probe headsare held very rigidly in place so that cableforces from coaxial cables connected to theprobe heads do not move the probe headssignificantly with respect to the wafer.Available, but not shown here, are similarmounts for these probe heads for a Rucker& Kolls 1032 Autoprober or Electroglass1034 or 2001 Autoprober. Also available areprobe cards which mount in standard 4 1/2"wide probe card mounts on any manual orautoprobe station.

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MICROWAVE PROBE HEAD

WAFER PROBE (TOP VIEW)

This is a photo of a single line microwaveprobe head. This head can be considered tobe an adapter from coax to device bondpads. This shows the coaxial connector andthe probe tip where contact to the chip ordevice on wafer is made. This head ismounted on the probe station with screwsthrough the back three holes.

5326

This diagram is approximately what you seethrough the microscope when two probes arecontacting an FET. In this, and the nexttwo slides, each probe head has a ground­signal-ground contact configuration. In thisview, the contacts are visible just beyondthe edge of the body of the probe tip. Eachprobe contacts the source connection oneither end of the FET with its grounds.The gate contacted on one side and thedrain on the other side with 50 ohm signallines from each probe.

WAFER PROBE (SIDE VIEW)

Probe

3.2 Grams Per Mil

Probe

5327

The probe heads are built from 10 milalumina and skate when contacting so thatthey wipe the contact pads on the deviceunder test. The ends of the probe tips arecarefully beveled in approximately thisshape. The probe heads approach the deviceunder test at about a 1:5 slope so that if theprobe is vertically overtravelled 5 mils, itwill skate across the pad about 1 milhorizontall y.

Semiconductor Wafer

5328

8

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WAFER PROBE (BOTTOM VIEW)

ProbeContacts

This shows a view if the probes were turnedover or as if we're looking at a probe tiparea from the die. We see the basicallycoplanar waveguide approaching the probetip and the probe contact metal plated ontothe coplanar waveguide metal at the probetips. Center-to-center spacings of thecontacts are fixed, and are available from 2mils to greater-than 10 mils.

ProbeContacts

5329

A typical setup for network analyzermeasurements at the probe tips simply hooksthe 8514 or 8515 test set through a goodcable to the probe heads. Cables need to below reflection and good repeatability.Typically, .141" semi-rigid coaxial cable isappropriate since the probe heads moverelatively little in any test. The bias forFET's or other active devices can besupplied through the bias tees in the test set.Normal connections between the 8510 andsweeper or synthesizer are used.

.141 semi-rigidcable

MEASUREMENT SYSTEM BLOCK DIAGRAM

5330

9

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BASIC ON-WAFER CALIBRATION

Probe

,.".

-' .~ 100Q

~ 100Q

Open Circuit(In Air)

SOQ Termination

Probe

ShortCircuit

SOQThrough Connection

Dc:::::J

D

: .-._"

5331

This is a basic reflection calibration for aground-signal-ground probe head. The opencircuit standard is probe lifted in air. Theshort circuit standard shorts the threecontacts on a small bar of conductor,usually a gold metalization. The 50-ohmload for a ground-signal-ground probe usesa pair of trimmed 100 ohm resistors that areabout 2 mils long making a minimum sized50-ohm load.

The typical through calibration uses a veryshort 50-ohm coplanar wave-guide throughconnection, shown with and without probescontacting it in this figure. The length ofthe through line (about 6 milsjps) is enteredinto the cal kit for the 8510.

10

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,IMPEDANCE STANDARD SUBSTRATE (ISS)

MEASUREMENT CONSIDERATIONS• Effects of Incorrect Calibration Standards Definitions

• Understanding "Planar" Impedance Standards

• Verification Techniques

• Other Sources of Error

A standard accessory supplied with Cascadeprobe stations and probe cards is animpedance standard substrate, ISS, which isused for calibrating and verifyingcalibrations at the probe tips. Each row inthe pattern corresponds to a probe contactconfiguration. For example, the top row isfor signal-ground probes. On the third roware a ground-signal-ground calibration andso forth. Each column refers to a differenttype of standard. On the first column areinductors for various probe types, thencapacitors, various resistor values, then theshort 50-ohm and through calibrationstandards, then verification patterns fortwo-port measurements including 10 and 20dB pads and other miscellaneous patterns,

5332 especially transmission lines.

We will now explore more deeply some ofthe accuracy considerations in thecalibration at the probe tips by first lookingat what are the effects of havingincorrectly specified calibration standards,then relating these to the knowns andunknowns of planar impedance standards.Then we will discuss verification techniqueswe've found to be useful and finally othersources of errors.

5333

•

ERROR CORRECTION BASICS

fm •=tOt =m fa

fm1 • :Q] f1

:Q]Known

fm2 • f2 Standards

fma • :Q] fa

The basic approach when making a one portcalibration on an ANA is to assume that allthe errors from the analyzer or from thecables or fixturing are described by an erroradaptor. This error adaptor's parameterscan be solved by placing three knownimpedance standards, Gamma 1, Gamma 2,and Gamma 3 at the calibration plane. Foreach of these, a reflection coefficient ismeasured, Gamma Ml, Gamma M2, andGamma M3 by the network analyzer. Afterthe error adaptor parameters have beencalculated, then the actual gammas for adevice under test can be calculated from theerror adaptor's parameters and the measuredreflection coefficient Gamma M.

5334

11

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EFFECT OF INCORRECT

CALIBRATION STANDARDS

r, = 1 Open

r2 = -1 Short

r3 = 0 Load

r _r, r2 2rm + r2 r3 (1 - rm) + r, r3 (-1 - rm)

a - r, (rm - 1) + r2 (rm + 1) + r3 (-2rm)

5335

EFFECT OF INDUCTANCE IN LOAD STANDARD

OPEN-I +J0SHORT--! + j0TERH-0+J. 025

5336

EFFECT OF INDUCTANCE IN SHORT CIRCUIT STANDARD

OPEN-l +JBSHORT--l+ J-. 05TERH-0+J0

5337

12

Instead of simply solving for the erroradaptor's S-parameters with three knownstandards, we keep all the unknowns in theequation, i.e. keep the three known standardimpedances, Gamma 1, Gamma 2, andGamma 3 as unknowns because we don'tcompletely know about them. Then wepostulate that we have cascaded the erroradaptor with an arbitrary adaptor whichhas resulted in a condition of Gamma M1 (aperfect open), Gamma M2 is a perfect short,and Gamma M3 is a perfect load. Gammaactual is then given by this equation. Inother words, this equation shows howimperfect standards will cause measuredreflection coefficients to be incorrectlymapped by the error adaptor.

In the follo·wing slides, the solid lines showa normal Smith Chart, and the dotted linesdisplay a distorted Smith Chart, which isthe locus of points which would bemeasured having incorrectly specified someparameter of the calibration standards.(Think of each chart as being at a singlefrequency). This figure shows the effect ofhaving a perfect open and short impedancewith an inductance in the termination. Aswe see at the top of the Smith Chart, thedistortion is accentuated so that a measuredhigh-Q induction would be outside of theSmith Chart if it were on the edge of + j 50ohms. At the bottom of the Smith Chart, aminus j 50 ohm impedance would bemeasured to have several ohms real.

This chart shows the effect of having thecorrect open calibration standard, acorrection load standard, and the shortstandard having an incorrect offset. In thiscase, again, a high Q inductor would gooutside the Smith Chart and a high Qcapacitor would appear to be too resistive.

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•

t

r

EFFECT OF INCORRECT CoFOR OPEN CIRCUIT STANDARD

OPEN-l+j.05SHORT--)+ JBTERH-e+j0

EFFECT OF OFFSETTING BOTHSHORT AND OPEN CIRCUIT STANDARDS

OPEN-I +j. 05SHORT--l+ J-. 05TERM-B+ jB

Effectively Moves the Reference Plane

This chart now refers to the short beingcorrect, the load being correct, and the openhaving an incorrect offset. This causes adistortion on the top and bottom of thechart in the same fashion as the short offseton the previous slide.

5338

This figure shows the effect of having anoffset on the short and an equal offset inthe open. The termina tion is assumed to beperfect. In this case, the offsets in the openand short, as we might expect, cause arotation of the entire chart corresponding toa movement of the reference plane. Notethat this analysis is applicable to coaxial orother transmission media as well.

5339

13

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SHORT CIRCUIT STANDARD REFERENCE PLANEVS. OVERLAP DISTANCE

Probe

,,,,,,,

r- Overlap Distance

-2.,.

I.,.

-I .,.

o

dletanee to tlr.' edge 0' .hort (lnlla)

5340

One of the effects we've seen on the planarshort standard is that the reference plane isnot set by the physical planar contact to theshort pattern, but effectively by how muchthe probe overlaps the edge of the shortunderneath the probe tip. This graph showsthe relative delay versus the overlapdistance of the probe head over the shortmetal. We recommend normally that theshort standard be contacted with about a 1mil overlap. This is done by centering theprobes on a 2 mil wide short bar. This 1mil overlap is somewhat arbitrarily chosenand could be moved to some other overlapif the offset for the open circuit, for

example, was also moved. If the probe isjust contacting the short on the edge of themetal, the actual edge of the gold mushesout on the edge of the short circuit so thereis a relatively large scatter at zero overlap.Note that it is possible to make repeatablemeasurements within a fracture of a ps ifthe probe is carefully placed over the short­circuit. As the probe overlaps the shortmore and more, an assymptote of about 2 psof effective shortening is reached. This iswhere the probe is overlapping the short by25 mils or more. There is a similar but lesssensitive effect of the reactance of the 50ohm load standard as function of where itis contacted by the probe.

14

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"--- - ---~~

ONE-PORT VERIFICATION WITH RESISTORS511 ZREF 1.0 Un its

200.0 mUnits/

hp

50Q50Q

25Q 12.5Q(2) (1)

200Q 100Q(3)

Probe

START 0.130000000 GHzSTOP 26.500000000 GHz

5344

For each probe configuration, there areseveral resistor values available on the ISSwhich can be used for 1 port verificationstandards. For example, the ground-signal­ground probe is contacting a 200 ohmresistor, a 100 ohm resistor, a 25 ohmresistor and a 12.5 ohm resistor. The results

of such measurements are shown in theSmith Chart wherein marker 1 shows the12.5 ohm resistor measurement, 2 shows the25 ohm resistor, and 3 shows the 100 ohmresistor. The measured reactance of each ofthese is a slight function of where the probecontacts the actual metalization on theresistor ends, as with the short standard.

15

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TWO-PORT VERIFICATION WITH ATTENUATORS

521 log MAGREF -10.0 dB

1. 0 dB/

Probe Probehp

\ .----- I -~

"

~['J:. ,

.. ..: ."

b/, . , '

- '/

...c. -~

~1

10dB Attenuator

D

START 0.130000000 GHzSTOP 26.500000000 GHz

This is an example of a 2-port verification standard wherein a 10 dB pi­section pad is used to make a 50 ohm load on each port and about a 10 dBattenuation from the port 1 probe to the port 2 probe. A log-magnitude plotof S21 is shown to be very close to 10 dB of attenuation through 26 GHz forthis verification standard.

16

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5345

ONE-PORT VERIFICATION USING OPEN STUB

511 ZREF 1. 0 Un it.

200.0 mUnit./

hp

CA

START 0.131800000 GHzSTOP 26.500000000 GHz

5346

The best verification standard we havefound for I-ports (especially with ground­signal-ground probes) is a small coplanarwaveguide open stub. It is shown with andwithout a probe contacting it in this figure,with the Smith Chart of the resultingcorrectly calibrated reflection coefficient ofthe open stub. Since the lines in thecoplanar waveguide are only a couple milswide, the loss is relatively high. In thisexample, the length is about 120 mils or 3millimeters. The reflection coefficient ofthe stub starts out at an open, goes through

a short at about 10 GHz, goes back near anopen at 20 GHz, then stops at 26 GHz. Wewould expect that the magnitude Sll woulddecrease monotonically with frequencybecause the losses are increasing withfrequency. This is a sensitive measure ofwhether the reactances of the impedancestandards (the short, open, and load) havebeen correctly chosen for the calibration. Ifthey were incorrect, the reflectioncoefficient of the open stub would be toohigh or low at the top or the bottom of theSmith Chart as shown in the previousdistorted Smith Chart predictions.

17

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EFFECT OF INDUCTANCE IN LOAD AND OPEN CIRCUIT

OPEN-l+j.lSHORT--l+j0TERH-0+j.025

5347

The magnitude of the measured open stubreflection coefficient gives us a good insightinto whether the magnitudes are beingcorrectly measured. However, this distortedSmith Chart shows an example of anincorrect open offset as well as aninductance of the 50 ohm standard. Thecombined effect gives an accurate measuredreflection coefficient magnitude at the topand bottom of the Smith Charts, but itsphase is incorrect, especially near the open.

18

By minimizing the deviation from linearphase of the measured phase of the openstub, another one of the reactances of thethree impedance standards can bedetermined. Since all three of the standardsof the load, the short, and the open have, tosome extent, an unknown reactance, wehave three unknowns. Two then can beverified by the magnitude and the phase ofthe open stub verification standard, and thethird one effectively just sets the referenceplane of the measurement.

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LEAKAGE BETWEEN PROBES(GROUND-SIGNAL CONFIGURATION)

S12 log MAGREF 0.0 dB1 10.0 dBIV -43 779 dB

0r.----r---,---.......---r---,--..:....:.r--;..;...:-T:.::::...-...----,..--~~ ~

CI-:-~=±-:_==_~-+--~-_+-_lI----I----J.--~--1

A MARKER 1l 0. 0~-97f 1 Gf~z

.-20 .....--+---+---+----4---+--+---I---+--~-_I

START 0.131800000 GHzSTOP 2S.500000000 GHz

5341

Another potential source of errors in two­port measurements is crosstalk from thefix turing, i.e. probes in this case. Since theisolation error term of the normal 12­element model is only a partial isolationcorrection, we do not use it with waferprobes and omit the isolation (effectivelyzeroing out the EX error terms). This plotshows a series of different cases measuredthrough 26 GHz between a ground-signalprobe and a signal-ground probe. The toptrace corresponds to the two probe headsboth contacting short circuits which areseparated by about 10 mils. The next traceshows two 50-ohm terminations which arealso 10 mils apart. The next trace shows the

19

probes 10 mils apart lifted up in air. TraceD is the case where the probes are 30 milsapart in air. Trace E has the probes in 30mils apart, but both on 50 ohm terminations.Trace F is where the probes are 100 milsapart in error. One of the interestingproblems of measuring crosstalk is defininga standard for zero corsstalk. For example,you would expect there to be some mutualinductance in Trace A and mutualcapacitances in the open cases, and mutualinductances and capacitances in theterminated cases. However, you would notexpect to see the structure at 5 GHz from asimple coupled inductance or capacitancecase.

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LEAKAGE BETWEEN PROBES(GROUND-SIGNAL-GROUND CONFIGURATION)

S12 log MAGREF 0.0 dB

10.0 dB/

hp

1

fJ ~.-./

~

~

~...- 11 r Vi\~~0-..~ ~~~- -

~v ""--- ~

~r\/1 ~I fl

~j~((.~ f\J ~ ~ if{',

A \'\f\ ~' ~ n

~~ ~\lIV

~~•• WI

•

11

1

H

*

START 0.130000000 GHzSTOP 26.500000000 GHz

5342

Some similar corsstalk measurements donefor ground-signal-ground probes are shownhere. The top trace shows both probescontacting short circuits but separated byonly 4 mils. The second trace shows bothprobes in air about 8 mils apart, and thethird trace shows probes in air separated by

100 mils. There is about 20 dB bettercrosstalk with the ground-signal-groundprobe pair than with the ground-signal­ground probes from the previous figure.This apparently is due to the extra groundin the ground-signal-ground case effectivelyshielding the field from coupling tosomething else.

20

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Some typical measurements and applicationswe'll show are a diode measurement, adiscrete FET, an MMIC measurement, andthe parameter extraction of a GaAs FET forpurposes of data base development.

TYPICAL MEASUREMENTS AND APPLICATIONS• piode Measurement

• Discrete FET Measurement

• MMIC Measurement

• Parameter Extraction and Database Development

DIODE MEASUREMENT

5348

This figure shows an example of a diodemeasurement. The device under test isactually the gate source of a GaAs MESFETwith the drain open, but it is very similarfor other types of schottky or pn diodes.The measured data for trace 1 is when thereis one volt forward bias on the gate. We seethat the device is basically about 6.8 ohmsin series with an inductance of about 44 pH.It starts at low frequency at the 6.8 ohmintercept and goes inductive at higherfrequency. Bias condition 2 still has someconductance in the diode where at lowfrequencies the resistance is about 100 ohms.The third trace on the Smith Chart shows avolts on the gate, which clearly correspondsto a series resistance and the gate sourcecapacitance. This very clean measurementof S-parameters can result in convincingmodels using lumped equivalent circuits ofthe elements.

\

,;';C - - - - /...\ I .... "

\ ....... -~- .... ~\", ,,\..... / ,,\ \

.,.. ......... " >; - ..... "f )I ........

--::: ......

0.13000""ee GH:I26.500000000 GHz

STARTSTCP

hp

S11 ZREF 1.11' Unit.

200.0 mUni t.1

H

"ICj

Lseri.. Rserles

Simple Diode Model

5349

DISCRETE FET MEASUREMENT

We will demonstrate two examples of FETmeasurements. In each case, we've used thesame device at the same bias condition butprobed it in two different ways. First, we'llshow the device measured with a ground­signal probe on the drain side and a signal­ground probe on the gate side. We'll thenshow a ground-signal-ground on both thegate and the drain of the FET.

5350

21

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DISCRETE FET MEASUREMENT(GROUND-SIGNAL CONFIGURATION)

TEST FETS 11':':21812322

G5-5G V05-3 VG5-~

F~AD1U~:::=5

RRD1U::::=. 1

11/22/85

5351

When measured with the ground-signal and signal-ground probes,the $­parameters match a lumped equivalent circuit up to roughly 12 GHz, butextra parasitics are involved and the match to the simple equivalent circuitbecomes inaccurate. Also, the slight bumpiness of the data at above 20 GHzis probably due to the higher crosstalk of the ground-signal probes.

22

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•

DISCRETE FET MEASUREMENT(GROUND-SIGNAL-GROUND CONFIGURATION)

.............

....

....:.....

..:1;····

/~~..,._._------i-------__..._,'-/" .. ,

/.,/ •.... .. '--..,.,.; 1 nt,-) / ..... '"

/ ...:.:.... '\/ ~

I \!~. \

/....,. .'"""~:~~:;~~~>~.... \.,l ". ~ ;.: ...../ -,

ohms;' :' .II . .' . 'l8l.. . .,. \ohms " ." /......... . . .... # ..:Q :~. \

fl Il.[·.L5=l .•.... ) _·~c ••l-.l

R15

t"FfFfFp5Oh:llS

Gl-i= \GH:: !,\ ....~!:.....'.;.:...

\:.:...... ~c:>:

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..........................4 ·~.;::I............ . ; ~ ,' ...

..-- ----.---....---~---_:--~---

RADIIJS=5RADIU::::=.2

V05-3 VGS-l!I 11/22/B~FETS 115215125:22

TEST

gm 3;'.38Cg~ 412Gild 29C:d~ 99Tt e e.......rd 520R,n 4.20Rs 1. 20Rg .50Ld 23Lg 25Ls 1!I3mt-:! 19.4[;l

gm/C 12.03Ft =:4.;:'5

5352

When measuring the same device at the same bias condition with ground­signal-ground probes from the gate side and the drain side, notice that thereis now a very good fit to a basic lumped equivalent circuit for the FET.The source inductance has been decreased drastically, and the crosstalkfrom one probe to another is reduced by the ground-signal-ground probes.

23

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c

MMIC MEASUREMENTS21 log MAGREF 0.0 dB

2.0 dB/

hp

J "'-. -- ~........... ---..... --

I.....,

STARTSTOP

0.500000000 GHz20.000000000 GHz

5353

Here is an example of measuring a singleMMIC amplifier stage from 0.5 to 20 GHz.The measurements on single stages are oftencleaner than what people have been able tomeasure in their packages. To probeMMIC's using microstrip as the transmissionmedia, it is necessary to bring the ground

24

contact from the back side up to the surfaceat the inputs and outputs for contactingwith the coplanar probes. Usually a littlebit of thought at the layout stage of adevice or circuit can result in much easierand cleaner measurements later when thedevice is to be probed.

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EQUIVALENT CIRCUIT PARAMETER EXTRACTION

EquivalentCircuitModel

Cgd

Lg Au I Cg• ·v

I~Rd Ld

_ c

Rgs Rin Rd. Cit.

I-VcYm R.

Ym-Ome-iwTt L•

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The ability to manually probe andautoprobe FET elements on the wafer andvery accurately extract a lumped equivalentcircuit for the measured data has led manyfoundries to using the averages andstandard deviations of the lumped elementsas design limits for MMIC design instead of

25

the raw S-parameters. Besides decreasingthe volume of data to be specified, singleelement variations across the wafer orvariations due to temperature or variationsdue to bias conditions can be monitored.This type of data base also gives moredirect feedback to the process engineer foryield enhancement information.

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EXAMPLE DATABASE OF RF PARAMETERS

One of the difficulties specific to galliumarsenide design is that there is usuallyenough electron traps in the material thatthe dc parameters of a FET, especially drainconductance, are not at all similar to thedevice's RF parameters. The drainconductance is usually about 10 times ashigh at high frequencies in a GaAs FET.This histogram shows an example of thedrain conductance measured over 1000devices and fitted to an equivalent circuitfrom RF probed S-parameters of thetransistor elements. In this fashion, thedesigner can easily see parameter spreadsexpected by a process.

dlotwl'Ill 12 OUTI

VE 4llL 745DE.. 31.1214

55••••...•••...,..

- . - .

·I•• L2

••,•,

/•1

1\· nnl•2 ,•• 111111tH..:n :,~ ll:H1•

llXIIA-Sdote. Id./G-"F/rd/ll:ll'IIoP"I-'ll va•• l.S Vg•• ' _,..11

y RANGE 13 X UNITS/CELL ,

,.••••n••51

••···,,,,22211I

'd

5355

SUMMARY

5356

In summary, we've seen that thecombination of an 8510 network analyzerand Cascade wafer probes can provide avery beneficial tool for device and ICdevelopers at prototyping, development, andmanufacturing phases. We've looked atdetails of the calibration standards andways to verify that the calibration iscorrect. We've seen that it makes adifference how devices under test arecontacted, and it can save a great deal oftesting and modeling effort later if thedevice is simply laid out optimally fortestability. Finally, the capability togenerate a large data base of RF parametersis revolutionizing both IC design formicrowave frequencies and yieldenhancement techniques for MMIC andhigh-speed processes.

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r/i~ HEWLETTa:~ PACKARD

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October 1, 1986

Printed in U.s.A.5954-8351