High Power-PP Two Inductors

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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 2, MARCH 2008 813

A ZVS Bidirectional DCDC Converter WithPhase-Shift Plus PWM Control Scheme

Huafeng Xiao and Shaojun Xie, Member, IEEE

AbstractThe current-voltage-fed bidirectional dcdc con-verter, which refers to a current-fed inverter at low voltage sideand a voltage-fed inverter at high voltage side, can realize zerovoltage switching (ZVS) for the switches with the use of phase-shift(PS) technology. However, the current-fed switches suffer fromhigh voltage spike and high circulating conduction loss. In orderto solve these problems, a novel phase-shift plus pulsewidthmodulation (PSP) control ZVS bidirectional dcdc converter isproposed in this paper. By adopting active clamping branch andPSP technology, the converter can realize ZVS for all switches in awide range of load variation while input or output voltage varies.In addition, a novel control strategy with one port voltage regu-lation and another port current regulation is proposed to make

energy bidirectional conversion freely. The operation principle isanalyzed and verified by a 28V/270V conversion prototype ratedat 1.5kW.

Index TermsActive clamping, bidirectional dcdc converter,phase-shift plus pulsewidth modulation (PSP), pulsewidth modu-lation (PWM), zero voltage switching (ZVS).

NOMENCLATURE

Voltage of the converter port.

Voltage of the converter port.

Current of the converter port.

Inductance in the converter side.

Instantaneous current through inductance , and

, respectively.Total effective inductance in series with the winding

of transformer.Instantaneous current through inductance .

Clamping capacitor.

Instantaneous voltage across clamping capacitor

c.Number of turns of primary winding, and secondary

winding, respectively.Capacitor in the converter side.

Phase-shift angle between and .

Duty cycle of the switches and .

Angular frequency.

s Switching frequency.

Conversion efficiency.

Manuscript received March 1, 2007; revised July 26, 2007. Recommendedfor publication by Associate Editor C. Canesin.

The authors are with the College of Automation Engineering, Nanjing Uni-versity of Aeronautics and Astronautics, Nanjing 210016, China (e-mails: [email protected], or [email protected]; [email protected]).

Digital Object Identifier 10.1109/TPEL.2007.915188

Output power.

Effective value of the current .

Instantaneous voltage across the primary winding

of transformer.Instantaneous voltage across the secondary winding

of transformer.Output of the phase-shift angle controller.

Output of the duty cycle controller.

I. INTRODUCTION

IN recent years, the development of high power isolated bidi-

rectional dcdc converters (BDC) has become an impor-

tant topic because of the requirements of electric vehicle, unin-

terruptible power supply (UPS), distributed generation, energy

storage, and aviation power system [1][12]. In a typical UPS

system, the battery is charged when the main power source is

normal and discharges to supply power in case of the failure of

lose of the main power source. In the aircraft high voltage di-

rect current (HVDC) power supply system [6], when the 270 V

HVDC generator is in gear, it charges the 28 V battery and sup-

plies the 28 V key load by the BDC,and when the generator isin

failure, the 28 V battery discharges to supply 270 V key load bythe BDC. The high-low voltage conversion and electrical isola-

tion are necessary in the above-mentioned conditions. The cur-

rent-voltage-fed BDC is suitable for such system due to its high

voltage conversion ratio and low current ripple in the current-fed

port.

A dual active full bridge dcdc converter was proposed

for the high power BDC in [9] and [10], which employs two

voltage-fed inverters to drive each side of a transformer. Its sym-

metric structure enables the bidirectional power flow and ZVS

for all switches. A dual active half bridge current-voltage-fed

soft-switching bidirectional dcdc converter was proposed with

reduced power components [11]. However, the current stresses

in switches and are asymmetric. When the voltageamplitude of two sides of the transformer is not matched, the

current stress and circulating conduction loss become higher

in [9], [10], and [11]. In addition, these converters can not

achieve ZVS in a wide range of load variation while input

or output voltage varies. These disadvantages make them not

suitable for large variation of input or output voltage condition.

An asymmetric bidirectional dcdc converter with PWM plus

Phase shift (PPS) control was proposed in [12]. The circulating

conduction loss is reduced, however, it results the asymmetric

stresses in the main switches and a bias of the magnetizing

current which decreases the utilization of the transformer. So,

it is not suitable for high power bidirectional conversion.

0885-8993/$25.00 2008 IEEE


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814 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 2, MARCH 2008

Fig. 1. Novel PSP ZVS BDC (a) Main circuit. (b) Key waveforms in Boost mode. (c) Key waveforms in Buck mode.

It is proposed that a current-voltage-fed PSP ZVS BDC based

on a current-fed half bridge and a voltage-fed half bridge guar-

anteeing volt-second balance of the transformer by its capacitors

and in this paper, as shown in Fig. 1(a). The converter

utilizes an active clamping branch and to avoid the

voltage spike, achieve ZVS of and , and also restrain the

start-inrush current [13]. By PWM control of and , the

amplitude of and is well matched while input or output

voltage varies, which can reduce circulating conduction loss,

and realizes ZVS in a wide range of load variation. The control

strategy of Phase-shift (PS) plus PWM is realized by two indi-

vidual controllers. The operation principle of PSP ZVS BDC is

analyzed in detail. A 2232 V/270 V 1.5 kW prototype is builtto verify the operation principle of the proposed converter.

II. OPERATION PRINCIPLE

The BDC has two operation modes. It is defined as

Boost mode when energy flowing from side to side,

and the counterpart is defined as Buck mode. Before anal-

ysis, the following assumptions are given: 1) All the active

power devices are ideal switches with parallel body diodes

( and ) and parasitic capacitors

( , and ); 2) The inductance and

are large enough to be treated as two current sources with

value of ; 3) The transformer is an ideal one with series

leakage inductor . Fig. 1(b) shows the key waveforms in

Boost mode. One complete switching cycle can be divided intotwelve stages. Because of the similarity, only a half switching


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XIAO AND XIE: ZVS BDC WITH PHASE-SHIFT PLUS PWM CONTROL SCHEME 815

Fig. 2. Equivalent circuits in Boost mode for a half switching period (a) Stage 0 [before

] (b) Stage 1[

] (c) Stage 2[

] (d) Stage 3[

] (e) Stage

4[ ] (f) Stage 5[ ] (g) Stage 6[ ].

cycle is described in detail. The equivalent circuits are shown

in Fig. 2. As the two sides of the topology are symmetrical, the

operation principles in Buck mode are similar to those in Boost

mode. Fig. 1(c) shows the key waveforms in Buck mode.

1) Stage 0 [Before ]: Refer to Fig. 2(a). and are

conducting. At this stage, . The power flows

from side to side.

2) Stage 1 : Refer to Fig. 2(b). At is turned off.

and begin to resonate, is discharged and

is charged.

3) Stage 2 : Refer to Fig. 2(c). At , the voltage

across attempts to overshoot the negative rail.is therefore forward biased. During this pe-

riod, can be turned on under zero voltage. The

voltage across is clamped at . At this stage,

.

4) Stage 3 : Refer to Fig. 2(d). At is

turned off. and begin to resonate,

is charged, is discharged. At this stage,

.

5) Stage 4 : Refer to Fig. 2(e). At , the voltage across

attempts to overshoot the negativerail. is therefore

forward biased. During this period, can be turned on

under zero voltage. The voltage across is clamped at. The current of rises to a positive value.


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6) Stage 5 : Refer to Fig. 2(f). At is turned off.

and begin to resonate, is discharged and

is charged.

7) Stage 6 : Refer to Fig. 2(g). At , the voltage across

attempts to overshoot the negative rail. is therefore

forward biased. During this period, can be turned on

under zero voltage. The voltage across is clamped at. At this stage, . The power flows from

side to side. At , the second half cycle starts, which is

similar to the first half cycle.

III. CHARACTERISTICS OF THE NOVEL BDC

A. Output Power

The phase shift angle between

and [referring to Fig. 1(b) and (c)], which is defined to be

positive when is leading to in phase, is used to control

the magnitude and direction of the transmitted power. The duty

cycle of and is used to match the amplitude of and, that means the current keeps constant in stage 0 and

stage 6. Referring to the Appendix A, the duty cycle of and

is given by

(1)

Under PS control, the output power is [10]

(2)

Under PSP control, the output power is (Referring to the Ap-

pendix B)

P =

2 ( N V ) ( 1 0 d ) ( j j + d 0 1 : 5 )

( 2 N ) ! L

; [ 0 ; 0 2 ( 1 0 d ) ]

( N V ) [ + 2 ( 1 0 d ) 0 ( 1 0 d ) ( 2 d 0 1 ) ]

( 2 N ) ! L

; [ 0 2 ( 1 0 d ) ; 0 ]

2 ( N V ) ( 1 0 d ) [ 0 ( d 0 0 : 5 ) ]

( 2 N ) ! L

; [ 0 ; ( 2 d 0 1 ) ]

( N V ) [ 0 + 2 d 0 d ( 2 d 0 1 ) ]

( 2 N ) ! L

; [ ( 2 d 0 1 ) ; ] :

(3)

Fig. 3 shows the relations between the output power (nor-

malized by ) and phase-shift angle under

PS and PSP control. The bold curves are output power versus

under PSP control. The intersection curves are output power

versus under PS control. When the amplitude of and

is matching ( V, V), the both curves are su-perposed under PS control and PSP control. However, the max-

imum of output power under PSP control is higher than that of

PS control in low battery voltage. Evidently, PSP control im-

proves the capability of power transmission.

Fig. 3. Curves of the normalized output power versus the phase-shift angle

( V = 2 2 3 2 V ; V = 2 7 0 V ; N = N = 2 : 1 ) .

B. Circulating Current

When the transmitted power is , the current RMS of

under PS control in Boost mode is (Referring to the Appendix C)

(4)

where is equal to 2 , and is equal to .Under PSP control, the current RMS of in Boost mode is

(Referring to the Appendix D) as (5), shown at the bottom of

the page.

Fig. 4 shows the comparison of the current RMS of under

PS control and PSP control in Boost mode. It is evident that the

circulating current is low under PSP control, which can improve

the conversion efficiency in low battery voltage.

C. Range for Achieving Soft Switching

From Section II, it can be known that in order to achieve ZVS

for all switches, (6) should be satisfied in Boost mode.

(6)

(5)


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Fig. 4. RMS value ofi

. (V = 2 2 3 2 V ; V = 2 7 0 V ; N : N =

2 : 1 ; P = 1 : 5 kW, f = 1 0 0 kHz, L = 1 : 2 H).

Also, (7) should be satisfied in Buck mode.

(7)

The conventional dual active bridge converter with PS con-

trol scheme can achieve full control range under soft switching

while the amplitude matching of and is naturally

matching. However, when the amplitude of and is not

matching, the soft switching range is rapidly reduced [10]. By

adopting PWM control of and in this paper, the amplitude

matching of and is completely guaranteed in different

battery voltage. Therefore, this converter can satisfy (6) or (7)well from no load to full load under PSP control [12]. In other

words, compared with PS control, PSP control can expand the

ZVS range to maximum in entire battery voltage range.

IV. CONTROL STRATEGY

The control strategy of PSP is realized with two individual

controllers, as shown in Fig. 5. The BDC is difficult to control

stably because of the different small signal characteristics in dif-

ferent operation mode. In this paper, a novel control strategy

with one port voltage regulation and another port current regu-

lation is proposed. By sampling one port voltage (the port,

) and another port current (the battery port, ), the controllercan realize the voltage regulation and current regulation in dif-

ferent energy transmission direction, respectively. The control

strategy unifies the control system, simplifies the control circuit,

and makes energy bidirectional conversion free [5].

The block diagram of phase-shift angle controller is shown

in Fig. 5, which is used to control the magnitude and direction

of the transmitted power. When the voltage value on side is

higher than the reference , the converter operates in buck

mode and is controlled by single current closed-loop. The con-

stant-current setting for the low voltage side is decided by the

current limiter, which can be regulated according to the charge

condition of the battery. When the voltage value on side is

lower than the reference, the converter will operate in boostmode, and be controlled by current and voltage dual closed-

loops. The maximum discharge current of the battery is limited

by the current limiter. By selecting appropriate control parame-

ters, this variable structure controller can improve the steady and

dynamic performance of the system. The further studies about

theoretical model analysis and choice criteria of the controller

parameters will be presented in a coming paper.

The duty cycle controller realizes the amplitude matchingof and when varies. By sampling the voltage of

clamping capacitor and the voltage /2 of the port,

the controller can yield a signal which is transferred to the

PWM chip (SG3525).Here the voltage equals the amplitude

of , and the voltage is equal to the amplitude of

the secondary voltage reflected to the primary. When the

voltage value is higher than the clamping capacitor

voltage , the duty cycle controller makes the signal

rising. Sequentially, the duty cycle of and is increased

to raise the voltage in accordance with the signal . Fi-

nally, the amplitude matching of and can be achieved.

Contrarily, the duty cycle of and is reduced to match

the amplitude of and . As can be seen in Fig. 5, the dutycycle can be independently modulated when varies.

V. EXPERIMENTAL RESULTS AND DISCUSSIONS

In order to verify the operation principle of the proposed con-

verter, a 1.5 kW prototype was built in laboratory. The specifi-

cations of the converter are given as follows:

1) The battery voltage of side: VDC.

2) The rated voltage of side: VDC.

3) Rated power: kW.

4) The turns ratio of the transformer: : .

5) The leakage inductance of the transformer: H.

6) The inductance: H.7) The clamping capacitor: F.

8) The capacitors: F.

9) Switches and : APT20M11JFLL.

10) Switches and : APT77N60JC3.

11) Switches and : APT20M16LFLL.

12) Switching frequency: kHz.

Fig. 6(a) and (b) show the experimental waveforms of the

leakage inductor current , the primary voltage , and the

secondary voltage at V in Boost mode under PSP

and PS control, respectively. Since the amplitudes of and

are matched in this case, the maximum current of under PSP

control and PS control is the same. Fig. 6(c) and (d) show theexperimental waveforms of the leakage inductor current , the

primary voltage , and the secondary voltage at

V in Boost mode under PSP control and PS control. In this case,

the amplitudes of and are not matched under PS control.

Therefore, the current stress of with PS control rises rapidly.

As can be seen from Fig. 6(a) and (c), the amplitude matching of

and is guaranteed in different battery voltage. Therefore,

the duty cycle controller is valid.

Fig. 7(a), (b) and (c) showthe gate drive signal, voltage across

the drain and source, and the drain current of and

respectively, at V in Boost mode with 1.5 kW output

power under PSP control. Fig. 8(a), (b) and (c) show the gate

drive signal, voltage across the drain and source, and the draincurrent of and , respectively, at V and


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Fig. 5. Control scheme (UC3875 is a phase shift resonant controller, CD4098 is a CMOS dual monostable multivibrator, SG3525 is a regulating pulse widthmodulator, and IR2110 is a high and low side driver).

Fig. 6. Experimental waveforms at V = 3 2 V and V = 2 2 V. (a) PSP control at V = 3 2 V, V = 2 7 0 V, f = 0 : 3 5 ; d = 0 : 5 ; and P = 1 5 0 0 W. (b) PScontrol at V = 3 2 V, V = 2 7 0 V, = 0 : 3 5 ; d = 0 : 5 , and P = 1 4 9 0 W. (c) PSP control at V = 2 2 V, V = 2 7 0 V, = 0 : 8 8 ; d = 0 : 6 5 , and P = 1 4 1 0 W. (d) PS control at V = 2 2 V, V = 2 7 0 V, = 0 : 0 8 ; d = 0 : 5 , and P = 2 5 0 W.

A in Buck mode with 1.5 kW output power underPSP control. Fig. 7 and Fig. 8 illustrate that all the switches

realize ZVS. The experimental results are in agreement with thetheoretical analysis well.


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Fig. 7. Gate drive signal, the voltage across the drain and source, and the drain current of the switches at full load andV =

30V in Boost mode. (a)S

. (b)S

.(c)

S

.

Fig. 9 shows the dynamic experimental waveforms of energy

bidirectional conversion process, from up to bottom are voltage

and current . When the voltage on port is higher than

the reference value, the bidirectional dcdc converter charges

the battery with constant current. When the voltage on port

drops, the battery turns to discharge and maintains the voltage

at 270 VDC. The experimental results convinced that the novel

control strategy with one port voltage regulation and anotherport current regulation can control energy bidirectional conver-

sion freely. The response time of voltage rebuilding is 10ms.

Therefore, this converter has high steady and dynamic perfor-

mance.

Fig. 10(a) shows the overall efficiency curves at different

load, different transmission direction, and different voltage

with the PSP control. In this figure, the power transmitted from

to is defined as positive, and the power transmitted from

to is negative. We can see that the efficiency is higher in

high battery voltage (such as V, the highest

in Boost mode). Unfortunately, the efficiency is lower in low

battery voltage (such as V, the highest in

Boost mode). This degradation is due to the increase of conduc-tion loss with the battery voltage decreases. Fig. 10(b) shows

the efficiency curves of the converter under PSP control and PS

control in Boost mode. From Fig. 10(b), it can be easily found

that PSP control can achieve higher efficiency than PS control,

especially in low battery voltage. The experimental results are

in agreement with Fig. 4.

VI. CONCLUSION

A novel ZVS bidirectional dcdc converter with PS plus

PWM control is proposed in this paper, which has the following

advantages.

1) All switches realize ZVS in a wide range of load variation

while input or output voltage varies.

2) The PS plus PWM control reduces the circulating current.

3) The converter avoids the voltage spike of and with

the use of an active clamping branch and .

4) The control strategy realizes energy conversion freely,

which has high steady and dynamic performance.

These merits are verified by a 2232 V/270 V 1.5 kW pro-

totype. It can be concluded, this kind of converter is extremely

suitable for aircraft HVDC power supply system and UPSsystem.


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Fig. 8. Gate drive signal, the voltage across the drain and source, and the drain current of the switches at full load andV = 3 0 0

V in Buck mode. (a)S

. (b)

S

. (c)S

.

Fig. 9. Waveform of energy bidirectional Transmitted.

APPENDIX A

From Fig. 1(b), we can see that the controlling of andis to match the amplitude of and in the stage .

The following equation is satisfied:

(A1)

The average voltage of in one switching period is zero

(A2)

Further

(A3)

Substituting (A2) and (A3) into (A1), the following is found:

(A4)

APPENDIX B

This Appendix is provided to derive the relation of output

power versus phase-shiftangle and duty cycle , the processcan be divided into four intervals.


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Fig. 10. Conversion efficiency (The power transferred from V to V is definedas positive, and the power transferred from

V

toV

is defined as negative.) (a)Efficiency with PSP control under different output power, V , and V voltage.(b) Efficiency comparison in Boost mode under the PSP control and PS control.

a) , referring to Fig. 1(b)

(B1)

(B2)

Substituting (B1) into (B2), the following is found:

(B3)

b)

(B4)

(B5)


(B6)

c) , referring to Fig. 1(c)

(B7)

(B8)


(B9)

d)

(B10)

(B11)


(B12)

Combining (B3), (B6), (B9), and (B12), the expression(3) can be obtained.


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APPENDIX C

A Bidirectional dcdc converter with Phase-shift control

strategy was introduced in [10], the following is listed:

(C1)

a)

(C2)

b)

(C3)

The RMS value of can be expressed as follows:

(C4)

Substituting (C1), (C2), and (C3) into (C4), you can find

(C5), shown at the bottom of the page.

APPENDIX D

In order to derive the RMS value of in full load in

Boost mode under PSP control, first, we should decide which

interval the phase-shift is in under different . Referring to

Appendix B, the following equations can be listed:

(D1)

(D2)

In the proposed conditions ( V, :

kW, kHz, H), the results

( V, ) can be yielded from (D1),

and (D2).

a) When , referring

to Fig. 1(b))

For ,

(D3)

For ,

(D4)

For ,

(D5)

Substituting (B1), (D3), (D4), and (D5) into (C4),

can be expressed as (D6), shown at the bottom of the page.b) When

For ,

(D7)

For ,

(D8)

For ,

(D9)

Substituting (B4), (D7), (D8), and (D9) into (C4),

can be expressed as

(D10)

Combining (D6), and (D10), the expression (5) can be

obtained.

(C5)

(D6)


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ACKNOWLEDGMENT

The authors would like to thank M. Shi, NUAA, Y. Tang,

NUAA, L. Guo, NUAA, and F. Lin , MF, Inc., for their help

during the experiments and revisions.

REFERENCES

[1] S. Inoue and H. Akagi, A bidirectioanl isolated dcdc converter as acore circuit of the next-generation medium-voltage power conversionsystem, IEEE Trans. Power Electroni., vol. 22, no. 2, pp. 535542,Mar. 2007.

[2] F. Zhang, L. Xiao, and Y. Yan, Bi-directional forward-flyback dcdcconverters, in Proc. IEEE PESC, 2004, pp. 40584061.

[3] L. Zhu, A novel soft-commutating isolated boost full-bridgeZVS-PWM dcdc converter for bidirectional high power applica-tions, IEEE Trans. Power Electron., vol. 21, no. 2, pp. 422429, Mar.2006.

[4] H.-J. Chiu and L.-W. Lin, A bidirectionaldcdc converter for fuel cellelectric vehicle driving system, IEEE Trans. Power Electron., vol. 21,no. 4, pp. 950958, Jul. 2006.

[5] H. Xiao, D. Chen, and S. Xie, A ZVS Bi-directional dcdc converter

for high-low voltage conversion, in Proc. IEEE IECON, 2005, pp.11541158.

[6] A. Emadi and M. Ehsani, Aircraft power systems: Technology, stateof the art, and future trends, IEEE AES Syst. Mag., vol. 15, pp. 2832,Jan. 2000.

[7] J. L. Duarte, M. Hendrix, and M. G. Simoes, Three-Port bidirectionalconverter for hybrid fuel cell systems, IEEE Trans. Power Electron.,vol. 22, no. 2, pp. 480487, Mar. 2007.

[8] M. Marchesoniand C. Vacca, New dcdc converterfor energystoragesystem interfacing in fuel cell hybrid electric vehicles, IEEE Trans.Power Electron., vol. 22, no. 1, pp. 301308, Jan. 2007.

[9] R. W. De Doncker, D. M. Divan, and M. H. Kheraluwala, Power Con-version Apparatus for dc/dc Conversion Using Dual Active Bridge,U.S. Patent 5 027 264, 2005.

[10] M. H. Kheraluwala, R. W. Gascoigne, and D. M. Divan, Performancecharacterization of a high-power dual active bridge dc-to-dc converter,

IEEE Trans. Ind. Appl., vol. 28, no. 6, pp. 12941031, Nov. 1992.[11] F. Z. Peng, H. Li, and G.-J. Su et al., A new ZVS bidirectional dcdc

converter for fuel cell and battery application, IEEE Trans. PowerElectron., vol. 19, no. 1, pp. 5465, Jan. 2004.

[12] D. Xu, C. Zhao, and H. Fan, A PWM plus phase-shift control bidirec-tional dcdc converter, IEEE Trans. Power Electron., vol. 19, no. 3,

pp. 666675, May 2004.[13] S.-K. Han, H.-K. Yoon,and G.-W. Moon etal., A newactive clampingzero-voltage switching PWM current-fed half-bridge converter, IEEETrans. Power Electron., vol. 20, no. 6, pp. 12711279, Nov. 2005.

Huafeng Xiao was born inHubei,China,in 1982.Hereceived the B.S. and M.S. degree in electrical engi-neering from Nanjing University of Aeronautics andAstronautics (NUAA), Nanjing, China, in 2004 and

2007, respectively, where he is currently pursuing thePh.D. degree in electrical engineering.

His main research interests include high frequencysoft-switching conversion, and photovoltaic applica-tions.

Shaojun Xie (M05) was born in Hubei, China, in1968. He received the B.S., M.S., and Ph.D. degreesin electrical engineering from Nanjing University

of Aeronautics and Astronautics (NUAA), Nanjing,China, in 1989, 1992, and 1995, respectively.

In 1992, he joined the Faculty of Electrical Engi-neering, Teaching and Research Division, and is cur-rently a Professor at the College of Automation Engi-neering, NUAA. He has authored more over 50 tech-nical papers in Journals and Conference proceedings.His main research interests include aviation electrical

power supply systems and power electronics conversion.

High Power-PP Two Inductors

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