High Frequency High Power Converters for Industrial...

High Frequency High Power Converters for

Industrial Applications

A Thesis by Publication submitted in

Partial Fulfilment of the Requirement for the

Degree of

Doctor of Philosophy

Pooya Davari

M.Sc, B.Eng (Electrical Engineering)

Science and Engineering Faculty

School of Electrical Engineering and Computer Science

Queensland University of Technology

Queensland, Australia

2013

III

Acknowledgement

My first and sincere appreciation goes to my principle supervisor A/Prof. Firuz Zare for the

continuous support of my PhD study and research, for his patience, motivation, enthusiasm,

and immense knowledge. His guidance helped me in all the time of research and writing of

this thesis. I could not have imagined having a better advisor and mentor for my PhD study.

He was more like a friend to me for being an open person to ideas, and for encouraging and

helping me to shape my interest and ideas.

I would like to express my deep gratitude and respect to Prof. Arindam Ghosh my associate

supervisor for his continuous help and support in all stages of my PhD study. I would also to

thank Prof. Peter O’Shea for all his support in signal processing aspect of my project.

My sincere appreciation goes to Prof. Gerard Ledwich for all I have learned from him

especially in the Power Engineering Group monthly meetings.

My PhD research was a multi-disciplinary project. Hence I was fortunate to be involved in

different projects and group activities. I do thank my colleagues and friends Mr. Ashkan

Heidarkhan Tehrani, Mrs. Negareh Ghasemi, and Mr. Meisam Babaie whom I had

contribution in part of their PhD projects. My sincere thanks also go to my colleagues and

friends at the Power Engineering Group, International Laboratory for Air Quality and Health,

and Biofuel Group for their encouragement, sharing knowledge and for providing a warm

research atmosphere.

I would like to convey my sincere thanks to Queensland University of Technology (QUT)

for financial support and providing me with a pleasant research area and laboratory facilities.

I gratefully acknowledge the Australian Research Council for financial assistance throughout

my research via the ARC Discovery Grant. The assistance of Laboratory Technicians and the

staff of the Research Portfolio were also greatly appreciated.

And most importantly, I am deeply and forever indebted to my parents for their love,

support and encouragement throughout my entire life. Without them I was totally lost and

never been succeeded.

Abstract

IV

Abstract

High power converters have found their way into many industries. Variety of industries

processes have increased their requirements in power level, economy of scale, efficiency,

new control method, new topologies and using new technologies (like new semiconductors).

The incentive to move to higher frequencies is not only given by the reduced sizes of the

passive components, magnetic core losses, but also the application demands. It is thus clear

that much can be gained from moving to a design that enables operation at very high

frequency. Hence in the recent years, high frequency high power converters have been one of

the most active areas in research.

In this PhD research project high power converters have been considered, regarding to their

power level, as two categories of high instantaneous and high average power converters.

High frequency high instantaneous power converters mostly known as pulsed power supplies

are studied as the main aspect of this research, and high frequency high average power

converters are investigated with the specific intention on high power piezoelectric transducer

applications as the second aspect.

The research was conducted at two levels; first the system level which mainly encapsulated

the circuit topology and control scheme and second the application level which not only

involves with real-world applications but also evaluate the proposed and designed power

supplies. In this regard, this PhD research was a multidisciplinary project and contribution

with different research groups was made as listed below:

Biofuel Research Group, Queensland University of Technology (QUT)

The International Laboratory for Air Quality and Health, QUT

MQ Photonics Research Centre, Macquarie University

Pulsed Power Laboratory, Kumamoto University

Ultrasound-System-Development, Fraunhofer Institute for Biomedical Eng

This thesis is divided into two main sections. The first section as the major goal of this

dissertation was to develop, design and practical implementation of effective pulsed power

supply topologies with respect to the application requirements considering the power

electronics techniques and current existing commercialized devices.

To gain insight into the principle and characteristics of pulsed power technology, variety of

topologies and applications were investigated. The main advantage of non-solid state

methods found to be utilizing gas-state and magnetic switches as they possess a very high

blocking voltage and fast rise time. However, as their drawbacks, they are bulky, unreliable,

Abstract

V

have short lifetime span and low repetition rate. Even with the magnetic switches, which have

a higher repetition rate, the problems remain. These conditions limit the mobility, efficiency

and increase the cost and the size of the pulsed power system. On the other hand, advances in

solid-state switches and exploiting power electronics techniques and topologies have led to

compact, efficient, high repetition rate and more reliable pulsed power systems. The main

drawbacks of solid-state switches are their limited power rating and operation speed.

Hence, the main objective was to remedy these two issues by exploiting power electronics

techniques and topologies. As the first step, different configuration of power converters found

beneficial both in improving power and operating speed limits. Using two 600V IGBTs

(Insulated Gate Bipolar Transistor) the experimental setup based on flyback converter was

evaluated up to 4kV. Simulation and practical implementation of the proposed method are

presented in Chapter 2 and Chapter 3.

To evaluate the performance of the proposed method the series configuration of the flyback

converter was extended up to 10 modules. The experimental setup utilizing just two 1400V

IGBTS was implemented with the capability of generating 40kV, however due to laboratory

health and safety issues it was tested up to 20kV. The results and analysis are addressed in

Chapter 3.

The proposed method depicted superior advantages both in improving the power rating and

rate of rise. However, it is load dependent and suitable for high impedance load. To remedy

this issue while preserving the obtained advantages a pulsed power topology based on

modularity concept was developed. Benefiting from cascaded switch-capacitor units and

effective control scheme a load independent pulsed power supply was proposed. Chapter 4

provides proposed method description and analysis.

To investigate real-world application characteristics and effect of pulsed power supply, two

different applications were considered. Exhaust gas treatment and analysing particle matter

(PM) mass and distribution is studied as the first application. In this regards, a DBD

(Dielectric Barrier Discharge) reactor with a multipoint to plate geometry electrodes was

gradually designed. A pulsed power setup based on push-pull converter with bipolar output

voltage was developed, which provides controllable parameters to excite the DBD load at

different operation point. The experiments were conducted on real diesel exhaust gas at

different voltage level from15kVpp to 19.44kVpp at 10kHz. An optimum operating point was

investigated with respect to PM distribution and highest PM removal efficiency. The analysis

and experimentations are described in Chapter 5.

Abstract

VI

As the second application, driving Xenon filled plasma lamp using pulsed power supply

was studied with the main intention of controlling the lamp intensity. Considering the

obtained results over wide range of voltage levels and operating frequencies, the plasma lamp

power consumption regarding to its intensity depicted nonlinear behaviour of plasma lamp

and the possibility of selecting an optimum operating point was found. Chapter 6 provided

the detailed analysis.

The second part of this project focused on developing high power converter for high power

ultrasound applications. In order to understand the specifications of required excitation signal

which needed to be generated using a power converter, characteristics of high power

piezoelectric transducer using different medium and operating conditions were studied. The

experimentation and frequency response analysis are presented in Chapter 7.

Based on the piezoelectric transducer characteristics outcomes, an efficient adaptive

algorithm along with a multi-level inverter was developed in order to increase the efficiency

of high power piezoelectric applications. The proposed algorithm is capable of detecting

piezoelectric resonance frequency variations and adapting the inverter operating frequency in

order to get high power conversion efficiency. Chapter 8 depicted the evaluation of the

proposed technique and effect of employing such algorithm on performance of an ultrasound

interface.

Finally, the conclusions and possible future works are drawn in Chapter 9. It is to be noted

that dealing with practical issues such as noise, EMI (Electro Magnetic Interference), high

voltage insulation, transformer saturation, spikes across the IGBTs, DSC (Digital Signal

Controller) programming and etc, caused vast studying in addition to the main objectives. In

addition to the gathered knowledge concerning the energy conversion of the power converter,

the experiments and investigated applications also yielded a large amount of valuable data

concerning converter development under different pulse conditions and load configurations.

This data is presented in a systematic manner as seven chapters (journal and conference

papers) so it can be of value for future related research.

VII

Keywords

Buck-boost converter

Corona

Current and voltage sources

Dielectric barrier discharge (DBD)

Digital signal controller (DSC)

Exhaust gas treatment

Flyback converter

Full bridge

Half bridge

H-bridge inverter

High power ultrasound

High voltage

Insulated-Gate Bipolar Transistor (IGBT)

Marx generator (MG)

Modular

Multi-level inverter

Non-Thermal Plasma (NTP)

Particulate Matter (PM)

Piezoelectric Transducer

Plasma

Power electronics

Pulse generator

Pulse width modulation (PWM)

Pulsed power

Push-pull converter

Rate of rise

Reactor

Resonance

Solid-state technology

Spark gap

Switch-capacitor

Unipolar and bipolar modulation

VIII

List of Abbreviation

AC: Alternating Current

ARFI: Acoustic Radiation Force Impulse

BERF: Biofuel Engine Research Facility

CCM: Continues Conduction Mode

CO2: Carbon Dioxide

CPC: Condensation Particle Counter

CRT: Cathode Ray Tube

CT: Current Transducer

DBD: Dielectric Barrier Discharge

DC: Direct Current

DCM: Discontinues Conduction Mode

DNA: Deoxyribonucleic Acid

DSC: Digital Signal Controller

EMF: Electromagnetic Metal Forming

EMI: Electro Magnetic Interference

FET: Field-Effect Transistor

FFT: Fast Fourier Transform

FWHM: Full Width at Half Maximum

GCT: Gate Commutated Thyristor

GTO: Gate Turn-off Thyristor

HIFU: High Intensity Focused Ultrasound

HV: High Voltage

IGBT: Insulated-Gate Bipolar Transistor

IGCT: Integrated Gate-Commutated Thyristor

MBL: Multistage Blumlein Lines

MG: Marx Generator

MOSFET: Metal-Oxide Semiconductor Field-Effect Transistor

MPC: Magnetic Pulse Compressors

NOX: Nitrogen Oxides

NTP: Non-Thermal Plasma

PFC: Power Factor Correction

PFN: Pulse Forming Network

IX

PM: Particulate Matter

PWM: Pulse Width Modulation

PZT: Piezoelectric Transducer

RC: Resistive-Capacitive

SCFM: Short Circuit Failure Mode

SHM: Structural Health Monitoring

SiC: Silicon Carbide

SMPS: Scanning Mobility Particle Sizer

SO2: Sulfur Dioxide

THD: Total Harmonics Distortion

UV: Ultraviolet Radiation

VOCs: Volatile Organic Compounds

VUV: Vacuum UV

X

Table of Contents

Acknowledgement ................................................................................................................. III

Abstract .................................................................................................................................. IV

Keywords .............................................................................................................................. VII

List of Abbreviation ........................................................................................................... VIII

Contributions.................................................................................................................... XXIV

List of Publications .......................................................................................................... XXVI

List of Chapters According to Publications and Contributions .............................. XXVIII

Scholarships and Awards ................................................................................................ XXIX

Statement of Original Authorship ................................................................................... XXX

Chapter 1 .................................................................................................................................. 1

1. Introduction ...................................................................................................................... 1

1.1. Definition of the Research Problem ........................................................................... 2

1.2. Literature Review....................................................................................................... 5

1.2.1. Introduction .......................................................................................................... 5

1.2.2. Pulsed Power (high frequency high instantaneous power converter) .................. 6

1.2.2.1. System Level ................................................................................................ 7

1.2.2.2. Application Level ....................................................................................... 24

1.2.3. High Frequency High Average Power Converter for Ultrasound Applications 33

1.2.3.1. Piezoelectric Transducer Characteristics .................................................... 34

1.2.3.2. Power Electronics Topologies for High Power Ultrasound Systems ......... 38

1.2.3.3. High Power Piezoelectric Transducer Applications ................................... 43

1.3. Account of Research Progress Linking the Research Papers .................................. 45

1.3.1. Introduction ........................................................................................................ 45

1.3.2. Pulsed Power Systems ....................................................................................... 45

1.3.2.1. Benefiting from Parallel and Series Configurations of Flyback Converter

for Pulsed Power Applications..................................................................................... 46

1.3.2.2. Flexible and modular pulsed power supply ................................................ 60

1.3.2.3. Pulsed power applications .......................................................................... 70

1.3.3. High frequency high power converters for ultrasound applications .................. 88

1.3.3.1. Power electronic converters for high power ultrasound transducers .......... 89

1.3.3.2. Improving the efficiency of high power piezoelectric transducer .............. 95

1.4. References ................................................................................................................... 106

XI

Chapter 2 .............................................................................................................................. 117

2 . Parallel and Series Configurations of Flyback Converter for Pulsed Power

Applications .......................................................................................................................... 117

2.1. Keywords ............................................................................................................... 118

2.2. Introduction ............................................................................................................ 118

2.3. Topology ................................................................................................................ 119

2.4. Proposed Method ................................................................................................... 122

2.4.1. Single Module .................................................................................................. 122

2.4.2. Parallel Modules .............................................................................................. 123

2.4.3. Series Modules ................................................................................................. 125

2.4.4. Operating Conditions ....................................................................................... 126

2.5. Simulation Results ................................................................................................. 127

2.5.1. Case1 ................................................................................................................ 128

2.5.2. Case2 ................................................................................................................ 129

2.5.3. Case3 ................................................................................................................ 130

2.5.4. Case4 ................................................................................................................ 130

2.6. Conclusion ............................................................................................................. 131

2.7. References .............................................................................................................. 131

Chapter 3 .............................................................................................................................. 134

3 . High Voltage Modular Power Supply Using Parallel and Series Configurations of

Flyback Converter for Pulsed Power Applications .......................................................... 134

3.1. Index Terms ........................................................................................................... 135

3.2. Introduction ............................................................................................................ 135

3.3. Topology ................................................................................................................ 137

3.4. Proposed Method ................................................................................................... 140

3.4.1. Single Module .................................................................................................. 141

3.4.2. Parallel Modules .............................................................................................. 142

3.4.3. Series Modules ................................................................................................. 142

3.4.4. Operating Conditions ....................................................................................... 144

3.5. Experimental Results and Discussion .................................................................... 146

3.5.1. Evaluating Parallel and Series Configurations ................................................ 146

3.5.1.1. Case1 ........................................................................................................ 147

3.5.1.2. Case2 ........................................................................................................ 150

3.5.2. High Voltage Modular Power Supply.............................................................. 151

3.6. Conclusion ............................................................................................................. 153

3.7. References .............................................................................................................. 154

XII

Chapter 4 .............................................................................................................................. 157

4 . A Flexible Solid-State Pulsed Power Topology ............................................................ 157

4.1. Keywords ............................................................................................................... 158

4.2. Introduction ............................................................................................................ 158

4.3. Topology ................................................................................................................ 160

4.4. Control Strategy and Operating Modes ................................................................. 161

4.4.1. Current & Voltage Control .............................................................................. 161

4.4.2. Charging & Load Control ................................................................................ 163

4.5. Simulation Results and Analysis ........................................................................... 167

4.5.1. Case1 ................................................................................................................ 167

4.5.2. Case2 ................................................................................................................ 168

4.5.3. Case3 ................................................................................................................ 168

4.6. Conclusion ............................................................................................................. 170

4.7. References .............................................................................................................. 170

Chapter 5 .............................................................................................................................. 173

5 . Effect of Pulsed Power on Particle Matter in Diesel Engine Exhaust Using a DBD

Plasma Reactor..................................................................................................................... 173

5.1. Index Terms ........................................................................................................... 174

5.2. Introduction ............................................................................................................ 174

5.3. Experimental method ............................................................................................. 176

5.3.1. Experimental Setup .......................................................................................... 176

5.3.2. DBD Reactor .................................................................................................... 178

5.3.3. Bipolar Pulsed Power Supply .......................................................................... 178

5.4. Results and Discussion .......................................................................................... 181

5.4.1. Plasma effect on PM Size Distribution ............................................................ 181

5.4.2. PM removal efficiency ..................................................................................... 187

5.4.3. Plasma Effect on PM Mass Reduction............................................................. 190

5.5. Conclusion ............................................................................................................. 191

5.6. References .............................................................................................................. 191

Chapter 6 .............................................................................................................................. 196

6 . Analysing DBD Plasma Lamp Intensity versus Power Consumption Using a Push-

Pull Pulsed Power Supply ................................................................................................... 196

6.1. Keywords ............................................................................................................... 197

6.2. Introduction ............................................................................................................ 197

6.3. Experimental Setup ................................................................................................ 198

XIII

6.3.1. DBD Lamp ....................................................................................................... 198

6.3.2. Pulsed Power Supply ....................................................................................... 199

6.3.3. Measurements .................................................................................................. 201

6.4. Results and Discussion .......................................................................................... 203

6.5. Conclusion ............................................................................................................. 205

6.6. Acknowledgement ................................................................................................. 197

6.7. References .............................................................................................................. 205

Chapter 7 .............................................................................................................................. 208

7 . Power Electronic Converters for High Power Ultrasound Transducers ................... 208

7.1. Keywords ............................................................................................................... 209

7.2. Introduction ............................................................................................................ 209

7.3. Experimental Procedure ......................................................................................... 211

7.4. Conclusion ............................................................................................................. 221

7.5. References .............................................................................................................. 222

Chapter 8 .............................................................................................................................. 224

8 . Improving the Efficiency of High Power Piezoelectric Transducers for Industrial

Applications .......................................................................................................................... 224

8.1. Introduction ............................................................................................................ 225

8.2. Methodology and Approach to Estimate Resonance Frequency ........................... 228

8.2.1. Excitation Signal .............................................................................................. 228

8.2.2. Estimating Impedance ...................................................................................... 229

8.2.3. Extracting Resonant Frequencies ..................................................................... 231

8.2.4. Noise Issues ..................................................................................................... 233

8.3. Simulation and Experiment.................................................................................... 233

8.3.1. Simulation Results ........................................................................................... 234

8.3.2. Experimental Results ....................................................................................... 236

8.3.3. Ultrasound Interface......................................................................................... 240

8.4. Conclusions ............................................................................................................ 241

8.5. References .............................................................................................................. 242

Chapter 9 .............................................................................................................................. 245

9 . Conclusions and Further Research ............................................................................... 245

9.1. Conclusions ............................................................................................................ 246

9.1.1. Improving solid-state pulsed power supplies regarding to the low power rating

and operating speed limits of the current switching devices ......................................... 246

XIV

9.1.2. Considering load characteristics and effect of the load on pulsed power supply

248

9.1.3. Increasing the efficiency of high power ultrasound applications .................... 249

9.1.4. Summary of advantages and drawbacks of proposed topologies and methods

250

9.2. Further Research .................................................................................................... 256

XV

List of Figures

Chapter 1

Fig. 1.1 General scheme of a pulsed power system. .................................................................. 7

Fig. 1.2 Basic Capacitive (a) and inductive (b) schemes for pulse generation. ......................... 8

Fig. 1.3 Pulse shape and its parameters. .................................................................................... 8

Fig. 1.4 Using multiple switches to overcome the switches power rating problem, (a) parallel

for high current and (b) series for high voltage applications. .................................................. 10

Fig. 1.5 Conventional Marx generator. .................................................................................... 11

Fig. 1.6 Conventional pulse forming network [45].................................................................. 12

Fig. 1.7 PFN with reversal voltage polarity [18]. .................................................................... 13

Fig. 1.8 MBL or adder topology [18]. ..................................................................................... 13

Fig. 1.9 Basic MPC: (a) circuit, (b) Voltage waveforms [47]. ................................................ 15

Fig. 1.10 Benefiting from diodes in MG [61] and PFN [18] topologies. ................................ 17

Fig. 1.11 Conventional Marx generator based on solid-state switches [60, 62]. ..................... 17

Fig. 1.12 Pulsed power generator based on magnetic pulse compression [10, 23]. ................ 18

Fig. 1.13 Circuit of a module of 6 MOS-FETs, connected in parallel [72]. ............................ 19

Fig. 1.14 Test circuit of MOS-FET switching unit [72]. ......................................................... 19

Fig. 1.15 High voltage switches (stack of switches) optimized for pulsed power applications:

(a) ABB [49], (b) BEHLKE [73]. ............................................................................................ 20

Fig. 1.16 Electrical circuit of the pulsed power generator using SI-thyristor [72]. ................. 21

Fig. 1.17 Waveforms of L2 current and the output voltage [72]. ............................................ 22

Fig. 1.18 (a) flyback, and (b) forward topologies [53]. ........................................................... 23

Fig. 1.19 (a) half bridge, and (b) full bridge [53]. ................................................................... 24

Fig. 1.20 Electrical equivalent circuit of a DBD load [55]. ..................................................... 25

Fig. 1.21 Photograph of a discharge induced in water [87]. .................................................... 30

Fig. 1.22 Photograph of effect of water discharges in algae treatment [27]. ........................... 31

Fig. 1.23 a) Biological cell structure, b) Double-shell model of a biological cell [30]. .......... 33

Fig. 1.24 Resonance and anti-resonance frequencies in piezoelectric impedance frequency

response.................................................................................................................................... 35

Fig. 1.25 Depiction of typical resonance frequency variations of piezoelectric devices due to

structural changes..................................................................................................................... 36

Fig. 1.26 The Van Dyke Model [107]...................................................................................... 36

XVI

Fig. 1.27 The Guan model-unloaded piezoelectric [107]. ....................................................... 37

Fig. 1.28 The extended Van Dyke model-loaded piezoelectric [107]. .................................... 37

Fig. 1.29 The complete Guan model-loaded piezoelectric [107]. ............................................ 38

Fig. 1.30 Two-stage resonant inverter based on flyback topology [126]. ............................... 39

Fig. 1.31 Two-stage resonant inverter based on H-bridge topology [5]. ................................. 40

Fig. 1.32 Effect of using multi-level waveform in harmonic elimination. .............................. 41

Fig. 1.33 Three-level PWM inverter combined with LLCC-filter for driving high power

piezoelectric transducer [102]. ................................................................................................. 42

Fig. 1.34 Typical output voltage and current of employed hybrid three-level inverter when

applying to a piezoelectric transducer [102]. ........................................................................... 42

Fig. 1.35 Basis of the ultrasonic atomizer [5]. ......................................................................... 44

Fig. 1.36 Flyback converter circuit with transformer equivalent circuit model. ..................... 47

Fig. 1.37 Flyback converter series and parallel connection (secondary side), a) Parallel, b)

Series. ....................................................................................................................................... 48

Fig. 1.38 Energy transmission in a single module flyback converter. ..................................... 48

Fig. 1.39 Effect of damping ratio on output voltage and rate of rise in a single flyback

module...................................................................................................................................... 54

Fig. 1.40 Output voltage waveform of single, parallel and series modules in Case1

(Co1=Co2=Co). .......................................................................................................................... 55


(Cop=Cos=Co). .......................................................................................................................... 56

Fig. 1.42 Output voltage waveform of single, parallel and series modules with 0.2ms delay

between modules gate signals. ................................................................................................. 56

Fig. 1.43 The effect of the load and the damping factor on the system performance. ............. 57

Fig. 1.44 Hardware setup for two flyback modules. ................................................................ 58

Fig. 1.45 (a) current sharing at the primary side for both parallel and series connections, (b)

output voltage waveform of single, parallel and series modules in Case1 (Co1=Co2=Co). ..... 59

Fig. 1.46 Block diagram of the implemented setup. ................................................................ 61

Fig. 1.47 Hardware setup for 10-series flyback modules. ....................................................... 62

Fig. 1.48 Measured output voltage of 10 series modules (VLoad) and a single module (Vo1). 62

Fig. 1.49 Proposed pulsed power topology for n level switch-capacitor units. ....................... 64

Fig. 1.50 The charging and supplying the load operating modes. ........................................... 66

Fig. 1.51 Charging and supplying the load control algorithm flowchart, a) based on individual

voltage feedback of each unit and b) based only one voltage feedback. ................................. 67

XVII

Fig. 1.52 The effect of the load impedance on the generated output voltage. ......................... 68

Fig. 1.53 Voltage waveforms with relative switching signal patterns. .................................... 69

Fig. 1.54 Voltage across the critical components. ................................................................... 69

Fig. 1.55 Schematic diagram of plasma treatment system developed at QUT engine lab. ..... 72

Fig. 1.56 DBD reactor in Solidworks: a) schematic, b) cross-section. .................................... 73

Fig. 1.57 Pulsed power supply circuit schematic diagram (push-pull inverter) ...................... 74

Fig. 1.58 A typical measured output voltage of the employed pulsed power supply. ............. 75

Fig. 1.59 Electrical Hardware setup with the DBD load. ........................................................ 75

Fig. 1.60 DBD image. .............................................................................................................. 76

Fig. 1.61 Particle Size Distribution (2000rpm, 25%Load, 19.44 kVpp). ................................ 77

Fig. 1.62 Particle Size Distribution (2000rpm, 25%Load, 17 kVpp). ..................................... 78

Fig. 1.63 Particle Size Distribution (2000rpm, 25%Load, 15 kVpp). ..................................... 79

Fig. 1.64 PM removal as a function of PM size (2000rpm, 25%Load, 19.44 kVpp) .............. 80

Fig. 1.65 PM removal as a function of PM size (2000rpm, 25%Load, 17 kVpp) ................... 80

Fig. 1.66 PM removal as a function of PM size (2000rpm, 25%Load, 15 kVpp) ................... 81

Fig. 1.67 Dielectric barrier discharge lamp ............................................................................. 84



Fig. 1.70 V-Q cyclogram of the DBD lamp as a basis of power consumption calculation. .... 86

Fig. 1.71 Comparing different operation points regarding to the varied applied voltage levels

and repetition rates, a) applied voltage versus plasma lamp power consumption, b) applied

voltage versus plasma lamp intensity. ..................................................................................... 87

Fig. 1.72 block diagram of a lab prototype for test 1 and test 2 .............................................. 90

Fig. 1.73 (a) Input signals at 39 kHz (b) output signals at 39 kHz (c) Input signals at 61 kHz

(d) output signals at 61 kHz. .................................................................................................... 91

Fig. 1.74 (a) test 1: summation of two output signals (at 39 kHz & 61 kHz) for Vin=15V and

Vin=30V, (b) test 2: two output signals for Vin=15V and Vin=30V, (c) comparing the results

of test 1 and test 2 for Vin=15V, (d) comparing the results of test 1 and test 2 for Vin=30V. 92

Fig. 1.75 Schematic diagram of the setup using an inverter. ................................................... 92

Fig. 1.76 Hardware Setup. ....................................................................................................... 93

Fig. 1.77 Captured results of third experiment ........................................................................ 93

Fig. 1.78 Applying filter to the inverter output. ....................................................................... 94

Fig. 1.79 Captured results of forth experiment (applying a filter). .......................................... 94

Fig. 1.80 Captured results of the last experiment. ................................................................... 94

XVIII

Fig. 1.81 Exploiting online resonance frequency estimation in high power applications of

piezoelectric devices. ............................................................................................................... 97

Fig. 1.82 Frequency responses: a) ideal pulse frequency response, b) practical pulse

frequency response................................................................................................................... 98

Fig. 1.83 Extracting resonant frequencies: a) change of slope at relative minimum (resonant

frequency), b) flowchart of the proposed algorithm. ............................................................. 100

Fig. 1.84 Experimental setup for the proposed method. ........................................................ 102

Fig. 1.85 Experimental results obtained for the piezoelectric devices impedance response: 103

Fig. 1.86 Obtained results for the ultrasound interface: (a) applied uni-polar pulse at 39 kHz

in time domain, (b) frequency response of the applied uni-polar pulse (input signal), (c)

frequency response of the output signal when the input is a uni-polar pulse, (d) applied multi-

level pulse at 39 kHz in time domain, (e) frequency response of the applied multi-level pulse

(input signal), (f) frequency response of the output signal when the input is a multi-level

pulse. ...................................................................................................................................... 105

Chapter 2


Fig. 2.2 Operating modes in a flyback converter. .................................................................. 122


Series. ..................................................................................................................................... 124

Fig. 2.4 Energy transmission in a single module flyback converter. ...................................... 125


(Co1=Co2=Co). ......................................................................................................................... 129


(Cop=Cos=Co). ......................................................................................................................... 129

Fig. 2.7 Output voltage waveform of single, parallel and series modules with 0.2ms delay

between modules gate signals. ............................................................................................... 130

Fig. 2.8 The effect of the load and the damping factor on the system performance. .............. 131

Chapter 3


Fig. 3.2 Operating modes in a flyback converter. .................................................................. 139


Series. ..................................................................................................................................... 140

XIX

Fig. 3.4 Energy transmission in a single module flyback converter. ..................................... 141

Fig. 3.5 Effect of damping ratio on output voltage and rate of rise in a single flyback module.

................................................................................................................................................ 146

Fig. 3.6 Hardware setup for two flyback modules. ................................................................ 147

Fig. 3.7 (a) current sharing at the primary side for both parallel and series connections, (b)

output voltage waveform of single, parallel and series modules in Case1 (Co1=Co2=Co). ... 149

Fig. 3.8 Voltage across the switches in Case1. ...................................................................... 150

Fig. 3.9 Voltage across the diodes in Case1. ......................................................................... 150


(Cop=Cos=Co). ........................................................................................................................ 151

Fig. 3.11 Block diagram of the implemented setup. .............................................................. 152

Fig. 3.12 Hardware setup for 10-series flyback modules. ..................................................... 152

Fig. 3.13 Measured output voltage of 10 series modules (VLoad) and a single module (Vo1).

................................................................................................................................................ 153

Chapter 4

Fig. 4.1 Proposed pulsed power topology for n level switch-capacitor units. ....................... 160

Fig. 4.2 Operating modes for positive buck-boost converter. ................................................ 162

Fig. 4.3 The charging and supplying the load operating modes. .......................................... 164

Fig. 4.4 Charging and supplying the load control algorithm flowchart (based on individual

voltage feedback of each unit). .............................................................................................. 165

Fig. 4.5 Charging and supplying the load control algorithm flowchart (based on only one

voltage feedback). .................................................................................................................. 166

Fig. 4.6 The effect of the load impedance on the generated output voltage.. ........................ 168

Fig. 4.7 Voltage waveforms with relative switching signal patterns. .................................... 169

Fig. 4.8 Voltage across the critical components. ................................................................... 169

Chapter 5

Fig. 5.1 Schematic diagram of plasma treatment system developed at QUT engine lab. ..... 177

Fig. 5.2 DBD reactor in Solidworks: a) schematic, b) cross-section. .................................... 178


Fig. 5.4 A typical measured output voltage of the employed pulsed power supply. ............. 180


XX

Fig. 5.6 The measured electrical parameters: a) output voltages across the DBD load, b)

output current, and c) voltage stress across the switch. ......................................................... 182

Fig. 5.7 DBD image. .............................................................................................................. 183

Fig. 5.8 V-Q cyclogram of the DBD load as a basis of power consumption calculation. ..... 183

Fig. 5.9 Particle Size Distribution (2000rpm, 25%Load, 19.44 kVpp). ................................ 185

Fig. 5.10 Particle Size Distribution (2000rpm, 25%Load, 17 kVpp). ................................... 186

Fig. 5.11 Particle Size Distribution (2000rpm, 25%Load, 15 kVpp). ................................... 187

Fig. 5.12 PM removal as a function of PM size (2000rpm, 25%Load, 19.44 kVpp) ............ 188

Fig. 5.13 PM removal as a function of PM size (2000rpm, 25%Load, 17 kVpp) ................. 188

Fig. 5.14 PM removal as a function of PM size (2000rpm, 25%Load, 15 kVpp) ................. 189

Chapter 6

Fig. 6.1 Dielectric barrier discharge lamp ............................................................................. 199



Fig. 6.4 Output voltage of the employed pulsed power supply: a) at 10 kHz, b) typical

measured output voltages at 10 kHz and 5 kHz. .................................................................... 201

Fig. 6.5 V-Q cyclogram of the DBD lamp as a basis of power consumption calculation. .... 202

Fig. 6.6 Comparing different operation points regarding to the varied applied voltage levels

and repetition rates, a) applied voltage versus plasma lamp power consumption, b) applied

voltage versus plasma lamp intensity. ................................................................................... 204

Fig. 6.7 The DBD lamp light intensity measured for varied applied voltage levels.............. 204

Chapter 7

Fig. 7.1 (a) Van Dyke Model, (b) a power converter, (c) output voltage of power converter.

................................................................................................................................................ 211

Fig. 7.2 A block diagram of a lab prototype for test 1 and test 2. ......................................... 213

Fig. 7.3 (a) input signals at 39 kHz (b) output signals at 39 kHz (c) input signals at 61 kHz

(d) output signals at 61 kHz ................................................................................................... 214

Fig. 7.4 (a) Test 1: summation of two output signals (at 39 kHz & 61 kHz) for Vin=15V and

Vin=30 V (b) Test 2: two output signals for Vin=15V and Vin=30 V. (c) Comparing

the results of test 1 and test 2 for Vin=15 V. (d) Comparing the results of test 1 and test 2 for

Vin=30 V. .............................................................................................................................. 215

Fig. 7.5 The experimental setup. ............................................................................................ 216

XXI

Fig. 7.6 A block diagram of test 3. ........................................................................................ 216

Fig. 7.7 The results of test 3 (a) input signals and (b) output signals. ................................... 217

Fig. 7.8 A block diagram of test 4. ........................................................................................ 218

Fig. 7.9 The results of test 4 (a) input signals and (b) output signals. ................................... 219

Fig. 7.10 A block diagram of test 5. ...................................................................................... 219

Fig. 7.11 The results of test 5 (a) input signals and (b) output signals. ................................. 220

Fig. 7.12 A block diagram of test 6. ...................................................................................... 220

Fig. 7.13 The results of test 6 (a) input signals and (b) output signals. ................................. 221

Chapter 8

Fig. 8.1 Effect of using multi-level waveform in harmonic elimination. .............................. 227

Fig. 8.2 Exploiting online resonance frequency estimation in high power applications of

piezoelectric devices. ............................................................................................................. 228

Fig. 8.3 a) Ideal pulse frequency response, b) Captured voltage c) Captured response

(current), d) Practical pulse frequency response. ................................................................... 230

Fig. 8.4 Extracting resonant frequencies: a) change of slope at relative minimum (resonant

frequency), b) flowchart of the proposed algorithm. ............................................................. 232

Fig. 8.5 Simulation results for piezoelectric Type A: a) circuit model, b) impedance response.

................................................................................................................................................ 235

Fig. 8.6 Experimental setup for the proposed method. .......................................................... 237

Fig. 8.7 Experimental results obtained for the piezoelectric devices impedance response: (a)

Type A, (b) Type B. .............................................................................................................. 238

Fig. 8.8 Obtained results for the ultrasound interface: (a) applied uni-polar pulse at 39 kHz in

time domain, (b) frequency response of the applied uni-polar pulse (input signal), (c)

frequency response of the output signal when the input is a uni-polar pulse, (d) applied multi-

level pulse at 39 kHz in time domain, (e) frequency response of the applied multi-level pulse

(input signal), (f) ) frequency response of the output signal when the input is a multi-level

pulse. ...................................................................................................................................... 241

XXII

List of Tables

Chapter 1

Table 1.1 Various switch parameters ....................................................................................... 16

Table 1.2 Summary of typical pulsed power applications requirements [30]. ........................ 26

Table 1.3 Dielectric strength of common materials. ................................................................ 28

Table 1.4 Output voltage and rate of rise when Co1=Co2=… CoN=Co. ..................................... 51

Table 1.5 Output voltage and rate of rise when Cop=Cos=Co. ............................................... 51

Table 1.6 Simulation Parameters ............................................................................................. 55

Table 1.7 Simulation Parameters in Case 1 and 2 ................................................................... 55

Table 1.8 Simulation Parameters ............................................................................................. 67

Table 1.9 PM Mass Reductions at Different Voltage Levels .................................................. 82

Table 1.10 Considered voltages and frequencies for different experiments ............................ 87

Table 1.11 Test conditions and setups ..................................................................................... 90

Table 1.12 Estimated resonant frequencies of the Type A piezoelectric device ................... 103

Table 1.13 Estimated resonant frequencies of the Type B piezoelectric device ................... 104

Chapter 2

Table 2.1 Output voltage and rate of rise when Co1=Co2=… CoN=Co..................................... 126

Table 2.2 Output voltage and rate of rise when Cop=Cos=Co. .............................................. 126

Table 2.3 Simulation Parameters ............................................................................................ 128

Table 2.4 Simulation Parameters in Case 1 and 2 .................................................................. 128

Chapter 3

Table 3.1 Output voltage and rate of rise when Co1=Co2=… CoN=Co. ................................... 143

Table 3.2 Output voltage and rate of rise when Cop=Cos=Co. ................................................ 143

Chapter 4

Table 4.1 Simulation Parameters ........................................................................................... 168

Chapter 5

Table 5.1 PM Mass Reduction at Different Voltage Levels .................................................. 190

Chapter 6

Table 6.1 Considered voltages and frequencies for different experiments ............................ 203

XXIII

Chapter 7

Table 7.1 Test conditions and setups ..................................................................................... 213

Chapter 8

Table 8.1 Values of the Components in the Electrical Circuit Model ................................... 236

Table 8.2 Estimated Resonant Frequencies in Simulation..................................................... 236

Table 8.3 Estimated resonant frequencies of the Type A piezoelectric device ..................... 238

Table 8.4 Estimated resonant frequencies of the Type B piezoelectric device ..................... 239

Table 8.5 Survey on advantages and drawbacks of mentioned methods ............................... 239

XXIV

Contributions

Effective Flyback Converter Configurations for Pulsed Power

Applications

Improving the voltage level and rate of rise

Studying the effect of load impedance on system performance

Investigating the fault conditions

Developing a High Voltage Modular Pulsed Power Supply

Applying modularity concept on flyback topology

Studying the effect of load impedance on system performance

Investigating the performance of parallel and series

configurations

Reducing the number of switches to min imum

Improving the voltage level and rate of rise

Ability to produce up to 40kV pulses

Easy diagnostic due to modularity concept

Proposing a Flexible Pulsed Power Topology

Utilizing cascaded switch-capacitor topology

An efficient load independent topology

Studying the effect of load impedance on the system performance

Using modularity concept

Ability to operate at high repetition rate

Developing a smart control algorithm which can control the flow

of energy and prevent from any possible faults

Easy diagnostic

Effect of Pulsed Power on Particle Matter Distribution

Efficient method for exhaust gas treatment

Developing a DBD load

Studying the effect of pulsed power at different voltage levels

Optimizing the system operation

Studying the particle matt er distribution and mass variations

XXV

Developing High Frequency High Power Converter for Plasma

Lamp Applications

Studying the plasma lamp power consumption regarding to its

intensity

Investigating DBD plasma lamp characteristics at varied power

levels

Designing a power converter for plasma lamp applications with

variable voltage and frequency

Studying Desirable Characteristics of Power Converters for High

Power Ultrasound Applications

Investigating high power piezoelectric transducer requirement s

Studying linear or non-linear behaviour of high power

piezoelectric transducer

Investigating effect of different medium on ultrasound system

performance

Improving the efficiency of power converter based on high power

piezoelectric transducer behaviour variati ons

Applying an efficient frequency adaptation algorithm

Developing frequency detection algorithm

Studying the behaviour of piezoelectric transducer

Increasing the performance of high power ultrasound system

XXVI

List of Publications

The Queensland University of Technology (QUT) allows the presentation of a thesis for the

degree of Doctor of Philosophy in the format of published or submitted papers, where such

papers have been published, accepted or submitted during the period of candidature. This

thesis is composed of seven published/submitted papers, of which six have been published

and one is accepted. Note that each paper is selected for the thesis as one chapter.

Peer Reviewed Journal Articles:

1. P. Davari, F. Zare, A. Ghosh, and H. Akiyama, "High Voltage Modular Power

Supply Using Parallel and Series Configurations of Flyback Converter for Pulsed

Power Applications," Plasma Science, IEEE Transactions on, vol.40, no.10, pp.2578-

2587, Oct. 2012.

2. P. Davari, N. Ghasemi, F. Zare, P. O'Shea, and A. Ghosh, "Improving the efficiency

of high power piezoelectric transducers for industrial applications," Science,

Measurement & Technology, IET, vol. 6, pp. 213-221, 2012.

3. M. Babaie, P. Davari, F. Zare, Z. Ristovski, and R. Brown, "Effect of Pulsed Power

on Particle Matter in Diesel Engine Exhaust Using a DBD Plasma Reactor," Plasma

Science, IEEE Transactions on, vol.41, no.8, pp.2349-2358, Aug. 2013.

Peer Reviewed International Conference Articles:

4. P. Davari, F. Zare, and A. Ghosh, “Analyzing DBD Plasma Lamp Intensity versus

Power Consumption Using a Push-Pull Pulsed Power Supply", EPE, Lille, France,

2013 (accepted).

5. P. Davari, F. Zare, and A. Ghosh, “A Flexible Solid-State Pulsed Power Topology",

International Power Electronics and Motion Control (EPE-PEMC), 2012 the 15th

IEEE Conference on, 2012.

6. P. Davari, F. Zare, and A. Ghosh, “Parallel and Series Configurations of Flyback

Converter for Pulsed Power Applications", Industrial Electronics and Applications

(ICIEA), 2012 the 7th IEEE Conference on, 2012.

XXVII

7. N. Ghasemi, F. Zare, P. Davari, P. Weber, C. Langton, and A. Ghosh, “Power

Electronic Converters for High Power Ultrasound Transducers", Industrial Electronics

and Applications (ICIEA), 2012 the 7th IEEE Conference on, 2012.

XXVIII

List of Chapters According to Publications and Contributions

High Frequency High Power Converters for


Literature Review

"Stage two, confirmation and final thesis"

Research Problem #1

Power Electronics Converters for Pulsed

Power Topologies

Chapter 2:

Applying Flyback Converter for Pulsed Power

"Parallel and Series Configurations of Flyback Converter for Pulsed Power Applications"

The 7th IEEE Conference on Industrial Electronics and Applications ICIEA2012

Chapter3:

High Voltage Modulare Pulsed Power Supply

"High Voltage Modular Power Supply Using Parallel and Series Configurations of Flyback

Converter for Pulsed Power Applications"

IEEE Transactions on Plasma Science, vol.40, no.10, pp.2578-2587, 2012 Chapter4:

Flexible Pulsed Power Supply

"A Flexible Solid-State Pulsed Power Topology"

The 15th IEEE Conference on Power Elctroncis and Motion Control, EPE-PEMC 2012

Research Problem #2

Investigating Effect of Pulsed Power in Different

Applications

Chapter 5: Effect of Pulsed Power on Particle Matter

Distribution

"Effect of Pulsed Power on Particle Matter in Diesel Engine Exhaust Using a DBD Plasma

Reactor"

IEEE Transactions on Plasma Science, vol. 41, no.8, pp.2349-2358, 2013

Chapter 6:

High Frequency High Voltage Converter for Plasma Lamp Applications

"Analyzing Intensity Versus Power Consumption in DBD Plasma Lamp Using a High Frequency Push-

Pull Converter"

15th European Conference on Power Electronics and Applications, EPE'13 ECCE Europe

Research Problem #3

High Frequency High Power Converters for High Power Piezoelectric Transducers

Chapter 7:

Power Converter for High Power Ultrasound Applications

"Power Electronic Converters for High Power Ultrasound Transducers"

The 7th IEEE Conference on Industrial Electronics and Applications ICIEA2012

Chapter 8:

Desgining a Power Converter based on High Power Piezoelectric Charactrestics

"Improving the Efficiency of High Power Piezoelectric Transducers for Industrial

Applications"

IET Sci. Meas.Technol, vol.6, no.4, pp.213-221, 2012

XXIX

Scholarships and Awards

Queensland University of Technology Postgraduate Research Award

(QUTPRA) for PhD degree for three years (2010-2013).

Tuition Fee Waiver Scholarship by Queensland University of Technology

for PhD degree for three years (2010-2013).

Two Supervisor Top-Up Scholarships for PhD degree, one with the

period of 1.5 year (2011-2013) and one for the period of one year (2012-

2013).

Outstanding HDR Student of the Month, Science and Engineering

Faculty, Queensland University of Technology, June 2012.

Queensland Safe Work Awards (2011)

QUT was highly commended for best solution to an identified electrical safety issue.

I was being part of the team in designing a prototype acrylic LV testing enclosure to

house circuiting and prevent contact with exposed terminals. The LV test box is

a flexible solution that reduces the risk of potential electrical incidents.

QUT grant-in-aid for attendance in ICIEA conference in Singapore, 2012.

QUT grant-in-aid for attendance in EPE-PEMC conference in Serbia,

2012.

XXX

Statement of Original Authorship

I declare that the work contained in this thesis has not been previously submitted to meet

requirements for an award at this or any other higher education institution. To the best of my

knowledge and belied, the thesis contains no material previously published or written by

another person except where due reference is made.

Pooya Davari Date: 21 Feb 2013

QUT Verified Signature

1

Chapter

1. Introduction

1

Chapter 1

2

1.1. Definition of the Research Problem

High power converters have found their way into many industries. Variety of industries

processes have increased their requirements in power level, economy of scale, efficiency,

new control method, new topologies and using new technology (like new semiconductors).

The incentive to move to higher frequencies is not only given by the reduced sizes of the

passive components, magnetic core losses, but also the application requirements. It is thus

clear that much can be gained from moving to a design that enables operation at very high

frequency. Hence in the recent years, high frequency high power converters have been one of

the most active areas in research [1-7].

Generally, high power converters regarding to their power level can be divided into two

different categories of high instantaneous and high average power converters. High

instantaneous power converters are mostly known as pulsed power supplies. The need for

generating high instantaneous power pulses was started with defence-related applications.

But over the last two decades, non-military applications of pulsed power technology have

been gained lot of interest. More than one hundred possible applications can now be listed [2,

8-43].

Varieties of topologies have been introduced and used for pulsed power supplies such as

Marx generators (MG) [18, 44, 45], pulse forming network (PFN) [18, 45, 46], magnetic

pulse compressors (MPC) [18, 45, 47, 48], and multistage Blumlein lines (MBL) [18, 45].

The main advantage of these methods (non-solid state) is utilizing gas-state and magnetic

switches as they possess a very high blocking voltage and fast rise time [10, 30]. However,

Gas-state switches require special operating conditions such as high pressure, vacuum

equipments and gas supplies. In addition, they are bulky, unreliable, have short lifetime span

and low repetition rate. Even with the magnetic switches, which have a higher repetition rate,

the problems remain. These conditions limit the mobility, efficiency and increase the cost and

the size of the pulsed power system [10, 30].

There have been great improvements in the pulsed power area in the recent years. Advances

in solid-state switches and exploiting power electronics techniques and topologies have led to

compact, efficient, high repetition rate and more reliable pulsed power systems. The main

drawbacks of solid-state switches are their limited power rating and operation speed. The new

developed solid-state switches such as Insulated-Gate Bipolar Transistor (IGBT), Integrated

Gate-Commutated Thyristor (IGCT) and Silicon Carbide Devices (SiC) have high power

rating [30, 49-51], but their lower operation speed comparing with the gas-state switches and

http://en.wikipedia.org/wiki/Insulated-gate_bipolar_transistor

Chapter 1

3

high cost still put limits in the pulsed power supplies. From these circumstances, Problem #1

arises:

1. Problem # 1: Solid-state pulsed power supplies drawbacks including low blocking

voltage of switching device, operating speed, and lack of flexibility

One way to increase the pulsed power supply performance and cover the switch

limits is to explore alternative circuit topologies. Since the above mentioned

topologies carries complexity, inflexibility and inefficiency as their main

drawbacks, applying power electronics topologies and techniques to remedy these

problems was one of the major objectives in this research project. Power electronic

topologies are considered not only as an alternative way to overcome the switch

limits but also in developing flexible and compact systems. Thus, several power

electronics topologies and control methods are studied and investigated in order to

optimize them for supplying pulsed power applications with maximum flexibility.

In addition to switching devices and circuit topology, power converter requirements are

dependent on the characteristics of the load as well. In other word this is the application

which defines the power supply specifications. Pulsed power applications present one of the

most varied ranges of loads in terms of load behavior and impedance. Hence, various load

conditions adversely affect the power supply flexibility which leads to the second research

problem:

Problem # 2: Load dependency problem

To remedy this problem, two different approaches can be studied. The first is to

design a pulsed power supply based on the application characteristics. This requires

clear study of the application or load behaviour, which can be results in a most

appropriate power supply for that specific application. The second method is to

design a flexible pulsed power supply. Here, flexibility stands for load

independency, which means to design a pulsed power supply which can cover wide

range of applications. Hence, to cover both methods varieties of pulsed power

applications were considered in order to study their behaviour and effects on the

pulsed power supply.

Chapter 1

4

The high frequency high average power converters (second category) in this project were

specifically considered for high power piezoelectric transducer applications. The critical

behaviour of a piezoelectric device is encapsulated in its resonance frequencies because of its

maximum transmission performance at these frequencies. An ideal scenario is to have a

sinusoidal excitation signal at the resonant frequency. But at high power and high frequency,

which is the focus area of this thesis, generating pure sinusoidal signals is not possible. The

most efficient way is to benefit from high power high frequency converters to excite the

piezoelectric device properly. This arises the last research problem:

Problem # 3: Increasing the efficiency of high power piezoelectric application

Piezoelectric devices typically have multiple resonant frequencies, but only the

major resonant frequency is generally targeted for excitation in practice. Structural

and environmental changes of a piezoelectric system can affect variations in the

resonant frequencies [5, 52]. Therefore it is important to estimate the main resonant

frequency in order to maintain efficient system operation. Regarding this issue,

piezoelectric behaviour and circuit model was fully investigated in order to optimize

the power supply. In addition, as the performance of the piezoelectric application

increases by adapting the power converter switching frequency with variations in

piezoelectric resonance frequency different frequency estimation methods were

studied.

Chapter 1

5

1.2. Literature Review

1.2.1. Introduction

High power converters have been employed in industries for many years. However, in recent

years, developing and investigating high power converters have gained substantial interest, as

the variety of industries processes have increased their requirements in power level, economy

of scale, efficiency, new control method, new topologies and using new technology. In

addition, interest in operating at higher frequencies has increased as well. The inducement in

moving to higher frequency levels is not only given by the reduced sizes of the passive

components, magnetic core losses, but also the application demands. It is thus clear that much

can be gained from moving to a design that enables operation at high frequency. Hence in the

last decade, high frequency high power converters have been one of the most active areas in

research [1-7].

Since the advantages of employing power electronics technologies have been exploited,

varied types of power converters are introduced and manufactured. The power converters

have been categorized in varied ways regarding to the application requirements and

specifications. One can classified the power converters based on the voltage level, frequency

level or power level. As the main goal of this PhD research was high power converters and

due to the presence of varied applications, in this research high power converters are divided

into two different categories:

High frequency high instantaneous power converters

High frequency high average power converters

High frequency high instantaneous power converters are mostly known as pulsed power

supplies. Pulsed power is rapid release of stored energy as electrical pulses into a load, which

can result in delivery of large amounts of instantaneous power over a short period of time.

The need for generating high instantaneous power pulses was started during the Second

World War for radar application. From that time on, the defence-related applications were

one of the key driving forces behind pulsed power technology, primarily in connection with

nuclear weapons simulation, applications of high-power microwave sources, high-power

laser sources, electromagnetic guns, etc. In addition, over the last two decades, non-military

applications of pulsed power technology have been studied. More than one hundred possible

applications can now be listed [2, 8-43].

High frequency high average power converters (second category) are considered

specifically for high power ultrasound application in this research. One of the major issues

with high power ultrasound systems is the short life-time span of the piezoelectric transducer

Chapter 1

6

which cannot withstand such high power signals for a long time. Recently applying power

electronics techniques for high power ultrasound applications has gained interest, as the life

time of the piezoelectric transducer increases due to the fact that the power supply is not

applying the high power signal continuously and it is in the pulse manner. But the big

challenge is how to design a power supply which can generate a proper excitation signal for

the piezoelectric device. The critical behavior of a piezoelectric device is encapsulated in its

resonance frequencies because of its maximum transmission performance at these

frequencies. The resonance frequencies fall within the range of ultrasound which means

above 20kHz. Hence high frequency high average power converters need to be considered

(second category).

1.2.2. Pulsed Power (high frequency high instantaneous power

converter)

The need for generating high instantaneous power pulses was started during the Second

World War for radar application. From that time on, the defense-related applications were

one of the key driving forces behind pulsed power technology, primarily in connection with

nuclear weapons simulation, applications of high-power microwave sources, high-power

laser sources, electromagnetic guns, etc. In addition, over the last two decades, non-military

applications of pulsed power technology have been studied. Pulsed power supplies are a key

component in systems that the load needs to be pulsed.

Pulsed power is rapid release of stored energy as electrical pulses into a load, which can

result in delivery of large amounts of instantaneous power over a short period of time [44,

45]. Interest in pulsed power technology has been growing extremely fast since 1923, the

year that Erwin Marx invented a very high voltage generator (Marx generator) [18]. Since

then, pulsed power technologies have found variety of applications; especially it is subjected

to the clean technology.

There have been great improvements in the pulsed power area in the recent years. The

highest energy and power that have been achieved in a single pulse are at present of the order

of 100MJ and a few hundred terawatts respectively. The highest voltage and current

amplitudes are obtained at 50MV and 10MA, respectively [44]. In addition to its power and

energy, the pulse rise time, fall time and repetition rate are also important, which are achieved

in a fraction of nanosecond and few MHz. Setting these features is completely depending on

the pulsed power system sections.

Chapter 1

7

Generally, as the Fig. 1.1 shows, a pulsed power system consists of three main sections of

energy storage, pulse generator and the load. Considering this, study and design of a pulsed

power system can be considered at two levels (Fig. 1.1):

System Level

Application level

Fig. 1.1 General scheme of a pulsed power system.

In order to have a high performance system, careful consideration of each of these levels is

needed. Therefore, each level is described in the following sections.

1.2.2.1. System Level

This level consists of two important parts of energy storage and pulse generator. At this

level the main concern is to accumulate and store the energy and finally release the stored

energy rapidly to get required pulses. The whole procedure is accomplished by benefiting

from the different circuit topologies, components and control algorithms.

1.2.2.1.1. Background

Energy storage can be capacitive, inductive [44]. Fig 1.2 illustrated simple schemes of

capacitive and inductive storage pulsed power supplies.

In Fig 1.2(a), first the switch S is open and in this interval energy should be stored from a

high voltage power supply in the capacitor. In the second step the switch turns on and energy

stored in capacitor transfers to the load. In the second scheme, Fig 1.2(b), first S1 is close and

S2 is open. In this stage the inductor is charging. In the second stage, S1 and S2 turn off and

on, respectively. Therefore, the stored energy in the inductor transfers to the load.

As mentioned before, despite the voltage level, there are other pulse characteristics which

are quite important. Fig 1.3, shows the typical pulse wave form that appears across the load.

In this figure three important features of rise time, pulse width and fall time are depicted. The

pulse rise time, which is defined as the time it takes the voltage to rise from 10% to 90%, is

the time that stored energy suddenly released to the load. This means that the voltage across

Chapter 1

8

the load rises from 0 to a certain level in a time interval. Hence, this time interval completely

depends on the switch characteristics. Some applications require a fast rise time (in a fraction

of nanosecond) otherwise the required phenomena never happens and the load may be

damage, such as applying pulsed power to a living cell in biomedical applications [10, 21, 26,

29, 30].

LoadC

S

Pulse Generator

Capacitive

Storage

HV

Power

Supply

(a)

S2

LoadS1

L

Inductive

Storage

Pulse Generator

(b)

Fig. 1.2 Basic Capacitive (a) and inductive (b) schemes for pulse generation.

Fig. ‎1.3 Pulse shape and its parameters.

Chapter 1

9

Pulse energy depends on both its duration, sometimes mentioned as FWHM (Full Width at

Half Maximum), and its amplitude. The pulse duration depends on the amount of the stored

energy and the load impedance. This means that if the stored energy is low regarding the load

impedance then the pulse amplitude should increased so that the required energy transferred

to the load, otherwise the expected phenomenon won’t happen. Therefore, the energy should

be tuned in a way that it satisfies the load demands.

Except for the pulsed power supplies which generate pulses based on the fall time, the load

defines the fall time. In some applications the load has resistive characteristics [18, 38, 53]

that a spark happens at the output while in some applications the capacitive behavior of the

load [35, 41, 53-56] causes the output voltage never decreases and an efficient way to

discharge the power supply is required.

The final feature of the generated pulse is the repetition rate. The repetition rate depends on

storage unit and switches. If the storage unit is not able to recharge in a required time then it

decreases the repetition rate of the system. Same thing happens when the switches are not fast

enough. Normally, when the voltage level increases the repetition rate decreases. That is why

the extra high voltage systems are working in the single shot mode [18].

Regarding the above mentioned features, the most prominent part of a pulsed power system

is the pulse generator, which is based on the utilized switch and topology. Hence, the pulse

generator is the connecting part between the storage and the load. This means that the switch

and employed topology power capability and operating speed define the whole system

characteristics.

Due to switches power rating limits, in some applications it is required to use multiple-

switch base circuit. A conventional way to utilize multiple switches is to directly stack them

in series or in parallel, as shown in Fig 1.4. The series and parallel connections can be used in

generation of high level of voltage and current, respectively [18, 57]. For high power

demands, multiple switches are stacked in series and parallel.

The implementation of series stack requires careful attention. Differences in switches

characteristics and their drive circuit should be considered in a way that they get

synchronized. Moreover, the system has to be able to handle the failure of the switches.

These problems decrease the interest of using this configuration.

Chapter 1

10

LoadC

S2

S3

S1

(a)

LoadC

S2 S3S1

(b)

Fig. 1.4 Using multiple switches to overcome the switches power rating problem, (a) parallel for high current

and (b) series for high voltage applications.

1.2.2.1.2. Non Solid-State Pulsed Power

The simple schemes mentioned above become ineffective if it is required to produce

voltage pulses of amplitude V, for the lack of capacitors and switches designed for

such high voltages. In this case, voltage multiplication schemes are applicable [45]. In this

section the most known non solid-state pulsed power topologies are reviewed.

Marx Generator (MG)

Marx generator is one of the most common topologies used in variety of applications [18,

44, 45] (Fig 1.5). In this circuit, N capacitors are connected in parallel and charged through

resistors to a voltage V0. If all switches close simultaneously, capacitors become connected in

series and a voltage pulse with the voltage level of close to NV0 appears across the load. The

total capacitance value versus the load will be C/N, and hence the pulse FWHM will be

⁄ .

Chapter 1

11

Load

C CS

R

R

CS

R

R

S

R

C

R0

Fig. 1.5 Conventional Marx generator.

This topology was proposed by Marx in 1923 for high-voltage generation based on the

spark gap switches. The spark gap switch can be categorized in gas-state or non solid-state

switches. The high voltage and power rating of this switch is its main advantage. The main

benefit of this generator is that the multiple switches will be synchronized automatically. The

closing of the first switch leads to an overvoltage across the other switches that are not yet

closed. Subsequently, this overvoltage forces them to close.

In the conventional MG the resistors must be high to prevent the discharging of the

capacitors. The high value of the resistance limits the pulse generation rate due to the long

charging time of the capacitors besides being a loss component [44, 45]. To overcome the

drawbacks of energy loss and long charging time, a modified version of the MG has been

introduced, which the resistors are replaced by inductors [45]. The only problem with this

topology is tuning the capacitors charging time due and resonance between the inductors and

the capacitors.

Regarding the above mentioned features, the MG is used extensively in pulsed power

application, and many different Marx based systems have been developed.

Pulse Forming Network (PFN)

A more efficient method of voltage multiplication was proposed by Mesyats [18, 45, 46].

This topology is based on L-C ladder network. This scheme is known as pulse forming

network and it consists of N stages, each containing an oscillatory LC circuit and a spark gap

switch (Fig 1.6). The component values should be as follows:

(1-1)

Chapter 1

12

C0 is the capacitance of the smoothing filter of the rectifier. Resistors are

connected in the circuit to ensure complete discharging of all LC-circuit capacitors. The

resistors are generally selected as:

√

⁄ (1-2)

where i is the LC-circuit number.

V0

C0

L1S1

C1 R1

L2S2

C2 R2

L3S3

C3 R3

LNSN

LoadCN RN

HV

Power

Supply

Fig. 1.6 Conventional pulse forming network [45].

As the switch S1 operates, the capacitor C0 charged to a voltage of V0, discharges into the

capacitor C1. Considering a lossless circuit the voltage across the C1:

[ (

√

)] (1-3)

Considering the above equation, at time √ the maximum voltage V1max is

equal to 2V0. If the switch S2 closes at the time t1, then regarding equation (1-1) C1

discharges into C2 much faster than into C0. In √ , the voltage across the C2

becomes . Therefore, if switch closes at proper time then eventually the

maximum voltage across CN will be:

(1-4)

Therefore, this topology is more efficient as it generates higher voltage level with same

number of switches comparing with MG.

Different combinations of the LC-circuit have been addressed in the literature in order to

increase the performance of the PFN. For example in 1964 just one year after the

conventional PFN has been introduced [18, 45, 58], the modified version of the conventional

PFN was introduced (see Fig. 1.7). The output voltage can reach to the same level of voltage

as MG but the number of the switches is reduced by a factor of 2.

Chapter 1

13

Fig. 1.7 PFN with reversal voltage polarity [18].

Multi-Stage Blumlein Lines (MBL)

The third alternative for voltage multiplication is inductive adder which is also known as

multi-stage Blumlein lines [18, 45]. As shown in Fig 1.8 several pulse transformers with the

secondary side connected in series are used. The advantage of this topology is benefiting

from transformer turns ratio in order to increase the voltage. Secondly, as each core is

independent the currents will naturally be shared between all switches. Therefore, low power

rate switches can be used in this topology. The output voltage will be the sum of all voltages

in the primary side multiply by the transformer turns ratio.

Fig. 1.8 MBL or adder topology [18].

Chapter 1

14

Magnetic Pulse Compressor (MPC)

Magnetic pulse compressor behaviour is similar to PFN because both operate based on the

resonant converter. The only difference is that the MPC is used to compress the generated

pulse to obtain fast rise and fall times. Therefore, it can be used as the final stage in a pulse

generator unit. This topology uses magnetic switches instead of gas-state switches. Figure 9

shows a typical MPC circuit [47]. Suppose that C1 = C2 = C3 = C and MSs L1 MSu. MSu

and MSs refer to the unsaturated inductance and the saturated inductance of the MS,

respectively. On switch closure with the above conditions, the charged energy in C1, which is

initially charged to a potential V0, is transferred to C2 by C−L−C resonance. As the potential

on C2 reaches a point at which MS will saturate, the energy transfer takes place from C2 to C3

once more. In the latter loop, L1 and the switch are replaced by the MS. The voltage V1 and

the current I1 are compressed to V2 and I2, respectively, by the function of MS during the

energy transfer, as shown in Fig. 1.9(b).

Considering the above mentioned topologies variety of pulse generators have been

introduced based on the different combinations of the mentioned topologies [18, 44, 45, 47,

48]. Some of them have been implemented in large scales for extra high power generation

such as Z-Machine, at Sandia National Laboratory, which stores 12MJ of electrical energy in

36 Marx generators and the intermediate stores reach a peak voltage of 5MV. This machine

has been developed now to shoot 27 million amperes (previously 18million) in 95

nanoseconds with output power of 350 terawatts. Improvements on increasing the ability of

machine in order to support 1 petawatts ( ) is going on [59].

The switches in the above mentioned pulse generators are gas-state or magnetic switches

(non-solid state). Gas-state and magnetic switches have been widely used in pulsed power

technology, as they possess a very high electric strength and fast rise time. Gas-state switches

require special operating conditions such as high pressure ( 1), vacuum equipments and gas

supplies. In addition, they are bulky, unreliable, have short lifetime span and low repetition

rate. Even with the magnetic switches, which have a higher repetition rate, the problems

remain. These conditions limit the mobility, efficiency and increase the cost and the size of

the pulsed power system [10, 30, 60].

Chapter 1

15

(a)

(b)

Fig. 1.9 Basic MPC: (a) circuit, (b) Voltage waveforms [47].

1.2.2.1.3. Solid-State Pulsed Power

Variety of topologies has been introduced for solid-state based or combination of solid-state

and non-solid state switches. In this section the well-known ones are mentioned.

The solid-state high power converters have found wide spread applications in industry. The

development of solid-state high power converters started in the mid-1980s when 4500-V gate

turn off (GTO) thyristors became commercially available. The GTO was the standard for the

medium-voltage drive until the advent of high-power insulated gate bipolar transistors

(IGBTs) and gate commutated thyristors (GCTs) in the late 1990s [6]. These switching

devices have rapidly progressed into the main areas of high-power electronics due to their

superior switching characteristics, reduced power losses, ease of gate control, and snubberless

operation.

Solid-state switches are compact, reliable, cost effective, and have a long lifetime and

repetition rate. Moreover, using solid-state switches has the advantage of benefiting from the

power electronic controlling techniques, which results in compact and more efficient pulsed

power supply [10, 25, 30, 40, 44, 53, 60].

Chapter 1

16

The main drawbacks of solid-state switches are their limited power rating and operation

speed. For the applications that needs high repetition rate, high stability, and a long life time

the only options is using the solid-state switches. Therefore, their power rating and operation

speed limits should be solved. Table 1.1 compares the solid-state and non-solid sate switches

together.

Table 1.1 Various switch parameters

Switch Max hold-off voltage (kV) Peak current (kA) Switching speed

Spark gap 100 10 to >1000 <5ns

Pseudo spark 35 5-100 >10ns

Thyratron 30 1-10 >10ns

Thyristor/GTO 1-10 1-80 >1

IGCT 6.5 0.5-3.8 >1

IGBT 6.5 0.1-2 >0.4

MOSFET 1.2 0.05-1 >2ns

Benefiting from Diode

At the beginning of benefiting from solid-state switches in pulsed power, most topologies

introduced for non-solid state pulsed power were used in combination with solid-state

switches. Fig 1.10 (a) shows the MG topology, which instead of using resistors or inductors,

diodes have been used [61]. Using diode resolve the problem of energy loss, resonance and

charging time in MG. Same thing has been done for PFN (LC generator) as depicted in Fig

1.10 (b). The diode here prevents from resonance or oscillations.

Chapter 1

17

(a)

(b)

Fig. 1.10 Benefiting from diodes in MG [61] and PFN [18] topologies.

Solid-State Marx Generator

After the developments in solid-state switches, as mentioned above, in order to benefit from

the long life and high repetition rate the spark gap switches have been replaced by the solid-

state ones. Fig 1.11 shows the Marx generator topology with IGBTs switches [60, 62].

Variety of topologies based on modified version of the MG has been introduced in literature

[48, 60-71].

Fig. ‎1.11 Conventional Marx generator based on solid-state switches [60, 62].

Chapter 1

18

High voltage Magnetic Compression Modulator

Magnetic compression topology has been used widely along with solid state switches. Fig

1.12 shows one of the effective introduced topologies [10, 23].

Fig. ‎1.12 Pulsed power generator based on magnetic pulse compression [10, 23].

The advantages of such a technique are voltage multiplication and pulse compression

without costly re-magnetization techniques. It operates as follows: When switch S1 closes,

capacitor C1 discharges and through transformer T1, charges capacitors C2 and C3. Upon the

saturation of the T1 core, capacitor C2 recharges to the opposite polarity and the voltage

across capacitors C2 and C3 doubles. At this moment, the core of the MS saturates, and the

voltage is applied to load Z. The charge path for capacitor C3 is provided by load Z and

freewheeling diode D. 60 kV peak voltage with 15-20ns rise time pulse have been reported

using the above circuit. The reported result shows the high performance of the introduced

pulsed power supply.

Using Stack of Switches for High Voltage Generation

In solid-state switches the operating speed of the switch has inverse relation with its power

ratings (hold-on voltage and current). Hence, to increase the hold on voltage while preserving

the operating speed the stack of switches can be used. As mentioned before, this solution has

problem in synchronization and switches failure.

Fig 1.13 shows one of the methods based on using stack of switches [72]. Here, a repetitive

pulsed power generator using MOSFET (Metal-Oxide Semiconductor Field-Effect

Transistor) has been developed for accelerator applications. In order to increase the voltage

and current capacity, MOSFETs have been stacked with 8 in series and 6 in parallel. Fig 1.13

shows the circuit of a module of 6 MOSFETs, connected in parallel. MOSFET times, high-

http://www.patentgenius.com/patent/7459806.html

Chapter 1

19

speed ICs are used to control gate potential. In addition, the FETs (Field-Effect Transistor)

are carefully selected so that the response-time difference between them is reduced to

minimum. Finally, eight of such modules are connected in series (48 switches connected).

Fig 1.14 shows the test circuit. In this circuit a capacitor charged by 5kV (functions as the

energy storage), while the MOSFET unit turning on and off controlled by the trigger pulses.

A 70Ω resistor is used as the load. The pulses were generated at 5kV with rise time of 33ns,

fall time of 43ns and repetition rate of 2.1 MHz.

Fig. ‎1.13 Circuit of a module of 6 MOS-FETs, connected in parallel [72].

Fig. ‎1.14 Test circuit of MOS-FET switching unit [72].

Considering the synchronization issue, the manufacturing companies have optimized

semiconductor components for pulsed power applications. Hence, similar switches are

selected to resolve the synchronization problem and in the case of series configuration new

type switches which have SCFM (Short Circuit Failure Mode) are used. Fig 1.15(a) shows

ABB switch [49] for pulsed power applications which can handle 10Kv, 90kA with low

Chapter 1

20

repetition rate of 1Hz. But BEHLKE Company which is one of the companies who has

focused on extra high voltage switches has developed fast switches that can handle 90Kv,

100 A with repetition rate of 100 KHz [73] (Fig 1.15(b)). The only problem with these kinds

of switches their high cost and high rise time.

(a)

(b)

Fig. 1.15 High voltage switches (stack of switches) optimized for pulsed power applications: (a) ABB [49], (b)

BEHLKE [73].

1.2.2.1.4. Power Electronics Converters for Pulsed Power

Power electronic converters have been one of the fastest growing market sectors in the

electronics industry over the last 25 yrs. Power electronic devices are at the heart of many

modern industrial and consumer applications and account for $18 billion per year in direct

sales, with an estimated $570 billion through sales of other products that include power

Chapter 1

21

electronic modules [74]. Recently, applications of solid-state high power converters have

gained great interest.

The topologies that were not originally designed for the pulsed power have been used in

this area. The main reason is benefiting from power electronic techniques to control the

energy flow and increase the system efficiency [53]. In the following the most high

performance methods for pulsed power supplies are reviewed.

An ultra compact generator based on the boost topology using static-induction thyristor (SI-

Thyristor) is addressed in [72]. In this approach, a SI-thyristor is used to develop an ultra-

compact pulsed power generator that is aimed at automobile applications. The electrical

circuit is illustrated in Fig 1.16.

This topology is consists of two stages of inductive energy storage with opening switch.

The behaviour of stage 1 and 2 are controlled by FET1 and FET2, respectively. Although the

second stage is completely controlled by FET2, the SI-thyristor plays the role of opening

switch and holds the voltage during the output. The SI-thyristor used in this experiment is a

4cm device with nominal maximum voltage of 3 kV and current of 300 A. The typical

waveforms are shown in Fig 1.17. The capacitor C1 is charged by the DC voltage supply of

12 V. The capacitance of C1 is large enough so that its voltage keeps nearly constant during

operation. When FET1 is turned off, the inductive energy stored in L1 is transferred to C2,

resulting in the C2 charging voltage of 200 V. When FET2 is turned on, the current in L2 goes

up and reaches 150A. Finally, the opening of SI-thyristor causes a voltage across SI-thyristor

of 3kV, which is multiplied by the transformer to 12 kV at the output. The FWHM is 100ns

and the repetition rate is 2 KHz.

Fig. 1.16 Electrical circuit of the pulsed power generator using SI-thyristor [72].

Chapter 1

22

Fig. 1.17 Waveforms of L2 current and the output voltage [72].

The literature refers the use of flyback, forward, half-bridge and full-bridge converters for

applications such as rapid capacitor high-voltage charging, food processing, X-ray plasma

processing, and air and water pollution control [53, 75-78]. The advantage of all these

configurations is using the transformer which is advantageous in isolation and stepping up the

output voltage.

Flyback and forward converter are two well known topologies in power electronics [53, 76,

78-85]. Flyback topology has been used from long time ago in CRT (Cathode Ray Tube) TVs

for high voltage generation. Fig 1.18 shows these two topologies.

The forward and flyback topologies are suitable for the generation of unipolar pulses. In the

case of bipolar pulses, half-bridge or full-bridge configurations are required (Fig 1.19).

Comparing flyback with all other converters for the pulsed power applications shows that

flyback topology seems much more appropriate [53, 82, 83]. The major advantage of flyback

topology over other power electronic topologies is the way of using transformer. The

transformer in this topology stored energy, steps down reflected voltage across the switch and

isolation. The problem with this topology is high frequency issues due to using the

transformer [83, 86].

Chapter 1

23

(a)

(b)

Fig. 1.18 (a) flyback, and (b) forward topologies [53].

Most of the introduced method for solid-state based configurations didn’t apply to the real

applications the whole circuit has been just evaluated with a resistive load. In the real world

that the applications have different behaviour such as RC, C, or RL [53] these methods

become ineffective. As a conclusion, even with the recent developments on solid-state

switches, the solid-state topologies are still dealing with two important issues comparing with

the non-solid state ones in terms of voltage level and rise time.

Chapter 1

24

(a)

(b)

Fig. 1.19 (a) half bridge, and (b) full bridge [53].

1.2.2.2. Application Level

As mentioned earlier, the last section of a pulsed power system is the load. Pulsed power

applications present one of the most varied ranges of loads in terms of load behaviour and

requirements. Hence, various load conditions adversely affect the power supply performance.

Here, different load types associated with pulsed power systems are described.

1.2.2.2.1. Load Types

Resistive Loads

Pure resistive loads are not common in pulsed power applications; however high impedance

resistive loads are used frequently as a load reference for testing and evaluating different

pulsed power supplies. One of the most challenging load types in pulsed power area is low

impedance resistive loads, in applications such as water decontamination, liquid food

sterilization, and biomedical materials [12, 22, 26-28, 38, 47, 53, 87]. The problem with low

resistive loads is that the energy dissipated in the load before the output voltage reaches to

required voltage level. To overcome this problem designing a load independent pulsed power

supply is needed. Till now applying pulsed power supplies with rise time in the order of

fraction of nano-seconds have found to be a good solution [27, 53].

Chapter 1

25

Resistive Capacitive Loads

Most of the pulsed power applications have resistive-capacitive (RC) behaviour, such as

plasma and gas processing. In fact, one of the most challenging operating conditions is

related with operation with plasmas [39, 53, 54, 56].

A typical example is dielectric barrier discharge (DBD) loads. To understand the load

dynamic characteristics, it can be modelled with an RC electrical equivalent circuit as in Fig

1.20. Here Cg represents the gap capacitance and Cd is the dielectric capacitance. Once that

dielectric breakdown occurs, the displacement current flows from one electrode to the other,

through the tube walls, using the plasma as a conductor, and this current is limited by Cd in a

unique manner [55].There are two regions of operation for a DBD load. Prior to breakdown,

Cg is in series with Cd, and the discharge is off. During breakdown a nonlinear gaseous

discharge is in series with a capacitance Cd, which plays a major role in limiting the gap

current.

Fig. 1.20 Electrical equivalent circuit of a DBD load [55].

In high impedance-capacitive loads, it is extremely important that the power supply has the

capability to short circuit the load after applying high voltage pulse; otherwise the load can be

damaged as the load stays charged to almost full pulse voltage [53].

Inductive Loads

This type of the load presents when voltage pulses are applied to a coil. For instance, in

electromagnetic metal forming application (EMF) high voltage resonant power supply applies

high current pulses (kA) into very low-inductive system [53, 88]. With this type of loads, it is

important that the pulsed power supply clamped the load with an opposite polarity voltage

after high voltage pulse.

Chapter 1

26

1.2.2.2.2. Pulsed Power Applications

The load is actually the application which the pulsed power supply is going to apply to. It is

one of the important parts of the pulsed power system, as the whole system is designed based

on the load or application requirements. Each application is separated based on the generated

pulse characteristics such as pulse amplitude, rise time, full width at half maximum (FWHM),

fall time, etc. Table 1.2 shows the typical pulsed power applications requirements [30].

Table 1.2 Summary of typical pulsed power applications requirements [30].

Application Electrical Energy Peak Power/Pulse

High Energy Density Plasma Physics 20MJ >10’s TW

Intense Electron Beam Radiography 200 kJ <1 TW

High Power Microwave (Narrowband) 10 kJ 100 GW

High Power Microwave (Ultra-wideband) 10 J 10 GW

Ion Beam Modification of Materials <10 kJ 30 GW

Bioelectrics 0.1 mJ- few J 10kW – 100 MW

Generally these applications can be divided into three categories:

Military and Defence Applications

In fact, the development of equipment to drive magnetron oscillators for microwave

radar started the field of pulsed power. Since that time, applications have been

dominated by defence-related technologies [44, 89].


In recent years, several applications based on pulsed power technology have been

introduced, developed or commercialized. Pulsed power has covered a variety of

industrial applications such as, fusion systems, food pasteurization, water treatment,

ozone generation, etc [9-11, 13-15, 18, 19, 22-25, 27, 29-31, 33, 34, 36-43].

Biomedical Applications

Since demonstrated that lightly ionized air is extremely effective in killing harmful

bacteria and spores on contaminated surfaces, the pulsed power became an important

technology in the biomedical arenas. Pulsed power has had a great impact on

biomedical applications such as, cancer treatment, wound healing etc [2, 12, 17, 20, 21,

26, 28, 30, 32, 35].

Chapter 1

27

From another aspect the pulsed power applications can be divided into two different types

of pulse generation. One is generating electro-magnetic energy which is the main concern of

military applications and second is plasma pulsed generation. In the following sections the

major non-military applications of the pulsed power system, in order to show the importance

of this technology, are described.

Plasma

Plasma is defined as the fourth state of matter. It was first identified by Sir William Crookes

in 1879. Plasma is an ionized gas containing free moving charge carriers: electrons and ions.

Over 99% of the visible universe is made up of plasma. For example, the matter in stars or

nebulae is plasma. There are also man-made plasmas on our planet, daily used in industrial

and medical applications. One of the ways in generating plasma is using pulsed power

supplies [35, 54, 56, 90-96].

But why generating plasma is beneficial. Plasma is very effective in disinfection. This

makes plasma very useful for various medical and industrial applications. The factors that

make plasma advantageous are:

Ultraviolet radiation

Ultraviolet radiation (UV) causes DNA (Deoxyribonucleic Acid) damage. This DNA

damage inhibits the replication of bacteria. The wavelengths in the range 220 – 280 nm at the

irradiation intensities of several mW/cm2 are known to have the optimal inactivation effect.

However, studies showed that UV is not an important decontaminating factor in treatments

with low temperature atmospheric air plasmas [35].

Heat

Heat can inactivate bacteria. Conventional sterilization methods are based on heat.

Charged particles

Charged particles from the plasma cause cell membrane charging, which may rupture the

outer membranes of bacteria [11, 21, 35]. This is because the electrostatic force caused by the

build-up of charge on the membrane can overcome its tensile strength and cause rupture.

Reactive species

The reactive species in non-thermal atmospheric air plasmas are generated through electron

impact excitation and dissociation. In these air plasmas we can find nitrogen– and oxygen–

based species such as atomic oxygen, ozone, NOX (Nitric Oxides), and OH.

Chapter 1

28

When the electric field intensity increases, it affects the electrons in the atomic orbital and

may cause atoms or molecules to polarize or liberate electrons. The maximum electric field

that a dielectric material can withstand without conduction is known as the dielectric strength

of that material and is expressed in V/mm. Table 1.3 shows the dielectric strength of common

materials. When the electric field increases beyond the dielectric strength, the material

becomes conducting by a process called avalanche effect, whereby electrons collide with the

atomic or molecular structure, releasing more electrons which in turn lead to the further

breakdown of the material. Large currents are possible at breakdown. For example the

dielectric strength of air at normal temperature and pressure is 3 kV/mm. At this point air

ionizes rapidly and arcing occurs. This is what happens during a lightning strike. Therefore,

depending on the electrodes gap and material dielectric strength the required electric filed

intensity varies in different applications.

Table 1.3 Dielectric strength of common materials.

Substance Dielectric Strength (kV/mm)

Helium (relative to nitrogen) 0.15

Air (relative to nitrogen) 3

Window glass 9.8-13.8

Silicone oil, Mineral oil 10-15

Benzene 163

Polystyrene 19.7

Polyethylene 18.9-21.7

Neoprene rubber 15.7 - 26.7

Distilled Water 65-70

High Vacuum (field emission limited) 20 - 40 (depends on electrode shape)

Fused silica 470 - 670

Waxed paper 40 - 60

PTFE (Teflon, Extruded ) 19.7

PTFE (Teflon, Insulating Film) 60 - 173

Mica 118

Generally, plasma can be divided into three different types of:

1) Plasma Discharge

Plasma discharge, thermal plasma or hot spark happens when the electric field intensity

exceed from the dielectric strength of the matter. In this situation high current pass

http://en.wikipedia.org/wiki/Helium

http://en.wikipedia.org/wiki/Air

http://en.wikipedia.org/wiki/Dielectric_strength

http://en.wikipedia.org/wiki/Glass

http://en.wikipedia.org/wiki/Silicone_oil

http://en.wikipedia.org/wiki/Mineral_oil

http://en.wikipedia.org/wiki/Benzene

http://en.wikipedia.org/wiki/Polystyrene

http://en.wikipedia.org/wiki/Polyethylene

http://en.wikipedia.org/wiki/Neoprene

http://en.wikipedia.org/wiki/Water

http://en.wikipedia.org/wiki/Vacuum

http://en.wikipedia.org/wiki/Fused_silica

http://en.wikipedia.org/wiki/PTFE

http://en.wikipedia.org/wiki/Extrusion

http://en.wikipedia.org/wiki/PTFE

http://en.wikipedia.org/wiki/Mica

Chapter 1

29

through the material and makes spark or arc. Due to this fact this type of plasma is not

preferable as the material, probe and pulsed power supply may get damage.

2) Cold Plasma

Cold plasma can happen in low pressure and it is also kwon as non-thermal plasma.

This situation is readily achieved under reduced pressures, in the range of 10 to 1000

Pa. But, these plasmas must be confined in massive vacuum reactors, their operation is

costly, and the access for observation or sample treatment is limited.

3) Atmospheric non-Thermal Plasma/Corona

In order to achieve the characteristics of non-thermal plasma but at atmospheric

pressure the current should be limited when the material starts to conduct. This can be

done by controlling electric field intensity in way that it never goes above the dielectric

strength. The second way is using a dielectric barrier discharge (DBD) which is a

special way of placing the electrodes [39-41, 53-56]. This kind of plasma in known as

corona discharge [35, 90, 91, 93, 96] and it is quite desirable due its non-destructive

features.

Interest in pulsed power technology in biomedical field is increasing since its disinfection

feature has been found. The biological effects of pulsed power can be categorized as water or

liquid discharge, gas discharge, and electromagnetic field [2, 17, 20, 21, 26, 28, 30, 32, 35].

Except for generating electromagnetic field which can be used in cancer treatment all other

applications benefit from the plasma inactivation features (described in the previous section).

On the other hand, with the pulsed power major features as efficiency, compactness, low

maintenance, reliability and clean technology, pulsed power industrial applications are

growing fast.

In the following two sections, the pulsed power applications have been considered in three

different categories with respect to biological and industrial applications. First one is

biological applications and the second one is industrial applications.

1.2.2.2.3. Water/Liquid Discharge

The most typical application of pulsed discharges in liquid is water treatment. The objects

of treatment are in various fields in both biomedical and industrial applications, such as

sewage, river, lake and etc. Liquid discharges are selected for sterilization or inactivation of

bacteria in liquid decontamination. A streamer discharge is usually used in liquid [9, 10, 12,

Chapter 1

30

22, 27, 38, 47, 87]. These streamer discharges in liquids are able to produce a high electric

field, high energy electrons, ozone, chemically activate species, ultraviolet rays, and shock

waves, which readily sterilize microorganisms and decompose molecules and materials. Fig

1.21 shows a discharge induced in water [87].

Fig. 1.21 Photograph of a discharge induced in water [87].

In addition to the mentioned features of the water discharges, generated ozone due to the

plasma phenomena can be used in ozone therapy. Medical ozone is used in three principal

fields including: treatment of circulatory disorders, disease produced by viruses such as liver

disease (hepatitis) and herpes and healing badly infected topical wounds and inflammatory

processes including open ulcers on the legs, inflammatory intestinal conditions, burns, scalds

and infected wounds as well as fungal infections.

This method is expected to have high efficiency for the treatment and decompose the

pollutants that cannot be easily decomposed by even ozone oxidation. Pulsed power systems

not only are more effective, compact and cost effective and reliable but also by using pulsed

power supplies there is no need to employ chemical approaches for example in

decontaminating water. For example, pulsed power is employed in cleaning of lake and dam

algae bloom by discharges in the water. Fig 1.22 shows the effect of using pulsed power in

algae treatment [27]. Usually algae exist on the surface of the water. As the figure shows after

applying pulsed power the algae were precipitated to the bottom.

Chapter 1

31

Fig. 1.22 Photograph of effect of water discharges in algae treatment [27].

However, liquid discharge also has some problems. One is the high breakdown voltage of

the liquid. Therefore, good insulation is needed inside the pulsed power supply. Secondly, it

is difficult to make a widespread discharge in water for getting a wide treatment area. Finally,

as the liquid, especially water, has small resistance therefore generated pulses should have

either high energy which increases the size of the system, or get connected to the liquid after

the voltage level has reached to the required amplitude.

1.2.2.2.4. Gas Discharge

Gas discharges are selected for sterilization of Escherichia coli and other bacteria for

decontamination of air. It has been used mostly in treatment of exhaust gases but is has also

applied for crop growth and water treatment as well. For crop growth, gas discharges were

used for cultivation of mushrooms [10], and for water treatment gas discharges with droplets

of water is suggested. But as mentioned its main application is on exhaust gas treatment.

Recently, non-thermal plasmas have been increasingly used to control harmful gases and to

generate ozone [13, 15, 19, 24, 33, 34, 36, 37, 39].

The environmental pollution caused by the consumption of fossil fuel energy is increased

rapidly; it became very important to protect the environment and to develop technologies

with less energy consumption and less pollutant exhaust. One the ways is using non-thermal

plasma in exhaust gas treatment. Non-thermal plasmas have many kinds of chemically

activate radicals, such as O, O3, N, N*, and OH, which are generated by the dissociation

Chapter 1

32

and ionization of the ambient gases caused by the impact of energetic electrons. Using pulsed

power technology, non-thermal plasmas have been generated by a pulsed electron beam or a

pulsed streamer discharge, and have been used to treat nitric oxides (NOX), sulfur dioxide

(SO2), carbon dioxide (CO2) and volatile organic compounds (VOCs), and to generate ozone.

Particularly, the treatment of exhaust gases (NOX and SO2) using a pulsed streamer discharge

has been studied for the past decade.

1.2.2.2.5. Material Processing

Plasma-based ion implantation and deposition are one of the well-known methods which

are used for surface treatment of complex shape materials by employing pulsed power

supplies. This method can now be considered a nature technology for surface modification

and thin film deposition. In addition to this method pulsed power can be used in other

applications of material processing, such as surface heating by laser, or microwave

irradiation, synthesis nano-composite powders, and joining of solid materials [10, 88].

1.2.2.2.6. Electromagnetic Field

Pulsed electromagnetic fields are selected for sterilization or inactivation of bacteria, but

they have gained more attention for cancer treatment. The electromagnetic field yields

electroporation of the cell membrane or influences the cell nuclei. Electroporation is usually

used to sterilize bacteria. This technique is commonly applied for sterilization in food

processing, such as inactivation of microorganisms naturally contaminated in orange juice

[97].

Recently, many researchers have interests in using shorter pulse durations with fast rise

time. The reason is that the electrical pulses should reach to the cytoplasm without affecting

the membrane. To make this happen the pulse duration and rise time should be faster than the

charging time of the membrane. To understand the behaviour of biological cell in dealing

with electrical pulses, electrical circuit of a cell is considered [30]. Fig 1.23 shows the

capacitive-resistive model of a biological cell. In addition to rise time, voltage level or the

pulse amplitude should be sufficiently high in order to affect the intracellular organelles [30].

The varied range of applications for pulsed power shows the importance of this technology.

The study and research on this technology is going with respect to designing more compact

and reliable system and studying biomedical features of the generated pulse.

Chapter 1

33

(a)

(b)

Fig. 1.23 a) Biological cell structure, b) Double-shell model of a biological cell [30].

1.2.3. High Frequency High Average Power Converter for

Ultrasound Applications

Wide spread research has been conducted on piezoelectric transducer applications since

1880, the year that Pierre and Jacque Curie discovered the phenomenon of piezoelectricity

[52, 98]. Piezoelectric transducers convert electric power to acoustic power and vice versa,

with most of the applications to date being low-power ones. In the last decade, however,

high-power ultrasound applications have gained significant importance [3, 99-104]. These

types of applications have great potential in chemical and bio-technology processing,

specifically for enhancing chemical reaction kinetics and new reaction pathways. Such

enhancements allow changing the production from batch processing to continuous flow

processing, thereby reducing investment and operational costs.

Improvement of ultrasound systems has very significant environmental implications,

particularly in the area of renewable energy (bio-mass and bio-fuel), waste-water treatment

and biomedical applications. High-power ultrasound technology is already being used to

address these areas of need, but to a very limited extent owing to the very inefficient nature of

the state of the art in energy conversion.

The big challenge is how to design a power supply which can generate a proper excitation

signal for the piezoelectric device. The critical behaviour of a piezoelectric device is

encapsulated in its resonance frequencies because of its maximum transmission performance

at these frequencies. The resonance frequencies fall within the range of ultrasound which

means above 20kHz.

Chapter 1

34

Recently, the most efficient way to generate power signals to excite the piezoelectric device

found to be high frequency high power converters. In order to design a suitable excitation

signal using a power supply, the behaviour of the piezoelectric transducer needs to be

investigated. Hence, designing stage of a power supply for piezoelectric applications falls

within two major steps:

Analyzing piezoelectric characteristics

Design and adapting a high frequency high power supply based on the piezoelectric

behavior

1.2.3.1. Piezoelectric Transducer Characteristics

Large amount of research have been conducted on developing methods to study

piezoelectric resonator. Among them the most effective way is evaluating its impedance

frequency response [105-117]. In the impedance methods, changes in the transducer

impedance due to mechanical resonance are detected. A minimum in the impedance response

corresponds to a resonant frequency, fr (see Fig 1.24). The impedance frequency response is

the ratio of the voltage spectrum to the current spectrum. To calculate the piezoelectric

impedance response, a voltage source needs to be applied to the device as an excitation signal

and current needs to be measured simultaneously. In order to obtain the response of the

device for a specific range of frequencies, the excitation signal should cover the entire

frequency range.

Piezoelectric devices typically have two kinds of electrical resonances (see Fig 1.24), the

first known as a resonance and the other one known as anti-resonance [52]. In the last few

years, varieties of approaches have been introduced for obtaining piezoelectric impedance

characteristics [105-117]. A knowledge of such impedance characteristics is important for,

structural health monitoring (SHM) [114, 118] and for electrical circuit modelling [106, 107,

113, 115] of the piezoelectric devices. In the case of high power applications, knowledge of

the impedance characteristics is important so that the maximum transmission performance

can be achieved [5, 115-119]. In the other words, at resonance frequencies, piezoelectric

elements convert the input electrical energy into mechanical energy most efficiently, and so it

is necessary to accurately know these frequencies and excite the system appropriately.

Chapter 1

35

Fig. 1.24 Resonance and anti-resonance frequencies in piezoelectric impedance frequency response.

Piezoelectric devices typically have multiple resonance frequencies (as illustrated in Fig

1.25) but only the major resonance frequency is generally targeted for excitation in practice.

Structural and environmental changes can affect the transducer elements and cause changes

in the values of the resonance frequencies [5, 52]. As such changes are occurring, it is

important to estimate the main resonance frequency if one is to maintain efficient system

operation (see Fig 1.25 for an illustration).

To measure piezoelectric transducer impedance, generally a network analyser is used.

However, due to the fact that the network analyser is expensive, bulky and it can’t operate at

high power other methods can be employed. In addition to network analyser, these methods

can be categorized as below:

Traditional method: In this method several single frequency sine wave signal

regarding to the range of frequency interest are applied to the transducer separately.

The impedance can be estimated after the responses are captured for each single

frequency. The accuracy of this method depends on the frequency resolution. This

method is not practical for studying the response of the transducer for wide range of

frequency as it is labour intensive.

The second method is based on exciting the piezoelectric transducer using a broad-

band signal which covers wide range of frequencies. The concept of this method is

based on the system identification which is well-known in the field of signal

processing [120, 121]. This method is based on the impulse response estimation and

can utilize signals such as chirp, white noise, step, and etc [120, 121]. The

advantage of this method is low computational complexity and high accuracy.

Chapter 1

36

It is to be noted that for high power transducer it is not possible to study the characteristics

of the transducer at high power using a network analyser, due to the fact that a network

analyser operates at low power. To remedy this issue the aforementioned methods should be

employed.

Fig. 1.25 Depiction of typical resonance frequency variations of piezoelectric devices due to structural changes.

In addition to impedance frequency response, a good understanding on the electrical

characteristics of piezoelectric transducer is essential in the design and analysis of sensory

systems relying on piezoelectricity, such as impedance-based SHM systems [107, 115].

Usually, an equivalent circuit model is used to describe the electrical characteristics of the

piezoelectric materials. Moreover, an equivalent circuit model makes piezoelectric-based

systems simulation possible.

The most basic equivalent electrical circuit model characterizing piezoelectric transducer is

the Van Dyke Model [107] (see Fig 1.26). The Van Dyke Model is a parallel connection of a

series RLC representing mechanical damping, mass, and elastic compliance and a capacitor

representing the electrostatic capacitance between the two parallel plates of the PZT

(Piezoelectric Transducer) patch [122].

Fig. 1.26 The Van Dyke Model [107].

As the Van Dyke model cannot accurately model the piezoelectric behaviour other models

have been proposed. The Guan Model, which is the most recently proposed, estimates values

Chapter 1

37

of the electrical components based on the electrical behaviour of the piezoelectric ceramics

[107, 115]. The Guan model, as shown in Fig. 1.27, adds a series resistor Rs and a parallel

resistor Rp to C0 of the Van Dyke Model to account for the energy dissipation. Determination

of the values of the electric components C0, C1, L1 and R1 relies on visual inspection on the

magnitude and phase of the impedance, and values of Rs and Rp are decided by the amount of

energy dissipation [107].

Fig. 1.27 The Guan model-unloaded piezoelectric [107].

When a piezoelectric ceramic is mounted to a mechanical structure, the mechanical

boundary conditions of the piezoelectric ceramic change [107, 115], and, accordingly, a

different circuit model is required for a loaded piezoelectric ceramic. Since a loaded

piezoelectric ceramic experiences multiple resonances, a circuit model for a wide frequency

range with multiple resonant frequencies can be employed to model the behaviour of a loaded

piezoelectric ceramic. As shown in Fig 1.28, additional R-C-L branches are added in parallel

to the R1-C1-L1 branch of the Van Dyke Model [107, 122]. The Guan Model is also extended

to accommodate a loaded piezoelectric ceramic based on the extended Van Dyke Model, as

shown in Fig. 1.29. Each series RLC branch physically stands for a mechanical resonant

mode (as shown in Fig 1.25).

Fig. 1.28 The extended Van Dyke model-loaded piezoelectric [107].

Chapter 1

38

Fig. 1.29 The complete Guan model-loaded piezoelectric [107].

Estimating an accurate model representing the piezoelectric transducer behaviour such as

linear and nonlinear characteristics and a systematic procedure to find an equivalent electrical

circuit model is an undergoing research due to the varied behaviour of the piezoelectric

device under different situation.

1.2.3.2. Power Electronics Topologies for High Power Ultrasound

Systems

As mentioned above, the critical behaviour of a piezoelectric device is encapsulated in its

resonance frequencies because of its maximum transmission performance at these frequencies

[52, 98, 114, 116, 119]. Hence, ideal scenario is to have a sinusoidal excitation signal at the

resonant frequency. But at high power and high voltage, which is the focus area of this paper,

generating pure sinusoidal signals is not possible.

The most efficient way to generate power signals is to use power electronics inverter

topologies. Specifically, switch mode inverters are used for piezoelectric high power

applications due to their high power density, efficiency, low cost and size compared to

conventional linear power supplies [5, 102, 123]. However, the harmonics present in the

output waveform produce undesired side bands which are not suitable in many applications.

Moreover, they also cause unnecessary power dissipation which reduces the efficiency of the

power converter [124, 125].

In the last decade several power electronics topologies have been introduced for high power

piezoelectric applications. To overcome the harmonic issue, generally either a filtering circuit

or a harmonic elimination technique is employed [125]. Here the most important topologies

recently introduced are briefly described.

Chapter 1

39

Two Level Inverters

Generally, with this type of inverters a resonant converter is employed. Resonant converter,

normally, is based on a typical power electronics topology and a resonant stage formed by

applying filter component and the piezoelectric transducer itself (as it has RLC

characteristics).

The first stage can be based on flyback, full-bridge, half-bridge, and etc. Fig 1.30 depicted a

flyback based resonant inverter. The flyback converter is supplied with a full-bridge rectifier.

This stage in addition to providing smoothed dc-link voltage performs a PFC (Power Factor

Correction) function by shaping the input current to be sinusoidal and in phase with the ac

input voltage [126]. The second stage is a high frequency resonant inverter.

Another alternative is depicted in Fig 1.31 using a full-bridge (H-bridge) inverter combined

with a resonant stage [5]. As can be seen different resonant configurations can be used which

totally depends on the piezoelectric transducer and the range of the operating frequency.

Fig. 1.30 Two-stage resonant inverter based on flyback topology [126].

Chapter 1

40

Fig. 1.31 Two-stage resonant inverter based on H-bridge topology [5].

Multilevel Inverter

Employing multilevel inverters is a well-known method in order to increase the generated

signal quality and reduce number of the harmonics close to the fundamental frequency.

Moreover, for ultrasound applications which operate at high fundamental frequency power

converters based on a PWM (Pulse Width Modulation) strategy with high switching

frequency index (fs/fo) cannot be a practical solution. In such applications, the maximum

possible fundamental frequency of a converter is restricted by the switching transients of each

power switching event and the number of switching events per cycle.

In this regard, to reduce the number of switching transients and eliminate the undesired

harmonics, the most effective way is to use multilevel inverters [127-130]. Through these

inverters, it is possible to produce quasi-sine waves with low total harmonic distortion at high

power. Multi-level inverters can increase the quality and efficiency of the high voltage supply

compared to the conventional 2-level inverters. This permits the semiconductor devices to

operate at lower switching frequencies with higher efficiency as well as lower voltage stress

across switches and loads which minimize electromagnetic and ultrasound noise emissions.

Chapter 1

41

Fig 1.32 depicts typical voltage waveforms and their harmonic spectra. It can be seen that the

harmonics are depressed for multilevel inverters such that they can be easily filtered out from

the frequency range of interest.

Fig. 1.32 Effect of using multi-level waveform in harmonic elimination.

Fig 1.33 illustrated one of the multilevel methods introduced for high power piezoelectric

excitation. The employed output filter is resonant LLCC type, which can reduce the total

harmonic distortion (THD). By combining a PWM controlled inverter and resonant filter, the

generated voltage can be varied in a suitable range of frequency. The inverter topology is a

hybrid three-level PWM inverter based on diode clamped topology [102]. Fig 1.34 shows a

typical generated output voltage and current and the output voltage across the piezoelectric.

Chapter 1

42

Fig. 1.33 Three-level PWM inverter combined with LLCC-filter for driving high power piezoelectric transducer

[102].

Fig. 1.34 Typical output voltage and current of employed hybrid three-level inverter when applying to a

piezoelectric transducer [102].

As can be seen even by using a multilevel topology filtering is still employed. However, as

the advantage of multilevel inverter is to push the high frequency contents far away, when

using filters for attenuating the remaining harmonics the filter cost and size decreases [127,

130].

Although several improvements have been achieved in this area, but increasing the number

of the voltage levels with low circuit complexity, adapting with resonance frequency

Chapter 1

43

variations, and employing effective harmonic elimination techniques are still under

investigation.

1.2.3.3. High Power Piezoelectric Transducer Applications

To show the importance of high power piezoelectric transducer, in this section a brief review

on high power piezoelectric applications is provided. Applications of high-intensity

ultrasonics are those wherein the ultrasonic energy is used for producing some permanent

physical change in the treated medium [3]. These effects can be attributed to various

mechanisms, the most important of which are: radiation pressure, heat, streaming,

cavitations, agitation, interface instabilities and friction mechanical rupture. These

mechanisms are involved in a wide range of applications, such as machining, welding,

metal forming, powder densification, etc, in solids; cleaning, particle agglomeration and

flocculation, liquid atomization, drying and dewatering, degassing, etc., in fluids [3]. For

instance Fig 1.35 shows the application of high power ultrasonic atomizer in manufacturing

polymer powder [5].

Recently, high power piezoelectric applications have gained interests due to new

improvements in high power supplies. High-power piezoelectric devices, such as ultrasonic

motors and piezoelectric transformers, are being intensively investigated [3, 99-104, 131].

Because piezoelectric devices often have significant advantages over conventional

electromagnetic devices at smaller sizes, miniaturization of piezoelectric devices is attractive.

Multilayer structures offer better generative force, electromechanical coupling, and energy

density properties than other configurations such as bimorph [103, 132, 133] and moonie

structures [103, 134]; thus they are promising designs for compact devices [103, 135].

Applications including ultrasonic motor, transformers and medical ultrasonic, such as

acoustic radiation force impulse (ARFI) imaging and ultrasound-guided HIFU (High

Intensity Focused Ultrasound) therapy are now shifting to commercialization stage. Although

great improvements have been achieved in this area, but high power applications are still

limited. This limitation causes by restrictions on excitation signal (power level and quality)

and internal power dissipation which is related to the piezoelectric material.

Chapter 1

44

Fig. 1.35 Basis of the ultrasonic atomizer [5].

Chapter 1

45

1.3. Account of Research Progress Linking the Research Papers

1.3.1. Introduction

This research commenced with studying high instantaneous power supplies, generally known

as pulsed power. Varieties of previous introduced methods, which mainly include non-solid

state techniques, were investigated in order to understand their advantages and drawbacks.

Meanwhile, high average power supplies with specific intention on high power piezoelectric

device applications were also investigated. In both categories, the main aim was to employ

and adapt power electronics techniques to increase the system performance, compactness,

simplification, and flexibility based on the application characteristics.

The preliminary literature review emphasised on trends for utilizing power electronics

techniques and topologies in high frequency high power supplies. Hence, studying and

reviewing converters and topologies capable of providing high power at high frequency was

considered as the initial step. Following that, to design a suitable power supply, load and

application requirements and their electrical characteristics were investigated.

Simulation and practical implementation of varied power converters and exciting different

loads at varied operating points were conducted after the literature review stage. Dealing with

practical issues such as noise, EMI (Electro Magnetic Interference), high voltage insulation,

transformer saturation, spikes across the IGBTs, DSC (Digital Signal Controller)

programming and etc, caused vast studying in addition to the main objectives.

1.3.2. Pulsed Power Systems

This research started with an extensive study on non-solid state topologies, such as Marx

generators (MG) [18, 44, 45], pulse forming network (PFN) [18, 45, 46], magnetic pulse

compressors (MPC) [18, 45, 47, 48] and multistage Blumlein lines (MBL) [18, 45], with

respect to their advantages and drawbacks. These techniques possess a very high electric

strength and fast rise time. But they are bulky, unreliable, have short life time span, low

repetition rate, and etc. On the other hand, solid-state techniques are compact, reliable, cost

effective, and have a long lifetime and repetition rate. As the power electronics techniques

are solid-state based, applying power electronics techniques found to be suitable way to

remedy aforementioned issues existing with the non-solid state power supplies.

Recently, significant advances in solid-state switches (both in peak power and operation

speed) and exploiting power electronics techniques and topologies have led to compact and

Chapter 1

46

more reliable pulsed power systems. However, power electronics techniques still show two

main disadvantages of limited power ratings and limited operating speed comparing with the

non-solid state ones. Hence, the major concern was to improve these two important limits.

Moreover, as the power supply is designed based on the load requirements, the load and

application specifications needed to be taking into account along with the two

aforementioned drawbacks.

Regarding to the mentioned issues, the main objectives of studying, designing and practical

implementation of new alternative topologies and configurations of pulsed power supplies

were as followings:

1- Limited power ratings of the solid-state switches

2- Limited operating speed

3- Load/application characteristics

1.3.2.1. Benefiting from Parallel and Series Configurations of Flyback

Converter for Pulsed Power Applications

Interest in applying power electronics topologies and techniques to increase pulsed power

supply flexibility and reliability is growing fast. In the last decade, research and studies

designate the advantage of using power converters in pulsed power applications [10, 136,

137]. However, generation of extra high voltages is still problematic due to components hold-

on voltage limits (such as capacitors, solid-state switches and etc).

One solution is to combine pulsed power supplies. Variety of converter configurations have

been introduced for improving the power supply specifications such as output ripple, high

input voltage, output power ratings, etc. Parallel and series connections are one of the well

known combinations. Designing power supplies based on series or parallel connection are

widely employed for different applications.

Therefore, the main contribution of this part was to benefit from parallel and series

connections of flyback converter modules to develop power rating and rise time of the pulsed

power supply using conventional low voltage switches.

The proposed approach utilizes flyback converter (see Fig 1.36) as it is simple, has only one

switch and magnetic component, is able to generate high voltage, can provide multiple

outputs, isolation, etc [82, 138]. But when it comes to pulsed power applications, in addition

to the above-mentioned features it shows more advantages as follows:

Chapter 1

47

The transformer, in addition to electrical isolation and energy storage also steps down the

reflected voltage across the switch; therefore lower voltage rate switches are needed

unlike other topologies.

Fault tolerant, as the switch is in the off-state during the output pulse.

High voltage output with low input DC voltage.

As the pulsed power applications mostly have R-C characteristic, a current source

topology (such as flyback) is a suitable candidate.

Suitable for controlling the energy flow as it acts as both current source and voltage

source.

Lm

Vs

C0 Load

1 : n

Ll

Vo

+

-

Fig. 1.36 Flyback converter circuit with transformer equivalent circuit model.

1.3.2.1.1. Proposed Method

The proposed method which employs the advantage of parallel and series configuration of

the flyback converter as the pulsed power supply to increase the voltage level and the rise-

time is shown in Fig 1.37. As depicted the parallel and series connections are only considered

for the secondary side, while the primary side of both configurations are connected in

parallel.

Chapter 1

48

Ll

Lm

Vs

1 : n

Co2

Ll

Lm

n : 1

Co1Load

Vo

+

-

Converter (1)Converter (2)

(a)

Ll

Lm

Vs

1 : n

Co1

Load

Ll

Lm

1 : n

Co2

Vo

+

-

Converter (2)

Converter (1)

(b)

Fig. 1.37 Flyback converter series and parallel connection (secondary side), a) Parallel, b) Series.

CoIo Vo

+

-

Load

Fig. 1.38 Energy transmission in a single module flyback converter.

Chapter 1

49

To understand and compare the parallel and series configurations, first a single module

characteristic is described. Regarding to the fact that a flyback converter can operate as a

current source, circuit schematic shown in Fig 1.36 is simplified as in Fig 1.38. Comparing

these two figures, the estimation of the output voltage can be summarized as below:

First, the magnetizing inductance is charged by the DC power supply. This mode lasts

depending on the duty cycle of the PWM (Pulse Width Modulation) signal. The relationship

between Lm and Vs can be expressed as:

s

mm

s

mmsms

TD

IL

TD

ILV

t

iLV

.0.

0

(1-5)

where Vs is the dc supply voltage, Im is the maximum current, D is the duty cycle and Ts is

the switching period. The charged energy stored in the inductor Lm under ideal situation can

be expressed as:

2

2

1mmpri ILE (1-6)

Regarding (1-5) and (1-6) it is possible to control the energy flow in each pulse by limiting

the Im which can be done by varying the duty cycle or the input voltage. This stored energy

will be transferred to the output capacitor when the switch is in the off-state. The capacitive

stored energy is:

2sec

2

1ooVCE (1-7)

If a lossless system considered, then the stored energy in the primary side, Epri, is equal to

the stored energy in the secondary side, Esec. Therefore, the output voltage can be derived

based on equations (1-6) and (1-7) as:

o

mmo

C

LIV

(1-8)

The rate of rise for the generated voltage is defined as dv/dt. When the switch is turned off,

the magnetizing inductance current is at its peak value (Im). Since the current through a

capacitor is proportional to the time-rate of the stored voltage, the rate of rise is:

o

m

o

o

nC

I

C

I

dt

dv

(1-9)

Chapter 1

50

This equation is valid till the current through the capacitor is approximately constant;

otherwise the rate of rise completely depends on the resonant frequency.

The idea of parallel and series configurations of the pulsed power modules is inspired by

both equations (1-8) and (1-9), as these equations show the effect of stored current level,

magnetising inductance and the output capacitor on the generated output voltage and the rise-

time.

Considering Fig 1.37(a) and the fact that each transformer acts as a current source, the

stored energy in the primary side is doubled when two modules are connected in parallel.

This stored energy will be transferred to the output capacitors; hence considering (1-6) and

(1-7) the output voltage magnitude and its rate of rise are as follows:

op

mmparallelo

C

LIV 2

(1-10)

op

o

Parallel C

I

dt

dv 2

(1-11)

where Cop=Co1+Co2. Equations (1-10) and (1-11) indicate the importance of the output

capacitor. Even if the modules are paralleled, the output capacitance can affect the output

voltage level and rise-time and can keep the performance same as a single module. Therefore,

the parallel configuration is beneficial in pulsed power applications when Cop=Co. This

happens in the case of R-C load when the load capacitance is much bigger than the power

supply output capacitance.

Second alternative is series connection of the pulsed power modules, as illustrated in Fig

1.37(b). Here, the injected energy to the output is also doubled. But as described, the

generated voltage features also depend on the output capacitance. Taking into account the

equations (1-6) to (1-9), the output voltage magnitude and its rate of rise can be expressed as:

os

mmSerieso

C

LIV 2

(1-12)

os

o

Series C

I

dt

dv

(1-13)

Chapter 1

51

where Cos is o2o1

o2o1

CC

CC

. In series modules, same as in parallel modules, the output voltage

level and rise time improve as the output capacitance decreases. Here the series modules are

beneficial in both level of voltage and its rate of rise when Cos<Co.

In pulsed power applications two different cases may occur. One is when the load

capacitance is lower than the output capacitance of the pulsed power supply. In this situation

series modules exhibit better performance as illustrated in Table 1.4. The second case is when

the load has much higher capacitance, therefore all series, parallel and single modules have

same output capacitance. Under this circumstance the parallel connection has better

performance (see Table 1.5).

Hence, depending on the load capacitance one of the modules is beneficial to use.

Regarding the voltage level and rate of rise as presented in Table 1.4 and 1.5, it is possible to

generate wide range of voltage levels and improve dv/dt by connecting N modules in series or

parallel. This feature increases the flexibility of the pulsed power supply in varied

applications.

Table 1.4 Output voltage and rate of rise when Co1=Co2=… CoN=Co.

N Modules connected in oV

dt

dv

Parallel Singleo

o

mm V

C

LI

Singleo

o

dt

dv

C

I

Series Singleo

o

mm NV

C

LNI

Singleo

o

dt

dvN

C

IN

Table 1.5 Output voltage and rate of rise when Cop=Cos=Co.

N Modules

connected in

oV

dt

dv

Parallel Singleoo

mm VN

C

LIN

Singleo

o

dt

dvN

C

IN

Series Singleo

o

mm VN

C

LIN

Singleo

o

dt

dv

C

I

Chapter 1

52

In addition, regarding (1-5) and the calculated output voltages in Table 1.4 and 1.5, the

output voltage can be easily adjusted for the required level whether by Vin or D. For example

(1-14) shows the output voltage of series module based on the results presented in Table 1.5

as:

.TDCL

VNV

om

ino

(1-14)

Usually this adjustment is performed by limiting the current by selecting suitable duty cycle

(D). But Im should be calculated in a way that Im<Isat, where Isat is the current saturation level

of the transformer.

As Fig 1.37(b) shows, in the series connection each switch withstands its own module

reflected output voltage, while in the parallel connection; each switch should tolerate the

whole reflected output voltage. This fact makes the series connected modules more

appropriate for generating high voltage waveforms using low voltage switches. Therefore, the

reflected voltage across the switch for N-series module can be rewritten as below:

maxmax1

)( osswitch vNn

Vv (1-15)

By comparing Fig 1.37(a) and 1.37(b) it is obvious that the idea of parallel and series

connections of the flyback modules is applied to the secondary side, while the primary sides

for both cases are paralleled. The main advantage of paralleling the primary side of the pulsed

power modules are: current sharing so the proposed idea is applicable for high power

applications, reducing the number of the power supplies to one at the input side and ability to

employ low current rating switches.

1.3.2.1.2. Operating Conditions

The above mentioned calculations are correct if three important points are considered in the

designing procedure of the pulsed power modules. These are:

1) The output capacitor should be completely discharged during each period; otherwise

the whole stored energy in the magnetising inductor will not transfer to the capacitor.

2) Synchronization of each switch gate signal is required. Delays between each module

gate signal reduce the performance of the system. This is important when higher

number of modules is used.

3) The damping factor (ζ):

Chapter 1

53

As shown in Fig 1.38, the entire circuit acts as a parallel RLC circuit (the current source is

an inductor). Hence, the output voltage and the rise time are the results of the resonance

effect of this circuit. The output voltage can be expressed as:

tstso eAeAtV 21

21)( (1-16)

where s1 and s2 are given as:

20

22,1 S

(1-17)

here, RC2

1 is known as neper frequency and

LC

10 is the resonance frequency.

0)0( to

V and oItCI

)0( are the primary conditions. I0 is the stored current in the

magnetizing inductor (Im=nIo) transferred to the secondary side. The coefficients A1 and A2

are determined by the primary conditions as:

02

02

20

212

LIAA

(1-18)

Now depending on the damping factor coefficient can have two different values:

Underdamped ζ <1 or α <w0 , C

L

R2

1

C

LIA

2

01

(1-19)

Overdamped ζ >1 or α >w0

01 RIA

(1-20)

Equation (1-19) shows that if the circuit is underdamped, the generated voltage amplitude is

independent of the load impedance, which is the situation in which the proposed method is

valid. But under overdamped condition as in (1-20), the output voltage amplitude depends on

the load impedance and the initial current. Hence, in overdamped condition, the parallel

connection has better performance.

In Fig 1.39, the effect of damping factor on the output voltage and rise time is illustrated. A

decreasing damping factor results in a higher voltage level and faster rise time.

Chapter 1

54


Considering the above mentioned features, this effect should be considered especially in

low impedance applications such as the liquid discharge. Therefore, regarding the mentioned

characteristics, the proposed method is more beneficial for high impedance-capacitive load

such as DBD (Dielectric Barrier Discharge) loads.

1.3.2.1.3. Simulation Results

To verify the proposed method and the theoretical analysis, simulations under different

conditions were carried out. Here the performance of parallel and series flyback configurations

(Fig 1.37) is compared with a single flyback converter (Fig. 1.36). The software that was used

for simulation was Matlab 7.10. Table 1.6 shows the parameters value used in simulation

corresponding to Fig 1.36. In order to make the simulation results more close to reality, the

transformer core and copper losses (Rc and Rw) are also considered.

To evaluate the proposed method in developing the features of a single module pulsed

power supply, four distinct cases are considered. Case 1 and 2 have been conducted regarding

the conditions presented in Table 1.4 and 1.5, respectively. In these cases, to simulate a plasma

phenomenon, the load gets connected to the output after the output voltage reaches to a certain

voltage level (threshold) [137]. The different parameters in each case are indicated in Table

1.7.

Chapter 1

55

Table 1.6 Simulation Parameters

Vs

(v) Fs

(kHz) D

(%) Lm

(µH) Llp

(µH) Lls

(mH) Rc

(kΩ) Rw1,2

(Ω) n

10 1 8.9 160 2 4 1 0.1 10

Table 1.7 Simulation Parameters in Case 1 and 2

Parameters Value Case1 Case2

Single Parallel Series Single Parallel Series

Threshold 1000v 1000v 1978v 1000v 1410v 1410v

Co 4nF 8nF 2nF 4nF 4nF 4nF

Co1=Co2 4nF 4nF 2nF 8nF

Case1

In this case the performance of parallel and series connections of two identical power

supplies is compared with the single one. Therefore, in this case the output capacitor of each

module is equal (see Table 1.7). Fig 1.40 shows the obtained output voltage waveform. As can

be seen in the series connection, both voltage level and rate of rise are increased with the

factor of N, (N=2 is considered here).

Case2

If the load capacitance, which is paralleled with the power supply output, is much bigger

than the power supply output capacitance, then the entire single, parallel and series modules

will have the same equivalent output capacitance. In this case the output capacitance of all

modules is considered to be equal to 4nF (see Table 1.7). As depicted in Fig. 1.41, both

parallel and series modules have same voltage level but the parallel connection shows better

performance in the case of dv/dt.

Fig. 1.40 Output voltage waveform of single, parallel and series modules in Case1 (Co1=Co2=Co).

Chapter 1

56

Case3

In the case of time delay between gate drives, here the same situation as Case 2 is selected

but 0.2ms delay is considered between gate signals. Fig 1.42 shows the generated output

voltage for parallel and series modules. As can be seen the output amplitude never reached to

the threshold value. Therefore the load is not connected, so the capacitors are not completely

discharged. This creates another problem since the entire energy in the magnetizing inductor is

not fully transferred to the capacitor and the mentioned equations for estimating the output

voltage are not valid.

Case4

To consider the effect of low impedance load such as liquid discharge applications, in this

case series and parallel modules are considered, same as in Case 1, but RL=100Ω is directly

connected to the output of the modules. In such situations damping factor plays important role

in system performance. The selected resistive load puts all three power supplies in the

overdamped condition. As Fig 1.43 illustrates the parallel modules show better performance,

while series module had far better performance in Case1 when the system was underdamped.

The easiest way to overcome this problem, considering ζ in (1-19), is increasing the output

capacitance.

Fig. 1.41 Output voltage waveform of single, parallel and series modules in Case2 (Cop=Cos=Co).

Fig. 1.42 Output voltage waveform of single, parallel and series modules with 0.2ms delay between modules gate signals.

Chapter 1

57

Fig. 1.43 The effect of the load and the damping factor on the system performance.

Experimentation

Two laboratory prototypes for parallel and series connections, based on single module

flyback converter are implemented, to investigate performance of the proposed method

practically. Fig 1.44 shows the experimental hardware setup for two flyback modules.

Here 600V IGBT modules, SK25GB065, are used as power switches. Semikron Skyper 32-

pro gate drive modules are utilized to drive the IGBTs and provide the necessary isolation

between the switching-signal ground and the power ground. Four 1000 V diodes, STTH3010,

are connected in series for each module. A Texas Instrument TMSF28335 DSC (Digital

Signal Controller) is used for PWM signal generation. Two step-up transformer with an

UU100 core 3C90 grade material ferrite from Ferroxcube, are designed with N1 = 4 and N2 =

40. Magnetizing inductance and leakage inductance of each transformer is approximately

equal to 152µH and 1.6µH, respectively. Here a 5nF capacitor is placed across the switch to

damp the voltage spikes caused by leakage inductance. The circuit is implemented with Vs =

17V, fs = 1 kHz, 15% duty cycle and resistive load of 20kΩ.

Fig 1.45 depicted the measured results. Fig 1.45(a) shows the current sharing at the primary

sides. Due to parallel connection of the primary sides, in both parallel and series

configurations, the current is approximately equally shared between the two modules. As

mentioned before, parallel connection at the primary sides makes the proposed method

independent from using high current rate switches.

Chapter 1

58

The voltage pulses, in Fig 1.45(b), illustrate the better performance of series connection

over the other connections in both voltage level and rate of rise. In series modules the

maximum voltage level and rate of rise are 4.02 kV and 608 V/µs, respectively. While the

parallel and single modules achieved approximately same level of the voltage (2.37 kV) and

rate of rise (304 V/µs). As can be seen, the series modules performance is around 1.7 and 2

times better than the other modules in voltage level and rate of rise, respectively. From the

aforementioned theoretical analysis the performance of the series modules in voltage level

expected to be 2 times better, this difference is due to the resonance and dissipation

happening in the practical circuit. As the rate of charging is in terms of a time constant RC;

hence, as illustrated in the figure, series connection discharging time is shorter than the

parallel and single modules.

Fig. 1.44 Hardware setup for two flyback modules.

Chapter 1

59

Module1 primary current


(a)

(b)

Fig. 1.45 (a) current sharing at the primary side for both parallel and series connections, (b) output voltage

waveform of single, parallel and series modules in Case1 (Co1=Co2=Co).

Chapter 1

60

In this research the advantages of parallel and series configuration of flyback converters for

pulsed power applications are demonstrated. The proposed method aims at increasing the

voltage level and rise time of the generated pulses. Regarding to the above mentioned

evaluations by employing parallel and series connections it is possible to generate wide range

of voltage levels with improved rate of rise.

The extended analysis of the proposed method is presented in Chapter 2 and 3. The basic of

this work along with the simulation results were published and presented as a conference

paper entitled “Parallel and Series Configurations of Flyback Converter for Pulsed Power

Applications” at The 7th

IEEE Conference on Industrial Electronics and Applications in July

2012 in Singapore [139]. The hardware implementation and experimental results are

presented as a part of a journal paper entitled “High Voltage Modular Power Supply Using

Parallel and Series Configurations of Flyback Converter for Pulsed Power Applications”

published in IEEE Transaction on Plasma Science Journal, Oct 2012 [140].

1.3.2.2. Flexible and modular pulsed power supply

Pulsed power applications present one of the most varied ranges of loads in terms of load

behaviour and impedance. Hence, various load conditions adversely affect the power supply

flexibility. Regarding this, it is desirable to have flexible pulsed power supply that can cover

varied range of applications. This flexibility comes in terms of: adjustable output voltage,

controlling the flow of energy and repetition rate. Moreover, the generated voltage pulse

waveform must be load independent [53]. This is quite important especially in the low

impedance applications such as in water discharge.

1.3.2.2.1. High voltage modular f lyback topology

Considering the above study, combining the power converters found to be an efficient way to

improve the performance of pulsed power system based on generated voltage level and rate

of rise. Therefore, combining more than two power supplies sounds to be a good solution to

not only overcomes the power rating and operating speed problems but also increase the

system flexibility in generating different voltage levels. Moreover, due to modularity concept

easy diagnostics and ability to employ low voltage components are the other advantages.

Hence, as the first step to show the benefit of employing modular power supplies and

regarding the aforementioned advantages, series configuration is applied for 10 flyback

modules. Fig 1.46 illustrated the block diagram of the implemented hardware setup. To

decrease the number of switches, depending on the switch power ratings, a set of modules

Chapter 1

61

(here 5) is controlled with one switch. The transferred energy to the load can be controlled by

monitoring the primary stored energy. This is done through limiting the input current by

changing the duty cycle of the PWM signal in (1-5). As Fig 1.46 shows, a current sensor is

used for monitoring the input current. The experimental hardware setup for 10 series flyback

modules is depicted in Fig 1.47. Here 1700V IGBT modules (SKM200GB176D) are used as

power switches. Same gate drive modules and controller setup are used for this experiment as

the former one. Also same configurations for the transformers are used except here the

number of the cores per transformer is reduced to one. A HX10-NP is used as the current

transducer (CT).

Regarding the selected components each module is able to generate up to 4kV, which

makes the whole system capable of generating up to 40kV. The circuit is implemented with

fs=1kHz and VS=10V and 10% duty cycle. Each module has a 470pF capacitor (CO) and a

1MΩ high power resistor was connected as a load.

VSL

oa

d

Control

protocol of

switches

VLoad

+

-

Vo1

+

-

Vo2

+

-CO

CO

CO Vo3

+

-

CO Vo4

+

-

CO Vo5

+

-

Vo6

+

-

Vo7

+

-

Vo8

+

-

Vo9

+

-

Vo10

+

-

CO

CO

CO

CO

CO

Me

asu

rin

g th

e C

urr

en

t

Fig. 1.46 Block diagram of the implemented setup.

Chapter 1

62

Fig. 1.47 Hardware setup for 10-series flyback modules.


Fig 1.48 shows the measured practical results when the generated voltage is applied to a

resistive load. As can be seen from Fig 1.48 the voltage pulse of the series modules obtained

the peak amplitude of 20.8kV, while the single module has the peak voltage of 2.15kV. The

measured output voltage shows the rate of voltage rise of 8kV/µs in the series connection.

The peak voltage level and rate of voltage rise indicate approximately the 10 times (number

of series modules) better performance in comparison with a single module.

The proposed methods depicted advantage of generating high level of the voltages with

improved rise time, however, it is suitable for high impedance capacitive loads (such as DBD

loads) that means the load impedance can affect the system performance.

Chapter 1

63

Further analyses are presented in Chapter 3. The proposed method was published as a part

of a journal paper entitled “High Voltage Modular Power Supply Using Parallel and Series

Configurations of Flyback Converter for Pulsed Power Applications” published in IEEE

Transaction on Plasma Science Journal, Oct 2012 [140].

1.3.2.2.2. Modular switch-capacitor units

To design a flexible pulsed power supply which can be independent of load impedance but

still based on the modularity concept, here an efficient scheme that utilizes modular switch-

capacitor units in obtaining high voltage levels with fast rise time (dv/dt) using low voltage

solid-state switches is proposed. The proposed pulsed power supply has flexibility in terms of

controlling input energy and generating broad range of voltage levels. The energy flow can

be controlled by adjusting the utilized current source at the first stage of the system and

desirable voltage level can be obtained by connecting adequate number of switched-capacitor

units. The proposed topology is able to operate independently from the load while preserving

high repetition rate. Moreover, the control algorithm is designed in a way that it can prevent

from any possible faults and protect the system from the over-voltage.

Topology

Fig 1.49 illustrates the proposed method with a brief controlling algorithm scheme.

Generally, as can be seen the proposed topology consists of three different stages. The first

stage is a positive buck-boost converter as a current source and the energy storage section.

The next stage is the switch-capacitor units, which generates the required voltage level and

acts as a pulse generator, and the last stage is the load.

In order to have a flexible pulsed power supply, here the positive buck-boost converter

control the amount of stored energy by employing appropriate duty cycle for Si and SV. The

desirable voltage level can be obtained by connecting adequate number of switch-capacitor

units in series. This feature also makes the system capable of taking the advantage of

employing low voltage switches, as lower voltage rating switches have intrinsically better

behaviour [141].

In addition to the common aforementioned three sections, this topology utilizes a load

switch for low impedance applications such as water decontamination in which it disconnects

the load from the pulsed power supply before the required voltage level is obtained. This

makes this topology load independent, because the switch-capacitor units are not connected

Chapter 1

64

to the load while they are charging. In order to preserve a same break down voltage for the

load switch as the other switches, the switching signals need to be controlled precisely.

S11

S12

S21

S22

Sn1

Sn2

SL

Load

Si SVVS

L

D1

D2

C1

C2

Cn

Posi

tive

buck

-boost

conver

ter

Load Switch

Control

protocol of

switches

Sw

itch

-cap

acit

or

unit

s

Hysteresis

current control

Hysteresis

voltage control

Over

-rid

e si

gnal

Over

-rid

e si

gnal

Voltage

Feedback

Synchronizing

PWM Signal

Fig. 1.49 Proposed pulsed power topology for n level switch-capacitor units.

Control Strategy

The output current of the positive buck-boost converter need to be adjusted on a suitable

level. A hysteresis control is used for stabilizing this parameter. This control technique is

selected due to its simplicity, robust performance and stability [129, 142]. To charge switch-

capacitor units equally, the output voltage needs to be adjusted on an appropriate level. This

can be done same as the current control by utilizing the hysteresis method.

The charging process is synchronized with a PWM signal, which also determines the

repetition rate of the pulsed power supply. Here the procedure is explained for two switched-

capacitor units (Fig 1.50) which can be extended to n units. The procedure is controlled by

Sn1 and Sn2, which n is the capacitor unit number.

Chapter 1

65

When the PWM signal is high, the charging process begins with the first switch-capacitor

unit. At this mode (Fig 1.50a), S22 turns on and the stored energy form the current source

starts charging the capacitor of the first unit via anti-parallel body diode of S11. When the

capacitor voltage reaches to the required level the second mode (Fig 1.50b) starts by turning

S12 on and S22 off.

When the last unit charged up (here the second unit), SL and SV should be turned on prior to

applying all units voltages to the load (turning on the high-side switches). The reason is to

protect these switches from over voltage, as each switch blocking voltage is considered to be

equal to one capacitor unit voltage. As can be seen from Fig 1.50c in the third mode the load

is connected to the last unit and the current source is disconnected as SV is in the on-state.

Depending on the load impedance and its time constant in the worst scenario case a

low impedance load may fully discharge the last unit.

After SL and SV turned on, the forth mode occurred by turning on the high side switches of

the capacitor units. At this moment, as Fig 1.50d depicted, all of the capacitors connected in

series and a high voltage is applied to the load. Due to the last unit discharging in the

previous mode in the worst case the output voltage is:

Cout VnV )1(

(1-21)

where n is the number of switched-capacitor units and VC is the voltage of each unit.

Regarding (1-21) the voltage drop issue can be solved by increasing the number of the units.

Finally, when the PWM signal is low all of the high-side and low-side switches turn off and

on, respectively and the voltage across the load drops to zero (Fig 1.50e).

Regarding the mentioned charging and supplying the load process two different control

algorithms can be employed. The main difference between these two algorithms is the way

that the voltage feedback is provided.

One possible way to control the whole process is to provide individual voltage feedback

from each unit. Hence, the charging process of each unit can be started by monitoring the

previous stage charging level (Vch). Finally, when the unit’s capacitors charged to the

required level (VR) then it can be applied to the load. Fig 1.51a illustrated the algorithm

flowchart. The advantage of this algorithm is the ability of preventing from any possible

faults and indicating the failed unit as each unit’s voltage is measured. But the main

drawback is having multiple voltage feedback which increases the hardware implementation

complexity.

Chapter 1

66

The second potential way for the control algorithm is based on the measured voltage across

the output. This simplicity is the main advantage of this algorithm comparing with the

previous one, as it only employs one voltage feedback. Fig 1.51b shows the second algorithm

flowchart. As can be seen after charging each unit’s capacitor, low-side switches should turn

on in order to short circuit the output. Hence, at each charging stage the measured output

voltage corresponds to only one unit. By short circuiting the output after each charging stage

and using a flag (h in Fig 1.51b) it is possible to control the charging and supplying the load

process. The main advantages of the second algorithm are the ability to prevent from any

fault (as the output voltage is measured) and convenient practical implementation.

S11

S12

SL

LoadC1

S21

S22

C2

S11

S12

SL

LoadC1

S21

S22

C2

S11

S12

SL

LoadC1

S21

S22

C2

S11

S12

SL

LoadC1

S21

S22

C2

S11

S12

SL

LoadC1

S21

S22

C2

(a) (b)

(c) (d) (e)

Fig. 1.50 The charging and supplying the load operating modes.

Chapter 1

67

SK1: off

SK2: onK = 1,..., n

h =1

Start

If Sync

PWM=1

Yes

Yes

No

Sh1 & Sh2: off

Sg1: off

Sg2: on

g = 1,..., n – h

SL: off

If Vch=VR

NoYesIf VcK>VR

K =1,..., n

NoIf h = n

Yes

No

h = h + 1

Yes

SL & SV: on

Sn1: on

Sn2: off

SK1: off

SK2: on

K = 1,..., n-1 Delay

SK1: on

SK2: off

K = 1,..., n

If Sync

PWM=0

No

Charging unit h

SK1: off

SK2: on

Si: off

Fault detection

Supplying the Load

(a)

SK1: off

SK2: onK = 1,..., n

h =1

Start

If Sync

PWM=1

Yes

Yes

No

Sh1 & Sh2: off

Sg1: off

Sg2: on

g = 1,..., n – h

SL: off

If Vout=VR

NoYesIf Vout>VR

K =1,..., n

NoIf h = n

Yes

No

h = h + 1

Yes

SL & SV: on

Sn1: on

Sn2: off

SK1: off

SK2: on

K = 1,..., n-1 Delay

SK1: on

SK2: off

K = 1,..., n

If Sync

PWM=0

No

Charging unit h

SK1: off

SK2: on

Si: off

Fault detection

SK1: off

SK2: on

K = 1,..., n

Supplying the Load (b)

Fig. 1.51 Charging and supplying the load control algorithm flowchart, a) based on individual voltage feedback of each

unit and b) based only one voltage feedback.

Simulation Results and Analysis

In this section the proposed topology with three switched-capacitor units (n=3 in Fig 1.49)

is simulated in order to verify the performance of the proposed method. The software that

was used for simulation was MATLAB 7.10. Table 1.8 shows the parameters value used in

simulation corresponding to Fig 1.49. The upper and lower bands for adjusting the current are

selected as 20.5A and 19.5A, respectively. The charging level is selected as 1000V, and to

control the voltage of each unit on 1000V the bands are selected with 2V variations and fS is

the frequency of the synchronizing PWM signal. The delay time as mentioned in Fig. 1.51 is

selected as 1µs.


VS L C1,2,3 D fS iL VC1,2,3

100V 1mH 10nF %0.5 1kHz 20A 1000V

The effects of different loads impedance on the generated output voltage are depicted in Fig

1.52. For high impedance loads the considered delay for turning on the SL and SV doesn’t

Chapter 1

68

affect the output voltage level, but as the load impedance decreases the voltage drop across

the output voltage increases. This is due to the last unit capacitor discharging based on

the . For the low impedance applications this issue can be solved by increasing the

number of the switched-capacitor units.

Fig. 1.52 The effect of the load impedance on the generated output voltage.

The switching signals for the charging process and the generated output voltage are

demonstrated for 1kΩ load in Fig 1.53. As depicted during the charging process all of the

high-side switches are in the off-state. As can be seen, the last unit S31 turns on prior to the

other switches. Finally because the PWM signal becomes low, the output voltage drop to zero

before the load fully discharges the capacitors.

As mentioned before, in order to benefit from low voltage switches the break down voltage

of switches are selected to be equal to one capacitor unit voltage. Therefore, it is quite

important to turn on and off SL and SV effectively, otherwise they should tolerate the whole

output voltage. As Fig 1.54 shows, following the proposed switching strategy the voltage

across these two switches never exceeded from 1 kV. The only component in this topology

which must handle the whole output voltage is D2.

Chapter 1

69

Fig. 1.53 Voltage waveforms with relative switching signal patterns.

Fig. 1.54 Voltage across the critical components.

The proposed method is evaluated under different circumstances and obtained simulation

results indicate the effectiveness of the proposed approach. Various voltage levels can be

obtained by connecting different numbers of switched-capacitor units. The energy flow can

Chapter 1

70

be controlled via a positive buck-boost converter. The repetition rate can be adjusted using a

synchronizing PWM signal. Moreover, the proposed topology is load independent, as it is

disconnected from the load during the charging process. This feature makes the proposed

method suitable for supplying wide range of applications specially the one with low

impedance such as water discharge.

Chapter 4 presented extended analyses of this research. The proposed method was

published and presented entitled “A Flexible Solid-State Pulsed Power Topology” at the 15th

International Power Electronics and Motion Control Conference and Exposition, Novi Sad,

Serbia 2012 [143].

1.3.2.3. Pulsed power applications

After developing different pulsed power topologies studying and analysing different

application and load characteristics was considered at this stage. Applying pulsed power

supply at different voltage level and frequency and observing variety forms of plasma

generation were investigated. The designed pulsed power supplies under load condition not

only depicted a good performance but also the role of pulsed power for plasma generation in

industrial applications was also realized.

1.3.2.3.1. Effect of pulsed power on exhaust gas particle matter distribution

To evaluate pulsed power supply with a real world application, exhaust gas treatment was

selected as the first application. High voltage and high frequency pulses using a push-pull

based power electronics topology were applied to a DBD load and the effect of the generated

plasma was investigated.

Non-thermal plasma (NTP) treatment of exhaust gas is a promising technology for both

nitrogen oxides (NOX) and particulate matter (PM) reduction by introducing plasma into the

exhaust gases. This study considers the effect of NTP on PM mass reduction, PM size

distribution and PM removal efficiency. The experiments have been performed on real

exhaust gases from a diesel engine. The NTP is generated by applying high voltage pulses

using a pulsed power supply across a dielectric barrier discharge (DBD) reactor. The effects

of the applied high voltage pulses up to 19.44 kVpp with repetition rate of 10 kHz are

investigated. In this paper, it is shown that PM removal and PM size distribution need to be

considered both together, as it is possible to achieve high PM removal efficiency with

undesirable increase in the number of small particles. Regarding these two important factors

in this research, 17 kVpp voltage-level is determined to be an optimum point for the given

Chapter 1

71

configuration. Moreover, particles deposition on the surface of the DBD reactor was found to

be a significant phenomenon which should be considered in all plasma PM removal tests.

As mentioned, a pulsed power supply is employed to generate NTP using a dielectric

barrier discharge (DBD) reactor. Variety of DBD reactors existed, which essentially is a

multilayer capacitor and has been extensively used for various applications [40, 91, 144-149].

Also it can be a packed bed DBD reactor [36, 42, 150-155] or assisted by another catalyst or

adsorbent [24, 36, 151, 154, 156-160]. Here, a conventional DBD reactor with multipoint-to-

plane geometry is developed.

The particle size distribution of particulate matter from compression ignition engines has

become of increased concern. Particles differ in size, composition, solubility and therefore

also in their toxic properties. More than 90% of diesel exhaust-derived PM is smaller than 1

μm in diameter [161]. Most of the mass is in the 0.1–1.0 μm “accumulation” size fraction,

while most of the particle numbers are in the < 0.1 μm “nano-particle” fraction [162, 163].

These inhalable small particles are able to enter the bloodstream and even reach the brain

[164, 165]. It is believed that PM mass reduction and specially PM size distribution are not

fully considered in previous researches.

Regarding the aforementioned issues, the main concern of this study was to analyse the

effect of pulsed power on PM mass reduction and PM size distribution considering the pulsed

power effects on ultra-fine particles emitted from real diesel engine exhaust gas. In this

research, a pulsed power supply based on the push-pull inverter is developed to generate up

to 19.44 kVpp across the DBD load. The tests were conducted at different voltage levels with

fixed repetition rate of 10 kHz. PM mass reduction, PM removal efficiency and PM size

distribution are investigated by evaluating the results obtained.

Experimental Method

A. Experimental Setup

A schematic diagram of the experimental setup is shown in Fig 1.55. Experiments were

conducted on a modern turbo-charged 6-cylinder Cummins diesel engine (ISBe22031) at the

QUT Biofuel Engine Research Facility (BERF). The engine has a capacity of 5.9l, a bore of

102 mm, a stroke length of 120 mm, a compression ratio of 17.3:1 and maximum power of

162 kW at 2500 rpm. Particle number distributions are measured with a scanning mobility

particle sizer (SMPS) consisting of a TSI 3080 classifier, which pre-selects particles within a

narrow mobility (and hence size) range and a TSI 3025 condensation particle counter (CPC)

Chapter 1

72

which grows particles (via condensation) to optically detectable sizes. The SMPS software

increases the classifier voltage in a pre-determined manner so that particles within a 10-500

nm size range are pre-selected and subsequently counted using the CPC. The software also

integrates the particle number distribution to enable calculation of the total number of

particles emitted by the engine at each test mode. Gaseous emissions are measured with CAI

600 series gas analyses. CO2, NOX and CO concentrations can be measured by this gas

analyser, whereas particulate mass emissions are measured with a TSI 8530 Dust-Track II.

Fig. 1.55 Schematic diagram of plasma treatment system developed at QUT engine lab.

As depicted in Fig 1.55, three way valves to control the exhaust gas path ways are

employed. With this configuration, it is possible to measure both gaseous emissions and

particles before entering the reactor and after leaving it by changing the three way valve

directions. CO2 was used as a tracer gas in order to calculate the dilution ratio. CO2 was

measured either from the dilution tunnel with dilution ratios being calculated using the

following equation:

backgrounddiluted

backgroundexhaust

COCO

COCOratioDilution

,2,2

,2,2

(1-22)

Chapter 1

73

Laboratory background CO2 measurements were made before the commencement of each

test session. Every concentration measured after dilution should be modified by using a

dilution ratio.

B. DBD Reactor

A conventional dielectric barrier discharge reactor was designed for the experiments. Fig

1.56 shows a schematic of the reactor. As illustrated in Fig 1.56, it consists of two concentric

quartz tubes. Both tubes are 400 mm long and have a wall thickness of 1.5 mm. The outside

diameters of inner and outer quartz tubes are 20mm and 25mm, respectively. Exhaust gas

passes through the gap between these two quartz tubes. Based on pre-designed geometry, the

discharge gap is 1 mm. The DBD is connected to the pulsed power supply using internal and

external electrodes. The internal electrode is a copper cylinder and the external electrode is

made by a copper mesh that wraps the exterior part of the DBD. The electrodes are placed in

the middle of the DBD load with the length of 100 mm. Both tubes are fixed by two Teflon

caps at each end. Exhaust enters the reactor at the angle of 45 degree and flows throughout

the gap and leaves the reactor with the same angle.

(a)

(b)

Fig. 1.56 DBD reactor in Solidworks: a) schematic, b) cross-section.

C. Bipolar Pulsed Power Supply

Fig 1.57 shows a circuit schematic diagram of the pulsed power supply. As illustrated, it is

based on the push-pull inverter topology. The push-pull inverter contains two switches that

are driven with respect to ground. This is the main advantage of the inverter. This topology

Chapter 1

74

uses a centre-tapped transformer which is excited in both directions. A step up transformer is

used to boost the voltage and achieve galvanic isolation.

S2

S1

Vin

DB

D

NA

NB

NS

CS

CS

NA = NB << NS

Fig. 1.57 Pulsed power supply circuit schematic diagram (push-pull inverter)

The main reason to use a push-pull topology is to generate bipolar output voltage and

employing lower number of switches. The two switches S1 and S2 are switched alternately

with a controlled duty ratio to convert input DC voltage into high frequency AC voltage

suitable for exciting the DBD load. Hence, the generated output voltage is bipolar.

Adding a DBD load turns the push-pull inverter into a resonant stage with approximately

sinusoidal output. The frequency of the semi-sinusoidal shape signal is determined by an L-C

circuit comprising of the transformer inductance and capacitances of DBD and the

transformer. The repetition rate can be used to adjust the power and by optimizing the

resonance it is possible to obtain high frequency semi-sinusoidal waveform. A typical

measured output voltage of the employed pulsed power supply is depicted in Fig 1.58.

The first portion of the output voltage waveform is the resonant circuit dominated by the

magnetizing inductance of the transformer and the capacitances of the transformer and DBD.

The period of this signal is approximately 11.2 µs. The second one is the resonance

happening during the switches off-state between the leakage inductance and the capacitances

of the transformer and DBD. The period of this signal is equal to 34.8µs. As can be seen from

the figure the repetition rate is set to 10 kHz.

Fig 1.59 shows the experimental hardware setup for the pulsed power supply. Here 1200V

IGBT modules, SK75GB123, are used as power switches. Semikron Skyper 32-pro gate drive

modules are utilized to drive the IGBTs and provide the necessary isolation between the

Chapter 1

75

switching-signal ground and the power ground. A Texas Instrument TMS320F28335 DSC is

used for PWM signal generation. A centre-tapped step-up transformer with an UU100 core

3C90 grade material ferrite from Ferroxcube, are designed with NA = NB = 5 and NS = 293.

Here a 470pF capacitor (CS) is placed across each switch to protect them against the voltage

spikes. The output voltage is measured and captured using a Pintek DP-22Kpro differential

probe and RIGOL DS1204B oscilloscope, respectively.

Fig. 1.58 A typical measured output voltage of the employed pulsed power supply.

Fig. 1.59 Electrical Hardware setup with the DBD load.

Results and Discussion

A. Plasma Effect on PM Size Distribution

The effect of plasma on emission treatment is considered for varied experiments based on

the aforementioned Cummins diesel engine. In all experiments engine speed and load are

kept constant at 40 kW (25 %load) and 2000 rpm, respectively. A portion of raw exhaust gas

Chapter 1

76

directly from an iso-kinetic sampling port of the tailpipe was diluted with air and passed

through the reactor. Emissions concentration is measured before and after the applying pulse

to study the plasma effects. In addition, the median particle diameter which is another useful

parameter to study the effect of plasma technique is also measured. It is to be noted that all

illustrated results in each experiment have been obtained as an average over three consecutive

measurements.

The measured output voltages show the rate of voltage rise of 2840V/µs. In these

experiments, the output voltage amplitude is controlled by changing the input DC voltage

between 72.4 V, 84.4 V, and 94.8 V for generating output voltages of 15 kVpp, 17 kVpp, and

19.44 kVpp respectively. Under same situation, Fig 1.60 shows an image of DBD recorded at

19.44 kVpp. As can be seen, NTP is clearly occurring between the two electrodes.

Fig. 1.60 DBD image.

In the first experiment, the maximum voltage level (19.44 kVpp) is applied. To estimate

the deposition rate on the reactor surface, emissions in the reactor inlet (reactor inlet no pulse)

and reactor outlet (reactor outlet no pulse) without applying any pulse voltage are measured.

Finally, the pulsed power supply is applied across the DBD and the emissions in reactor

outlet are measured (reactor outlet with pulse). The same process is employed in all following

experiments.

Fig 1.61 illustrates the particle size distribution at 19.44 kVpp. There is a considerable

amount of PM deposition on reactor surface which is likely related to small gap between the

tubes (1mm). The median diameter in the reactor inlet is 70 nm, while in the reactor outlet is

decreased to 66 nm. This shows that larger particles deposited more on the reactor surface.

By applying pulsed power, as shown in this figure, the median diameter decreases

remarkably to 35 nm. This implies that lots of big particles have been oxidized or broken to

small particles by producing plasma inside the exhaust gasses at this voltage level.

Chapter 1

77

Fig. 1.61 Particle Size Distribution (2000rpm, 25%Load, 19.44 kVpp).

The peak value of particle number at the reactor inlet is around ⁄ .

This value declines to ⁄ at the reactor outlet due to deposition. By

applying pulsed power the graph peaks to 4.5 ⁄ which is approximately

four times bigger than the particle number at reactor outlet without any pulse. The number of

particles with diameter of less than 70 nm in the reactor outlet with applying pulse is higher

than the particle numbers at the reactor outlet without any pulse. This effect increases even

more at particle diameters less than 50 nm. At this level, the particle number at reactor outlet

surpasses the number of particles at reactor inlet. These findings imply that the 19.44 kVpp

pulse power at 10 kHz, increases the number of small particles considerably.

The effect of voltage level with 17 kVpp is considered in the second experiment. The

results obtained have been summarized in Fig 1.62. The figure shows the median diameter

changes from 70 nm at reactor inlet to 78 nm at reactor outlet without any pulse, and then

falls to 75 nm at reactor outlet with applying pulse. This shows that the larger particles are

deposited and also removed by plasma selectivity compared to smaller particles. As can be

seen, at this voltage level (17 kVpp) the number of small particles has not increased. This is

an important feature when compared with the previous experiment (19.44 kVpp).

Chapter 1

78

Fig. 1.62 Particle Size Distribution (2000rpm, 25%Load, 17 kVpp).

The last experiment is conducted by applying 15 kVpp pulses. The measured results are

depicted in Fig 1.63, which shows no growth in the number of small particles as well as

second experiment (17 kVpp). The maximum of PM concentration took place at around 71

nm. There is a small difference between PM concentrations with and without applying

plasma. Therefore, this level of voltage can remove particles at the same rate of deposition.

Comparing the values of median diameter in the reactor inlet and outlet shows that the

smaller particles are more likely to be deposited inside the reactor under the no pulse

condition. On the other hand, by applying the pulse the median diameter is in same range of

reactor inlet. There is no increase in the number of small particles at this voltage level. This

trend in median diameter variation is almost in complete agreement with the voltage of 17

kVpp.

Chapter 1

79


B. PM Removal Efficiency

PM removal can be calculated based on the following equation:

(1-23)

where and are the concentrations of PM (particle/cm3) at different PM

diameter. Fig 1.64 illustrates the PM removal efficiency at 19.44 kVpp. PM deposition on the

reactor surface increases with the PM size and gets to the maximum value of 70% removal at

around 80 nm. But after 80nm PM removal decreases again. For most of the particle sizes,

PM deposition on the reactor surface is more than 40%. When a 19.44 kVpp voltage is

applied to the reactor, PM removal efficiency reaches the value of 90% for larger particles.

Removal efficiency for particulate matter less than 80 nm is less than PM removal without

applying any pulse. This indicates undesirable operation of the plasma within this particle

size range. Also there is no removal for particles smaller than 50 nm. We suggest that this

incense in particle numbers can be related to the following two factors: Firstly, fragmentation

of larger particles by electron impact reactions or incomplete oxidation and secondly,

oxidation of exhaust gaseous emissions to particle by plasma ozone generated.

Chapter 1

80

Fig. 1.64 PM removal as a function of PM size (2000rpm, 25%Load, 19.44 kVpp)

Fig. 1.65 PM removal as a function of PM size (2000rpm, 25%Load, 17 kVpp)

Fig 1.65 Shows the PM removal at 17 kVpp which shows that PM removal by plasma is

more effective than deposition removal. This means that at this level of voltage, all deposited

particles and also some large particles inside the flow can be removed or oxidized. For

particles smaller than 35nm, PM removal percentage by deposition is higher than PM

removal by plasma. Apparently, smaller particles cannot be removed by plasma within this

Chapter 1

81


range (<35nm). However another possibility can be dissociation of larger particles to smaller

ones by electron impact reactions.

PM removal efficiency at 15 kVpp is illustrated in Fig 1.66. As can be seen the PM removal

for both graphs increases to an optimum value then decreases. The maximum PM removals in

reactor outlet with and without the pulse are 61% and 69% respectively. For particles larger

than the 60 nm diameter, PM removal when applying pulse voltage is slightly higher than PM

removal without any pulse. However, for particles with smaller diameters, these values are in

the same ranges.

Regarding the results obtained, it can be concluded that the voltage level has an important

role in the size dependent removal efficiency. At 15 kVpp the particle size distribution has

been affected slightly. The PM removal without producing small particles can be improved

as the voltage increases to 17 kVpp. The increase in the number of small particles has been

noticed when the voltage level goes up to 19.44 kVpp, while larger particles have been

reduced considerably. By taking into account all the above mentioned features, the 17 kVpp

experiment shows better efficiency in terms of particle size distribution for the given

configuration.

C. Plasma Effect on PM Mass Reduction

After studying the effect of different voltages on particle size distribution, in this section,

the effect of plasma on particle mass reduction is considered. In a similar way to the previous

Chapter 1

82

sections three different voltage levels have been applied to the DBD load. All results obtained

have been summarized in Table I. PM mass concentrations in the reactor inlet were 4.56 (

),

4.26 (

) and 5.14 (

) respecting to the variation of engine operating conditions for the

three tests conducted. These values decreased to 2.48 (

), 3.68 (

) and 4.37 (

) at reactor

outlet in the no pulse condition, respectively. This shows a significant particle deposition

inside the reactor. When the pulsed power is applied, plasma PM removal occurred. This

causes the PM mass concentration reduction of 43.9 %, 38.6% and 27.1% at 19.44 kVpp, 17

kVpp and 15 kVpp respectively.

The maximum PM mass reduction has been obtained when the voltage level is 19.44 kVpp.

However, according to the particle size distribution measurements, this voltage level

increases the number of small particles, which is not a desirable feature. The 17 kVpp applied

voltage shows a more suitable performance with good mass reduction of around 40% without

any increase in ultra-fine particle numbers. The 15 kVpp voltage level is found to be almost

the threshold breakdown voltage for the given configuration, below which no significant PM

mass reduction, PM removal, and PM size distribution which have not been affected too

much.

In this research the industrial application of pulsed power along with new findings, such as

PM distribution, PM mass reduction, and optimum operating point have been investigated. In

addition, the capability of employing power electronics topology to generate suitable voltage

levels with controllable parameters for a pulsed power application is demonstrated.

Table 1.9 PM Mass Reductions at Different Voltage Levels

Applied Voltage

Measurement

19.44 kVpp 17 kVpp 15 kVpp

Reactor Inlet PM

Concentration (3m

mg)

4.56 4.26 5.14

Reactor Outlet No Pulse PM

Concentration(3m

mg)

2.84 3.68 4.37

Reactor Outlet By-Pulse PM

Concentration (3m

mg)

2.56 2.62 3.74

Plasma PM Removal

Efficiency (%)

43.9 38.6 27.1

Chapter 1

83

Chapter 5 presented extended description about this work. This study entitled “Effect of

Pulsed Power on Particle matter in Diesel Engine Exhaust Using a DBD Plasma Reactor” is

published in IEEE Transactions on Plasma Science Journal on Aug 2013.

1.3.2.3.2. Analyzing plasma lamp intensity versus power consumption using

pulsed power supply

To study another type of DBD load with multi-point to multi-point geometry, a plasma lamp

is considered as the second application.

Recently, DBD lamps have gained much attention as they are mercury free, easily scalable

and simple to construct [166]. DBD lamps are mostly filled with rare gases and rare gas-

halides, which can provide efficient scheme for generating incoherent UV and VUV

(Vacuum UV) radiation [166]. The range of applications is broad owing to the variability of

output wavelength (88–310 nm) with the choice of gas fill.

DBD Lamps can be operated with continuous excitation or with pulsed excitation. Because

a pulsed discharge can operate at much higher peak voltages and peak currents for the same

average power as in a dc glow discharge, higher instantaneous sputtering, ionization and

excitation can be expected and hence better efficiencies [166]. Therefore, a solid-state pulsed

power supply can be a suitable choice for exciting a DBD lamp as it provides controllable

variables such as duty cycle, frequency, generated voltage, and etc.

One of the important features of a DBD lamp is the light intensity. The light intensity is quit

important in sensitive applications such as in therapy of skin diseases and for disinfecting the

skin surface. Hence, increasing or decreasing generated UV intensity can harm or not

efficiently affect the under treatment area respectively. The light intensity can be controlled

by factors such as frequency (pulse repetition rate) and applied voltage. However, another

important issue which needs to take into account is the lamp power consumption. Increment

in power consumption not only reduce the system performance but also can lead to short

operational lifetime of the lamp. Thus, finding the relation between the lamp intensity and

power consumption in order to select a proper operating point is important.

To study the behaviour of a DBD lamp, light intensity versus power consumption is

analysed at different voltage level and repetition rates. The DBD lamp is excited using a

push-pull based pulsed power supply. The light intensity is measured using a UV detector

and the power consumption is measured based on Lissajous V-Q diagram.

Chapter 1

84

Experimental Setup

A. DBD Lamp

Here a DBD lamp, as shown in Fig 1.67, is selected which is due to their particular interest

as they possess high efficiency [166]. The lamp tube has a coaxial geometry with a double

dielectric barrier, and is fabricated from UV grade fused silica tubing. The external electrodes

are wire/wire mesh yielding an active region of ~60mm length with a discharge gap of ~9mm

between the inner and outer dielectrics. The gas fill (sealed off) is a Xenon buffer at ~100mb

pressure, seeded with a low partial pressure (~1torr) of halogen (Cl) to yield XeCl excimers

upon discharge excitation which emit in the UV at 308m. Typical output irradiances from the

lamp are 1-10mW per cm2.

Fig. 1.67 Dielectric barrier discharge lamp

B. Pulsed Power Supply

Fig 1.68 shows a circuit schematic diagram of the developed pulsed power supply. As

illustrated, it is based on the push-pull inverter topology. The push-pull inverter contains two

switches that are driven with respect to ground. This is the main advantage of the inverter.

This topology uses a centre-tapped transformer which is excited in both directions. A step up

transformer is used to boost the voltage and achieve galvanic isolation.

The main reason to use a push-pull topology is to generate bipolar output voltage and

employing lower number of switches. The two switches S1 and S2 are switched alternately





Chapter 1

85



resonance it is possible to obtain high frequency semi-sinusoidal waveform.

Fig 1.69 shows the experimental hardware setup for the pulsed power supply. Here 1200V



switching-signal ground and the power ground. A Texas Instrument TMS320F28335 DSC is

used for PWM signal generation. A centre-tapped step-up transformer with an UU100 core

3C90 grade material ferrite from Ferroxcube, are designed with NA = NB = 5 and NS = 293.

Here a 470pF capacitor (CS) is placed across each switch to protect them against the voltage

spikes. To calculate the power consumption a 4.2nF capacitor (Cm) is connected in series

with the DBD lamp. Cm is selected large enough to not to affect the DBD lamp capacitance.

The output voltage is measured and captured using a Pintek DP-22Kpro differential probe

and RIGOL DS1204B oscilloscope, respectively.

S2

S1

Vin DB

D L

am

p

NA

NB

NS

CS

CS

NA = NB << NS

Cm



Chapter 1

86

C. Measurements

The lamp intensity was measured using a UV-photodetector (see Fig 1.69). The photo-

detector is equipped with JIC157 which has a spectral range of 210...390 nm. The averaged

intensity was considered for evaluation regarding to the area of the captured intensity

waveform and employed frequency.

Fig. 1.70 V-Q cyclogram of the DBD lamp as a basis of power consumption calculation.

To measure the power consumption of the DBD lamp, the energy transferred to the DBD

lamp has been calculated by applying the Lissajous (V −Q) diagram [167]. To measure Q one

capacitor (Cm) is placed in series with the DBD lamp, as depicted in Fig 1.68. Thus, by

measuring the voltage across Cm and multiplying it by Cm value it is possible to calculate Q.

The energy consumed by the plasma for one cycle is calculated from the area of V−Q curve

for different experiments (see Fig 1.70). Hence, by considering the employed repetition rate

(frequency) it is possible to calculate the average consumed power by the DBD lamp. Due to

presence of noise during measurement the captured data is filtered before power consumption

estimation stage (see Fig 1.70).

Results and Discussion

To analysis the lamp characteristics the lamp intensity versus the lamp power consumption

is considered at different voltage level and operating frequency. Input voltage (Vin) and

repetition rate (fr) were assigned as controlling parameters to determine the plasma lamp

characteristics. The high voltage pulses were applied to the plasma lamp at 7 different voltage

levels and 4 different repetition frequencies of 2kHz, 5kHz, 10kHz, and 15kHz (see Table

1.10). The measured results are illustrated as averaged intensity versus averaged power

Chapter 1

87

consumption (see Fig 1.71). It is to be noted that the results are normalized for a better

comparison.

Table 1.10 Considered voltages and frequencies for different experiments

Frequency

Output Voltage Level

2kHz

5kHz

10kHz

15kHz

Vin= 36V D11 D21 D31 D41

Vin= 40V D12 D22 D32 D42

Vin= 44V D13 D23 D33 D43

Vin= 48V D14 D24 D34 D44

Vin= 52V D15 D25 D35 D45

Vin= 56V D16 D26 D36 D46

Vin= 60V D17 D27 D37 D47

(a)

(b)

Fig. 1.71 Comparing different operation points regarding to the varied applied voltage levels and repetition

rates, a) applied voltage versus plasma lamp power consumption, b) applied voltage versus plasma lamp

intensity.

Chapter 1

88

The obtained results clearly indicate that the DBD lamp power consumption

correspondingly increases with the applied voltage level and the repetition rate. Despite the

power consumption which is always desirable to be as low as possible, it is the lamp intensity

which is the main goal. Obtaining high intensity at low power consumption in plasma lamp

applications is one of the major concerns. The obtained results regarding to the applied

voltage versus the lamp intensity are illustrated in Fig. 1.71. The measured results indicate

that not necessarily the higher intensity the high power consumption is. In the other word, the

intensity doesn’t have a linear relation with the power consumption. The measured results in

Fig. 1.71 depicted the possibility of achieving same or even higher intensity at lower power

consumption by increasing the repetition rate. For instance, comparing D44 with D36 shows

that, D44 has higher intensity than D36 while it has same power consumption. Such

behaviour can be realized by comparing D32 with D27 as well. This can be due to the

different slope of change regarding to the applied voltage and consumed power.

Regarding to the aforementioned facts, it is feasible to find an optimum operation point

where the required intensity can be achieved at lower power consumption. This can be

obtained by operating at lower voltage levels but with the cost of higher repetition rate.

Reducing the generated voltage level while obtaining the required performance is always

desirable due to high voltage insulation and power switches limited breakdown voltage

issues.

More analyses are provided in Chapter 6. This work entitled “Analysing DBD Plasma

Lamp Intensity versus Power Consumption Using a Push-Pull Pulsed Power Supply” is

accepted in 15th European Conference on Power Electronics and Applications to be held in

Lille, France, 2013. We thank Dr. Robert Carman of the Department of Physics and

Astronomy at Macquarie University, Sydney, Australia, for the helpful advice, and loan of

the XeCl lamp which was originally supplied by Prof. G. Zissis, LAPLACE Laboratory,

Universite Paul Sabatier, Toulouse, France.

1.3.3. High frequency high power converters for ultrasound

applications

The second aspect of this research project was improving the efficiency of high power

piezoelectric applications by applying high frequency high power converters. Extensive study

was conducted on high power piezoelectric device characteristics in order to generate suitable

excitation signal. Variety of power electronics topologies and control techniques were

Chapter 1

89

investigated in order to excite the high power piezoelectric device at the desirable range of

frequency. Two major steps were considered in this study:

Investigating the characteristics of a high power piezoelectric transducer

Improving the efficiency high power ultrasound system

1.3.3.1. Power electronic converters for high power ultrasound

transducers

Ultrasound power transducer performance is strongly related to the applied electrical

excitation [52, 98]. To have a suitable excitation for maximum energy conversion [5, 115-

119], it is required to analyse the effects of input signal waveform, medium and input signal

distortion on the characteristics of a high power ultrasound system (including ultrasound

transducer). In this research to investigate the features of a suitable excitation signal, varied

condition is considered. Different input voltage signals are generated using a single-phase

power inverter and a linear power amplifier to excite a high power ultrasound transducer in

different mediums (water and oil) in order to study the characteristics of the system.

Moreover, the effect of power converter output voltage distortions on the performance of the

high power ultrasound transducer using a passive filter is also analysed.

To study the performance of the piezoelectric transducer under the applied situation, an

ultrasound system was considered. The frequency responses of the applied voltage and the

measured voltage are considered as it is one of the most efficient ways to study the behaviour

of a system. In addition to that, it is possible to understand the linear or non-linear behaviour

of the system and the transducer (main goal) based on the applied signal and measured

response by employing the superposition method.

Experiments and Results

Table 1.11 shows a summary of all test conditions which have been carried out at different

mediums and input voltages. As can be seen, for low voltage evaluations sinusoidal

waveforms are generated by a linear power amplifier (OPA459) and a step-up transformer

(see Fig 1.72). But due to existing restrictions for the higher voltage levels, a single phase

inverter is used which results in a non-sinusoidal waveform.

The employed piezoelectric transducer has some resonant frequencies around 39 kHz. To

study the piezoelectric characteristics using superposition method and as the highest energy

Chapter 1

90

conversion happens at resonant frequencies one of the signals is selected at 39 kHz the other

one is selected at 61 kHz which is a non-resonant frequency.

Table 1.11 Test conditions and setups

Medium Excitation System

Input Voltage Magnitude

(peak)

Frequency of Input

Signal (s)

Test 1 water

Signal Generator, a power

amplifier and a high

frequency transformer

15 V and 30 V 39 kHz and 61 kHz

Test 2 water

Signal Generator, a power

amplifier and a high

frequency transformer

15 V + 30 V (time

domain) 39 kHz and 61 kHz

Test 3 oil three-level inverter 50 V, 100 V, 200 V and

300 V 39 kHz

Test 4 oil three-level inverter and a

filter

50 V, 100 V, 200 V and

300 V 39 kHz

Test 5 water three-level inverter 50 V, 100 V, 200 V and

300 V 39 kHz

Fig. 1.72 block diagram of a lab prototype for test 1 and test 2

In the first experiment, the first transducer was excited by applying a sinusoidal waveform

with two voltage levels (15V and 30V) at two different frequencies (39 kHz and 61 kHz).

The responses were captured across the second transducer in time domain. The obtained

results don’t show a proportional relation between the input and output voltages (see Fig

1.73). This can be due to the non-linear behaviour of the transducer. But to make sure the

system should be analysed under superposition principle. Based on this principle, if the

ultrasound system has linear characteristics, the responses of the system with two sinusoidal

signals (as follow) should be same:

a) Separately excited by two signals at 39 kHz and 61 kHz and adding the output signals

b) Simultaneously excited by two signals at 39 kHz and 61 kHz

Chapter 1

91

Therefore, in the second experiment, the same system is excited by two signals at 39 kHz

and 61 kHz which are added together at the input side of the power amplifier. It is to be

expected that the output voltages of the ultrasound system in two different tests should be

exactly same as each other if the system has a linear characteristics. To evaluate the linearity,

the previous obtained results are also added up together (based on superposition principle).

The measured result illustrated in Fig 1.74 shows that the output voltages (separately and

simultaneously excited) are not same, which implies the nonlinear characteristics of the

ultrasound system under the applied situation.

To study the non-linearity of ultrasound system at higher power ratings and different

mediums, experimentation were carried out using a single phase inverter. A coupling box of

laboratory setup is filled of oil or water for further experiments. A block diagram and the

experimental setup are shown in Fig 1.75 and Fig 1.76 respectively. The generated voltage

was a square wave uni-polar voltage waveform.

(a)

(b)

(c)

(d)

Fig. 1.73 (a) Input signals at 39 kHz (b) output signals at 39 kHz (c) Input signals at 61 kHz (d) output signals

at 61 kHz.

Chapter 1

92

(a)

(b)

(c)

(d)

Fig. 1.74 (a) test 1: summation of two output signals (at 39 kHz & 61 kHz) for Vin=15V and Vin=30V, (b) test

2: two output signals for Vin=15V and Vin=30V, (c) comparing the results of test 1 and test 2 for Vin=15V, (d)

comparing the results of test 1 and test 2 for Vin=30V.

Fig. 1.75 Schematic diagram of the setup using an inverter.

Chapter 1

93

Fig. 1.76 Hardware Setup.

(a)

(b)

Fig. 1.77 Captured results of third experiment

The third experiment was conducted at different voltage levels (50V, 100V, 200V and

300V) and at 39 kHz. Due to uni-polar modulation the generated signal has some harmonics.

To have a fare comparison all the input voltages are normalized so they have same amplitude.

Based on the same factor the captured output voltages are normalized as well (see Fig 1.77).

The results show that the ultrasound system has nonlinear characteristics at different voltage

levels and the output voltages are not same when they are normalized.

The output voltage of the power inverter generates voltage stress (dv/dt) across the

piezoelectric transducer. This voltage with the capacitive characteristics of the piezoelectric

transducer can generate significant current spikes which increases losses and high frequency

noise. The input voltage distortion can tend to deteriorate the output voltage quality and can

increase the power dissipation. Moreover, these distortions can cause the nonlinear behaviour

of the ultrasound system. Hence, a 990 μH inductor as a filter is placed between the power

converter and the transducer to reduce the input voltage distortion across the first transducer.

Chapter 1

94

Fig 1.78 shows a block diagram of the setup. The captured results after applying filter implies

that the ultrasound system still shows nonlinear characteristics (see Fig 1.79).

The final experiment was performed using water as medium same as third experiment but

this time by applying an inverter. Similar to the previous test results, when the input and

output voltages are normalized at 50 V, there are still significant differences between the

output voltages at different frequencies shown in Fig 1.80. The nonlinear behaviour of a high

power ultrasound system is obvious in this figure due to mismatch of the responses of the

ultrasound system to the normalized input voltages.

Fig. 1.78 Applying filter to the inverter output.

(a)

(b)

Fig. 1.79 Captured results of forth experiment (applying a filter).

(a)

(b)

Fig. 1.80 Captured results of the last experiment.

Chapter 1

95

Regarding to the mentioned experiments and evaluations, in this research, the nonlinearity

of the ultrasound system has been shown under varied situation. The obtained results indicate

the fact that for the high power ultrasound applications not necessarily the output voltage

increases with the increment in the input voltage. In addition, an inverter can be a suitable

choice for generating the required voltage even without filtering out the generated harmonics.

The extended analysis of this research is presented in chapter 7. This work was published

and presented entitled “Power Electronic Converters for High Power Ultrasound

Transducers” at The 7th

IEEE Conference on Industrial Electronics and Applications in July

2012 in Singapore [168].

1.3.3.2. Improving the efficiency of high power piezoelectric transducer

Regarding to the aforementioned outcomes the broad spectral content of the 2-level pulse

results in undesired harmonics which can decrease the performance of the system

significantly. On the other hand, it is crucial to excite the piezoelectric devices at their main

resonant frequency in order to have maximum energy conversion. Therefore a high quality,

low distorted power signal is needed to excite the high power piezoelectric transducer at its

resonant frequency. This paper proposes an efficient approach to develop the performance of

high power ultrasonic applications using multilevel inverters along with a frequency

estimation algorithm. In this method, the resonant frequencies are estimated based on relative

minimums of the piezoelectric impedance frequency response. The algorithm follows the

resonant frequency variation and adapts the multilevel inverter reference frequency to drive

an ultrasound transducer at high power. Extensive simulation and experimental results

indicate the effectiveness of the proposed approach.

The most efficient way to generate power signals is to use a 2-level power inverter.

Specifically, switch mode inverters are used for piezoelectric high power applications due to

their high power density, efficiency, low cost and size compared to conventional linear power

supplies [5, 102, 123]. However, the harmonics present in the output waveform produce

undesired side bands which are not suitable in many applications. Moreover, they also cause

unnecessary power dissipation which reduces the efficiency of the power converter [124,

125].

On the other hand, for ultrasound applications which operate at high fundamental frequency

power converters based on a PWM strategy with high switching frequency index (fs/fo)

cannot be a practical solution. In such applications, the maximum possible fundamental

Chapter 1

96

frequency of a converter is restricted by the switching transients of each power switching

event and the number of switching events per cycle.








Fig 1.32 depicted typical voltage waveforms and their harmonic spectra. It can be seen that

the harmonics are depressed for multilevel inverters such that they can be easily filtered out

from the frequency range of interest.

To improve the performance of the piezoelectric transducer for high power applications, in

addition to the multilevel converter, the device needs to be excited at its resonant frequency.

Piezoelectric devices typically have multiple resonant frequencies, but only the major

resonant frequency is generally targeted for excitation in practice. Structural and

environmental changes of a piezoelectric system can affect variations in the resonant

frequencies [5, 52]. Therefore, it is important to estimate the main resonant frequency in

order to maintain efficient system operation. Therefore, the multilevel converter needs to be

adapted with a suitable frequency estimation algorithm.

The most effective way to find the resonant frequencies of a piezoelectric transducer is by

evaluating its impedance frequency response [105, 108-113, 115, 117, 118]. A minimum in

the impedance response corresponds to a resonant frequency, fr. The impedance frequency

response is the ratio of the voltage spectrum to the current spectrum. To calculate the

piezoelectric impedance response, a voltage source needs to be applied to the device as an

excitation signal and current needs to be measured simultaneously. In order to obtain the

response of the device for a specific range of frequencies, the excitation signal should cover

the entire frequency range. The idea of calculating the piezoelectric impedance is inspired by

the general concept of performing system identification (i.e. finding the system transfer

function [121]). It is therefore possible to benefit from the existing knowledge base of

excitation signals for system identification. Considering the above mentioned features, a

broad-band excitation signal is the most appropriate candidate [120, 121].

This study advocates the advantage of using multilevel inverters along with an efficient

frequency estimation algorithm (as depicted in Fig 1.81) to develop the performance of the

Chapter 1

97

high power ultrasonic applications. In the proposed algorithm, a 1 kHz rectangular pulse with

10% duty cycle is applied to the device and the current is measured as the response. Then the

captured data is cropped to retain only one cycle of the applied input voltage and

corresponding output current. In the next step, FFT (Fast Fourier Transform) is applied to

both the signals and the impedance response is found as a ratio of the voltage to current

transforms. Finally, the relative minimum values are estimated and sorted according to both

impedance derivative and impedance magnitude. The FFT is used because it can be computed

relatively efficiently (with order NlogN operations), thus enabling real-time operations.

Finally, the multilevel inverter reference frequency is updated with the estimated resonance

frequency. The proposed method has been evaluated in simulations (using an electrical circuit

model) and experimentally (using two piezoelectric devices). The results obtained indicate

the efficiency and high performance of the proposed method.

DC/AC

Inverter

AC

Current Sensor

Piezoelectric

Frequency

Estimation

Fig. 1.81 Exploiting online resonance frequency estimation in high power applications of piezoelectric devices.

Methodology and Approach to Estimate Resonance Frequency

A. Excitation Signal

Piezoelectric devices have different resonance frequencies which are sometimes described

as vibration modes. To identify these frequencies, the excitation signal needs to be wide-

band. In the proposed approach, a sequence of rectangular pulses is applied to the device. The

pulse widths are 0.1ms and the pulses are reapplied every 1ms. It is quite easy to generate the

selected pulse stream and the pulses occur fast enough to enable the system to create regular

updates of the resonance frequency online.

Fig 1.82(a) shows the frequency response of a typical rectangular pulse. As can be seen the

energy in the spectrum is well spread, but varies considerably in intensity as a function of

frequency. This imperfection actually proves to an advantage because the practical spectrum

Chapter 1

98

does not suffer from the problem of having zero energy at some frequency components (see

Fig 1.82 (b)).

(a)

(b)

Fig. 1.82 Frequency responses: a) ideal pulse frequency response, b) practical pulse frequency response.

B. Estimating Impedance

The response due to the excitation signal (known as the ‘residual vibrations’) is given by:

n

k

kkk tatfSinAty1

)exp()2()( (1-24)

where n is the number of resonance frequencies, Ak is the amplitude, fk is the frequency and

ak is the damping coefficient of the thk resonance frequency.

A current sensor is used to capture the response (residual vibrations) of the device. The

current and voltage across the piezoelectric are captured simultaneously. To get the best

results, one cycle of the excitation pulse, along with the corresponding current response,

needs to be extracted from the captured signals. The starting point of the voltage waveform is

specified on the leading edge of the pulse. Then, from a knowledge of the sampling rate and

the length of the signal (which is 1ms) the end point is determined. Based on these starting

and end points the voltage and current waveforms are cropped from the captured signals.

After cropping, the FFT of the both signals are calculated as follows:

Chapter 1

99

1,...,1,0,)()(

1

1

2

NkenxkX

N

n

N

nki

(1-24)

where x(n) and X(k) are the discrete inputs and outputs respectively. To find the power

spectrum of the voltage and current signals the FFT outputs are multiplied by their conjugates

as per equation (1-25) and finally the impedance is calculated based on equation (1-26).

N

XXPx

.

(1-25)

NN

FffP

fPfP s

i

vz /)

2:0(,

)(

)()( (1-26)

where Fs denotes the sampling frequency and N is the number of FFT points.

C. Extracting Resonant Frequencies

The resonant frequencies correspond to the local (or relative) minimums of the piezoelectric

impedance. The relative minimums of a function are the points where that the slope of the

tangent changes from – to +. Fig 1.83(a) shows a power spectrum, of impedance and

the change of slope in the resonant frequency. At point A, is equal to zero. Moreover,

immediately to the left of this point at point B the slope is negative while at point C the slope

is positive.

Motivated by the above, the derivative of is calculated and all the points where

changes sign from – to + are extracted. As the final step, at the extracted frequencies, a

sorting is performed based on the magnitude of . The main resonant frequency is the

one with the lowest magnitude.

The procedure described above is depicted as an algorithm flowchart in Fig 1.83(b). Within

that flowchart R is a constant defining the number of resonant frequencies needing to be

extracted. The algorithm can be used both for offline and online systems. It is important to

note that, repeating the proposed algorithm, while applying a repetitive pulse and averaging,

results in an increase of the estimation accuracy.

Chapter 1

100

(a)

Input

captured

signals

Finding

starting and

end points

Cropping the

signals

Saving as

relative

minimum

Sorting using

amplitude

Yes

No

Finding the

minimum

amplitude

Number of extracted

frequencies = R

No

Yes

(b)

Fig. 1.83 Extracting resonant frequencies: a) change of slope at relative minimum (resonant frequency), b)

flowchart of the proposed algorithm.

Experimentation and Discussion

Chapter 1

101

The proposed algorithm is evaluated via both simulations and experimentation. The

comparisons were made based on three key types of alternative impedance analysing

methods. These alternative methods are:

a) The traditional method: In this method several single frequency sine waves from 30

kHz to 80 kHz have been applied separately to the piezoelectric. The frequency step

size was set to 1 kHz.

b) The Network Analyser method: An R&S ZVL3 vector network analyser has been used

to obtain the impedance frequency response of a piezoelectric device in the frequency

range of 30 kHz to 80 kHz.

c) The unit step and white noise excitation method: In order to compare the proposed

method with different wide-band excitation signals, the impedance frequency behaviour

has been obtained based on step response and white noise.

Step Pulse: From system identification theory, it is known that the Fourier transform

of the impulse response (h(t)) of a system gives the system transfer function. Since

generating an impulse ( ) at high power is highly impractical a step (u(t)) is

preferable. Here a 1ms step is applied to the device and as discussed in the previous

section the impedance response can be obtained by dividing the Fourier transform of

the output into the Fourier transform of the input.

White Noise: One of the most popular excitation signals used in system identification is

white noise [120, 121]. As white noise theoretically has a constant amount of energy per

frequency band, it is possible to simply look at the captured current to find the resonant

frequencies. To account for the fact that white noise spectra are not always perfectly flat

in practice, the impedance response is calculated in the same way as the other two

methods – by dividing the Fourier transform of the output into Fourier transform of the

input.

Two different piezoelectric devices Type A and Type B were considered for experimental

evaluation. As was the case with the simulation testing, the impedance of both devices was

obtained in the frequency range of 30 kHz to 80 kHz. For the proposed method a power

converter was used for generating pulses. For the traditional and broadband excitation

methods a G5100A function waveform generator was used as the signal generator source and

the signals were amplified using an OPA549. Fig 1.84 shows the experimental setup for the

proposed method. All signals were captured using a RIGOL DS1204B oscilloscope. As the

frequency was between 30 kHz and 80 kHz, the cut-off frequency of the filter was set to 100

Chapter 1

102

kHz. Hence capacitor and resistor values of Cf =1nF and Rf =1.59KΩ respectively were

selected for the low pass filter. The measured impedance frequency behaviour of the Type A

piezoelectric device for all methods is shown in Fig 1.85(a). Table 1.12 also shows the

extracted frequencies for the three strongest resonant frequencies.

Fig. 1.84 Experimental setup for the proposed method.

Fig. 1.85(b) shows the impedance frequency response of the second piezoelectric device.

The estimated frequencies for the first three resonant frequencies are shown in Table 1.12. As

can be seen from the measured results, the white noise method did not result in accurate

estimation of Fr2 and Fr3 compared to the other methods. The main reason is that in each

experiment the level of power in white noise changes randomly over time, and may even go

to zero at particular frequencies. Therefore for resonances which are not especially strong (as

was the case for Fr2 and Fr3 in the Type B device) the results are unreliable.

It should be noted that the electrical circuit model of the piezoelectric device is a simple

model which is not able to perfectly model the piezoelectric device nonlinearity, time

variations and high frequency behaviour. That is why there are small differences between the

simulation and test results. The advantages and drawbacks of the various methods are

summarized in Table 1.13. The proposed method offers simplicity and high performance in

addition to its ability to be used for online systems.

The performance of the proposed frequency estimation algorithm was evaluated in the

above sections, but as already mentioned, exciting the device at its resonant frequency is not

enough to achieve maximum power conversion. The excitation signal harmonics also need to

be considered. To do that an ultrasound interface is setup in the next section.

Chapter 1

103

(a)

(b)

Fig. 1.85 Experimental results obtained for the piezoelectric devices impedance response:

(a) Type A, (b) Type B.

Table 1.12 Estimated resonant frequencies of the Type A piezoelectric device

Method

Resonant Frequency(kHz) Fr1 Fr2 Fr3

Network Analyser 70.18 48.62 38.82

Impulse Response (Step Excitation) 70.31 48.83 39.06

White Noise 70.31 48.83 39.06

Traditional 70 49 39

Proposed 70.31 48.83 39.06

Chapter 1

104

Table 1.13 Estimated resonant frequencies of the Type B piezoelectric device

Method




White Noise 78.13 44.92 68.36


Proposed 78.13 46.88 66.41

In this section the advantage of exciting a piezoelectric transducer using a multi-level

waveform at the resonant frequency compared with a uni-polar waveform is illustrated. For

the comparison, one multi-level waveform and one uni-polar waveform were generated with

peak to peak voltage of 120V at 39 kHz (Type A device resonant frequency).

To perform the evaluation, one pair of the Type A piezoelectric transducers was placed face

to face as sender and receiver. For the first experiment, a unipolar pulse was applied to one of

the piezoelectric devices and the voltage across the other one was captured. Fig 1.86(a) shows

the applied voltages. To show their influence on piezoelectric devices the frequency

responses of applied and captured voltages are illustrated in Fig 1.86 (b). As can be seen from

Fig 1.86(b) the captured response contains several harmonics, due to the excitation signal

having much energy away from the fundamental frequency.

For the next test, a multi-level waveform was applied to the piezoelectric device. As can be

seen from Fig 1.86(c), harmonic levels are attenuated significantly. This was due to the use of

a multi-level waveform which damped the harmonics in the vast area around the fundamental

frequency.

Comparing the Fig 1.86(b) with Fig 1.86(c), illustrates that the maximum energy is

achieved at the resonant frequency. Higher efficiency is obtained for multi-level signal. In

particular the method is quite effective for reducing the harmonic content. In addition, the

presence of harmonics not only adversely affects frequency sensitive applications, but also

causes an increase in temperature and an increase in power loss. Moreover, when using filters

for attenuating remaining harmonics the filter cost and size decreases when multi-level

topology is employed.

Chapter 1

105

(a)

(d)

Input voltage (unipolar)

(b)

Input voltage (multilevel)

(e)

Output voltage (unipolar)

(c)

Output voltage (multilevel)

(f)

Fig. 1.86 Obtained results for the ultrasound interface: (a) applied uni-polar pulse at 39 kHz in time domain, (b)

frequency response of the applied uni-polar pulse (input signal), (c) frequency response of the output signal when

the input is a uni-polar pulse, (d) applied multi-level pulse at 39 kHz in time domain, (e) frequency response of the

applied multi-level pulse (input signal), (f) frequency response of the output signal when the input is a multi-level

pulse.

Further description of this study is provided in Chapter 8. This work entitled “Improving

the Efficiency of High Power Piezoelectric Transducers for Industrial Applications” was

published in IET Science, Measurement and Technology Journal, in Feb 2012 [169].

Chapter 1

106

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accepts overall responsibility for the publications;

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4- Potential conflicts of interest have been disclosed to (a) granting bodies, (b) the editor or publisher of

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5- They agree to the use of the publication in the student’s thesis and its publication on the Australasian

Digital Thesis database consistent with any limitations set by publisher requirements.

In the case of this chapter:

Parallel and Series Configurations of Flyback Converter for Pulsed Power Applications

Published in proceedings of: The 7th IEEE Conference on Industrial Electronics and Applications, ICIEA

2012, Singapore

Contributor Statement of contribution

Pooya Davari Proposed the initial design and conducted mathematical proof, simulation studies,

data analysis, and wrote the manuscript.

22 Feb 2013

Firuz Zare Proposed the initial idea and supervised the validity studies including: conducting

the simulations and writing the manuscript.

Arindam Ghosh Aided data analysis and writing the paper.

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117

Chapter

2 . P a r a l l e l a n d S e r i e s C o n f i g u r a t i o n s o f

F l y b a c k C o n v e r t e r f o r P u l s e d P o w e r

A p p l i c a t i o n s

* School of Electrical Engineering and Computer Science, Queensland University of Technology,

GPO BOX 2434, Brisbane, Australia

Presented and Published at: The 7th

IEEE Conference on Industrial Electronics and

Applications, ICIEA 2012, Singapore, July 2012.

2

Chapter 2

118

Abstract— Advances in solid-state switches and power electronics techniques have led to

the development of compact, efficient and more reliable pulsed power systems. Although, the

power rating and operation speed of the new solid-state switches are considerably increased,

their low blocking voltage level puts a limits in the pulsed power operation. This paper

proposes the advantage of parallel and series configurations of pulsed power modules in

obtaining high voltage levels with fast rise time (dv/dt) using only conventional switches. The

proposed configuration is based on two flyback modules. The effectiveness of the proposed

approach is verified by numerical simulations, and the advantages of each configuration are

indicated in comparison with a single module.

2.1. Keywords

Pulsed power, Flyback converter, Parallel and series connection, High voltage pulse

2.2. Introduction

Rapid release of stored energy as electrical pulses into a load can result in delivery of large

amounts of instantaneous power over a short period. This strategy is called pulsed power [1].

Generally, a pulsed power system consists of three main parts: energy storage, pulse generator

and the load. The most prominent part of a pulsed power system is the pulse generator, which

is based on utilized switch and topology. Hence, the switch is the connecting device between

the storage and the load. This means that the whole system characteristics such as rise time,

repetition rate, voltage rating, efficiency, cost, life time, etc, depend on the employed switch

specifications.

Gas-state and magnetic switches have been widely used in pulsed power technology, as

they possess a very high electric strength and fast rise time [2]. Gas-state switches require

special operating conditions such as high pressure, vacuum equipments and gas supplies. In

addition, they are bulky, unreliable, have short lifetime span and low repetition rate. Even with

the magnetic switches, which have a higher repetition rate, the problems remain. These

conditions limit the mobility, efficiency and increase the cost and the size of the pulsed power

system [2, 3]. On the other hand, solid-state switches are compact, reliable, cost effective, and

have a long lifetime and repetition rate. The main drawbacks of solid-state switches are their

limited power rating and operation speed [2, 3].


speed) and exploiting power electronics techniques and topologies have led to compact,

Chapter 2

119

efficient and more reliable pulsed power systems. The new developed solid-state switches

such as IGBT (Insulated-Gate Bipolar Transistor) and Integrated Gate-Commutated Thyristor

(IGCT) have high power rating [4], but their lower operation speed comparing with the gas-

state switches and high cost still put limits in the pulsed power supplies.

One way to increase the pulsed power supply performance and cover the switch limits is to

explore alternative circuit topologies. Varieties of circuit topologies such as Marx generators

(MG) [5], pulse forming network (PFN) [6], magnetic pulse compressors (MPC) [7] and

multistage Blumlein lines (MBL) [8] have been introduced. These topologies have been

widely used in pulsed power supplies, but complexity, inflexibility and inefficiency are their

main drawbacks [2, 9]. Interest in applying power electronics topologies and techniques to

increase power supply efficiency and reliability is growing fast. In the last decade, research

and studies designate the advantage of using DC-DC converters in pulsed power applications

[3, 9, 10]. However, if one wants to generate extra high voltages, due to components hold-on

voltage limits (such as capacitors, solid-state switches and etc), these schemes become

ineffective.

In addition to topology, another way is to combine power supplies. Variety of converter

configurations have been introduced for improving the power supply specifications such as

output ripple, high input voltage, output power ratings, etc [11, 12]. Parallel and series

connections are one of the well known combinations. Designing power supplies based on

series or parallel connection are widely employed for different applications [11, 12].

This paper proposes parallel and series connections of flyback converter modules to

develop power rating and rise time of the pulsed power supply using conventional low voltage

switches. Flyback converter is selected due to its unique properties in pulsed power as it is

discussed in the next section. The proposed approach is evaluated by numerical simulations,

and the advantages of each configuration are indicated in comparison with a single module.

The evaluated results indicate the effectiveness and efficiency of the proposed method.

2.3. Topology

The proposed approach utilized flyback topology, which is one of the well known

topologies in the power electronics [13]. Conventional flyback converter is usually preferred

as it is simple, has only one switch and magnetic component, is able to generate high voltage,

can provide multiple outputs, isolation, etc [13, 14]. But when it comes to pulsed power


Chapter 2

120

applications, in addition to the mentioned features, it has some more advantages which make it

more suitable. Main features of a flyback converter for pulsed power applications are:

The transformer, in addition to electrical isolation and energy storage also steps down

the reflected voltage across the switch; therefore lower voltage rate switches are needed

comparing with other topologies.



As the pulsed power applications mostly have R-C characteristics [9], a current source


Considering above mentioned features the current source topology is selected for the

proposed method. Here basic principle and operation modes of this converter are briefly

described.

The behaviour of a flyback converter can be realized by modelling the transformer with a

simple equivalent circuit consisting of an ideal transformer, magnetizing inductance (Lm) and

leakage inductance (Ll). Fig. 2.1 shows a flyback converter including the simple model of the

transformer connected to a load. In this figure CO is the converter equivalent output

capacitance. This capacitance can get affected in the case of an R-C load. It is to be mentioned

that, to keep the analysis simple, following assumptions are made:

1) The switch and diode are ideal (voltage drop across them is zero).

2) The switch and diode output capacitances are zero.

3) Conduction and switching losses are negligible.

4) The transformer stray capacitances are negligible.

Basically a flyback converter transfers energy from a source into the transformer

magnetizing inductance when the switch is on, and then transfers the stored energy to the load

while the switch is off. The proposed approach operates in the DCM (Discontinues

Conduction Mode). The operating modes are briefly summarized for 4 different modes

(illustrated in Fig. 2.2) below:

Mode 1: In this mode the switch is in the on-state. As Fig. 2.2 depicts, the current flows

through Lm and, during this stage, the energy is stored in the inductor. This mode lasts

Chapter 2

121

depending on the duty cycle of the PWM signal. The relationship between Lm and Vs can be

expressed as:

s

mm

s

mmsms

TD

IL

TD

ILV

t

iLV

.0.

0

(2-1)

where Vs is the dc supply voltage, Im is the maximum current, D is the duty cycle and Ts is the

switching period.

Mode 2: When the switch is turned off the magnetizing current circulates through the

primary side of the transformer and the diode in the secondary side is turned on. As the switch

is turned off the current flowing through Ll is decreased to zero, which induces voltage spike

across the switch according todt

diLv ll . This voltage may damage the switch if it exceeds

the switch break down voltage level. A snubber can provide a path for this current and damps

the spike to protect the switch [15].

Mode 3: The diode is turned on and the current stored in the magnetising inductance flows

to the secondary side. The maximum voltage across the switch at this stage is:

maxmax1

)( osswitch vn

Vv (2-2)

where, Vo is the maximum output voltage level and n is the transformer turns ratio.

Mode 4: As the converter operates in DCM in this mode, all the stored energy in Lm is

completely transferred to CO, causing the diode to get turned off.

Lm

Vs

C0 Load

1 : n

Ll

Vo

+

-


Chapter 2

122

Ll

Lm

VS

1 : n

LoadCo

VS VS

Mode 1

Mode 3 Mode 4

1 : n

Co

Mode 2

Ll

Lm

VS

Load

Lm Lm

Ll Ll

1 : n 1 : n

Co CoLoad Load

Vo

+

-

Vo

+

-

Vo

+

-

Vo

+

-

Fig. 2.2 Operating modes in a flyback converter.

2.4. Proposed Method

High voltage pulses with fast rise-time are the two main important features of a pulsed

power supply [1-3]. Fig. 2.3 illustrates the proposed method, which employs the advantage of

parallel and series configuration of the pulsed power modules to increase the voltage level and

the rise-time, while using the low-voltage switches. As the figure shows, the parallel and

series connections are only considered for the secondary side of the converter.

2.4.1. Single Module

To understand and compare the parallel and series configuration, first a single module

characteristic is described. As mentioned, a flyback converter can operate as a current source.

Therefore, circuit schematic shown in Fig. 2.1 is simplified as in Fig. 2.4 Comparing these two

figures, the estimation of the output voltage can be summarized as below:

First, as described earlier, the magnetizing inductance is charged by the DC power supply.

The charged energy stored in the inductor Lm under ideal situation can be expressed as:

2

2

1mmpri ILE

(2-3)

This stored energy will be transferred to the output capacitor when the switch is in the off-

state. The capacitive stored energy is:

Chapter 2

123

2sec

2

1ooVCE

(2-4)




o

mmo

C

LIV

(2-5)

The rise time for the generated voltage is defined as dv/dt. When the switch is turned off,



o

m

o

o

nC

I

C

I

dt

dv

(2-6)



The above equations show the effect of stored current level, magnetising inductance and

the output capacitor on the generated output voltage and the rise-time. The idea of parallel and

series configurations of the pulsed power modules is inspired by both equations (2-5) and (2-

6).

2.4.2. Parallel Modules

Considering Fig. 2.3.(a) and the fact that each transformer is acting as a current source, the

stored energy in the primary side is doubled when two modules are connected in parallel. This

stored energy will be transferred to the output capacitors; hence considering (2-3) and (2-4)

the output voltage magnitude and its rate of rise are as follows:

op

mmparallelo

C

LIV 2

(2-7)

op

o

Parallel C

I

dt

dv 2

(2-8)

Chapter 2

124

where Cop=Co1+Co2. Equations (2-7) and (2-8) indicate the importance of the output capacitor.

Even if the modules are paralleled the output capacitance can affect the output voltage level

and rise-time and keep them same as the single module. Therefore, the parallel configuration

is beneficial in pulsed power applications when Cop=Co. This happens in the case of R-C load

when the load capacitance is much bigger than the power supply output capacitance.

Ll

Lm

Vs

1 : n

Co2

Ll

Lm

Vs

n : 1

Co1Load

Vo

+

-

(a)

Ll

Lm

Vs

1 : n

Co1

Load

Ll

Lm

1 : n

Co2

Vo

+

-

(b)


Chapter 2

125

CoIo Vo

+

-

Load


2.4.3. Series Modules

Second alternative is series connection of the pulsed power modules, as illustrated in Fig.

2.3(b). Here, the injected energy to the output is also doubled. But as described, the generated

voltage features also depend on the output capacitance. Taking into account the equations (2-

3) to (2-6), the output voltage magnitude and its rate of rise can be expressed as:

os

mmSerieso

C

LIV 2

(2-9)

os

o

Series C

I

dt

dv

(2-10)

where Cos is Co1Co2/(Co1+Co2). Same as in parallel modules, in series modules, the output

voltage level and rise time improve as the output capacitance decreases. Here the series

modules are beneficial in both level of voltage and its rate of rise when Cos<Co. The influence

of N modules and equivalent output capacitor (for two different cases) on the output voltage

and rise time are summarized in Tables 2.1 and 2.2.

As Fig. 2.3(b) shows, in the series connection each switch withstands its own module

reflected output voltage, while in the parallel connection, each switch should tolerate the

whole reflected output voltage.

Chapter 2

126



dt

dv

Parallel Singleo

o

mm V

C

LI

Singleo

o

dt

dv

C

I

Series Singleo

o

mm NV

C

LNI

Singleo

o

dt

dvN

C

IN



dt

dv

Parallel Singleo

o

mm VN

C

LIN

Singleo

o

dt

dvN

C

IN

Series Singleo

o

mm VN

C

LIN

Singleo

o

dt

dv

C

I

This fact makes the series connected modules more appropriate for generating high voltage

waveforms using low voltage switches. Therefore, (2-2) can be rewritten as below for N-series

module:

maxmax1

)( osswitch vNn

Vv

(2-11)

2.4.4. Operating Conditions


designing procedure of pulsed power modules. These are:

1) The output capacitor should be completely discharged during each period; otherwise the

whole stored energy in the magnetising inductor will not transfer to the capacitor.

2) Synchronization of each switch gate signal is required. Delays between each module

gate signal reduce the performance of the system.

3) It is also important to consider the damping factor (ζ). As shown in Fig. 2.4, the entire

circuit acts as a parallel RLC circuit (the current source is an inductor). Hence, the

output voltage and the rise time are the results of the resonance effect of this circuit.

The output voltage can be expressed as:

Chapter 2

127

tstso eAeAtV 21

21)( (2-12)


20

22,1 S

(2-13)

here, RC2

1 is known as neper frequency and LC

1


0)0(

toV and oItCI

)0( are the primary conditions. Io is the stored current in the

magnetizing inductor (Im=nIo) transferred to the secondary side. The coefficients A1 and A2 are

determined by the primary conditions as:

02

02

20

212

LIAA

(2-14)


Underdamped ζ <1 or α < 0 , C

L

R2

1

C

LIA

2

01

(2-15)

Overdamped ζ >1 or α > 0

01 RIA

(2-16)

Equation (2-15) shows that if the circuit is in underdamped situation, then the generated

voltage amplitude is independent of the load impedance, which is the situation where the

proposed method is valid. But under overdamped condition as in (2-16), the output voltage

amplitude depends on the load impedance and the initial current. Hence, in overdamped

condition, the parallel connection has better performance.

2.5. Simulation Results

To verify the proposed method and the theoretical analysis, simulations under different

conditions were carried out. Here the performance of parallel and series flyback configurations

Chapter 2

128

(Fig. 2.3) is compared with a single flyback converter (Fig. 2.1). The software that was used

for simulation was Matlab 7.10. Table 2.3 shows the parameters value used in simulation

corresponding to Fig. 2.1. In order to make the simulation results more close to reality, the

transformer core and copper losses (Rc and Rw) are also considered.

To evaluate the proposed method in developing the features of a single module pulsed

power supply, four distinct cases are considered. Case 1 and 2 have been conducted regarding

the conditions presented in Table 2.1 and 2.2, respectively. In these cases, to simulate a plasma

phenomenon, the load gets connected to the output after the output voltage reaches to a certain

voltage level (threshold) [9]. The different parameters in each case are indicated in Table 2.4.

Like all other methods, the proposed method has its own drawbacks under certain

conditions. These conditions, mentioned as operating conditions in the previous section, are

considered in Case 3 and 4. In Case 3 the consequence of unsynchronized gate signals on the

system performance is described. Finally, in Case 4 the effect of damping factor on the

generated pulses is examined. Except in Case 4 for all the other cases a 1kΩ resistor is

selected as the load.


Vs

(v) Fs

(kHz) D

(%) Lm

(µH) Llp

(µH) Lls

(mH) Rc

(kΩ) Rw1,2

(Ω) n

10 1 8.9 160 2 4 1 0.1 10

Table 2.4 Simulation Parameters in Case 1 and 2

Parameters Value Case1 Case2

Single Parallel Series Single Parallel Series

Threshold 1000v 1000v 1978v 1000v 1410v 1410v

Co 4nF 8nF 2nF 4nF 4nF 4nF

Co1=Co2 4nF 4nF 2nF 8nF

2.5.1. Case1


supplies is compared with the single one. Therefore, in this case the output capacitor of each

module is equal.

Regarding the capacitor value mentioned in Table 2.4, 4nF for each module, the equivalent

capacitance of the series and parallel modules are equal to 2nF and 8nF, respectively. Fig. 2.5

shows the obtained output voltage waveform. As can be seen in the series connection, both

Chapter 2

129

voltage level and rate of rise are increased with the factor of N, (N=2 is considered here). This

has been shown in the analysis of section 2.3 and Table 2.1.

Regarding (2-2) and (2-11) the reflected voltage across the switches is about 110V for both

parallel and series modules. This shows ability of the proposed method in generating high

voltages with low voltage switches. So, it is possible to benefit from ultrafast switches, as

switch operation speed increases when its hold-on voltage decreases.

As the rate of charging is in terms of a time constant RC; hence, as illustrated in the figure,

series connection discharging time is shorter than the parallel and single modules.

2.5.2. Case2

If the load capacitance, which is paralleled with the power supply output, is much bigger

than the power supply output capacitance, then the entire single, parallel and series modules

will have the same equivalent output capacitance. In this case the output capacitance of all

modules is considered to be equal to 4nF (see Table 2.4). As depicted in Fig. 2.6, both parallel

and series modules have same voltage level but the parallel connection shows better

performance in the case of dv/dt.

Fig. 2.5 Output voltage waveform of single, parallel and series modules in Case1 (Co1=Co2=Co).


Chapter 2

130

2.5.3. Case3

Due to differences in semiconductor characteristics and the gate drive circuits, it is quite

important to select identical devices and synchronize the gate signals. Here the same situation

in Case 2 is selected, but 0.2ms delay between gate signals is considered to show the effect.

Fig. 2.7 shows the generated output voltage for parallel and series modules. As can be seen the

output amplitude never reached to the threshold value. Therefore the load is not connected, so

the capacitors are not completely discharged. This creates another problem since the entire

energy in the magnetizing inductor is not fully transferred to the capacitor and the mentioned

equations for estimating the output voltage are not valid.

2.5.4. Case4

In some applications, the load has low impedance like discharge plasmas in water [3]. In

these cases damping factor plays important role in system performance as it discussed in part

D of the previous section.

Here the series and parallel modules are considered, same as in Case 1, but RL = 100Ω is

directly connected. This value puts the all three power supplies in the overdamped condition.

As Fig. 2.8 illustrates the parallel modules show better performance, while series module had

far better performance in Case1 when the system was underdamped. This means that all the

mentioned features about parallel and series connections are valid when the system is

underdamped (see (2-15)). The overdamping effect of the load can be seen as the output

voltage levels are dropped tremendously compared to those in Case 1. Because when the load

resistance decreases, the time constant, τ=L/R, increases, and hence the inductor energy is not

completely transferred to the output capacitor. The easiest way to overcome this problem,

considering ζ in (2-15), is increasing the output capacitance.

Fig. 2.7 Output voltage waveform of single, parallel and series modules with 0.2ms delay between modules gate

signals.

Chapter 2

131

Fig. 2.8 The effect of the load and the damping factor on the system performance.

2.6. Conclusion

This paper demonstrates the advantages of parallel and series configuration of flyback

converters for pulsed power applications. The proposed method aims at increasing the voltage

level and rise time of the generated pulses using conventional low voltage switches. In this

method the flyback converter topology is employed as it shows beneficial characteristics

especially for pulsed power applications. The proposed method is evaluated under different

circumstances and obtained simulation results indicate the effectiveness of the approach.

2.7. References

[1] H. Bluhm, Pulsed Power Systems: Principle and Applications. Berlin Heidelberg: Springer-

Verlag, 2006.






[4] ABB, "High power semiconductors," in Short form catalogue, ed, 2003.

[5] T. Heeren, T. Ueno, D. Wang, T. Namihira, S. Katsuki, and H. Akiyama, "Novel Dual Marx

Generator for Microplasma Applications," Plasma Science, IEEE Transactions on, vol. 33,

pp. 1205-1209, 2005.

[6] T. G. Engel and W. C. Nunnally, "Design and operation of a sequentially-fired pulse forming

network for non-linear loads," Plasma Science, IEEE Transactions on, vol. 33, pp. 2060-

2065, 2005.




1744-1750, 2006.

Chapter 2

132

[8] D. Durga Praveen Kumar, S. Mitra, K. Senthil, A. Sharma, K. V. Nagesh, S. K. Singh, et al.,


Blumlein," Review of Scientific Instruments, vol. 78, pp. 115107-115107-4, 2007.




[10] S. Zabihi, F. Zare, G. Ledwich, and A. Ghosh, "A novel high voltage pulsed power supply

based on low voltage switch-capacitor units," in Pulsed Power Conference, 2009 IET

European, 2009, pp. 1-4.

[11] A. Cid-Pastor, L. Martinez-Salamero, C. Alonso, R. Leyva, and S. Singer, "Paralleling DC-

DC Switching Converters by Means of Power Gyrators," Power Electronics, IEEE


[12] B. R. Lin, H. K. Chiang, C. C. Chen, C. S. Lin, and A. Chiang, "Analysis and implementation

of soft switching converter with series-connected transformers," Electric Power Applications,

IET, vol. 1, pp. 82-92, 2007.



on, vol. 37, pp. 171-178, 2009.



1189-1193.

[15] M. Peipei, W. Xinke, Y. Jianyou, C. Henglin, and Q. Zhaoming, "Analysis and design

considerations for EMI and losses of RCD snubber in flyback converter," in Applied Power

Electronics Conference and Exposition (APEC), 2010 Twenty-Fifth Annual IEEE, 2010, pp.

642-647.

133













High Voltage Modular Power Supply Using Parallel and Series Configurations of Flyback Converter

for Pulsed Power Applications

Published in: IEEE Transactions on Plasma Science, Vol.40, No.10, pp. 2578-2587, Oct. 2012.


Pooya Davari Proposed the initial design and conducted simulation and data analysis.

Implemented the hardware setup, designed the control strategy, conducted

experimental verifications and wrote the manuscript

22 Feb 2013

Firuz Zare Proposed the initial idea and supervised implementation process, experimentation,

and writing the manuscript

Arindam Ghosh Aided data analysis and writing the paper

Hidenori Akiyama Provided general information about the pulsed power applications demands



Prof. Arindam Ghosh


134

Chapter

3 . H i g h V o l t a g e M o d u l a r P o w e r S u p p l y

U s i n g P a r a l l e l a n d S e r i e s C o n f i g u r a t i o n s

o f F l y b a c k C o n v e r t e r f o r P u l s e d P o w e r




† Kumamoto University, Japan

Published in: IEEE Transactions on Plasma Science, Vol.40, No.10, pp. 2578-2587, Oct.

2012.

3

Chapter 3

135

Abstract— To cover wide range of pulsed power applications, this paper proposes a

modularity concept to improve the performance and flexibility of the pulsed power supply.

The proposed scheme utilizes the advantage of parallel and series configurations of flyback

modules in obtaining high voltage levels with fast rise time (dv/dt). Prototypes were

implemented using 600V IGBTs (Insulated-Gate Bipolar Transistor) switches to generate up

to 4 kV output pulses with 1 kHz repetition rate for experimentation. To assess the proposed

modular approach for higher number of the modules, prototypes were implemented using

1700V IGBTs switches, based on 10 series modules, and tested up to 20kV. Conducted

experimental results verified the effectiveness of the proposed method.

3.1. Index Terms

Pulsed power, Flyback converter, Parallel and series connection, High voltage pulse

3.2. Introduction



To generate such pulses variety of research and studies have been conducted in pulsed power

area. The most prominent part of a pulsed power system is the utilized switch and topology.

This means that the whole system characteristics such as rise time, repetition rate, voltage

rating, cost, life time, etc, depend on the designed topology and employed switches

specifications.







system [2, 3]. On the other hand, solid-state switches are compact, reliable, cost effective, and

have a long lifetime and repetition rate. The main drawbacks of solid-state switches are their

limited power rating and operation speed [2, 3].


speed) and exploiting power electronics techniques and topologies have led to compact and


Chapter 3

136

more reliable pulsed power systems. The new developed solid-state switches such as IGBT

and Integrated Gate-Commutated Thyristor (IGCT) have high power rating, but their lower

operation speed comparing with the gas-state switches and high cost still put limits in the

pulsed power supplies [2, 4-6]. For example 5SHY42L6500 which is one of the recent

developed IGCT switches can handle voltage up to 6.5kV and its rise time is 1µs [6], while

GP-81B (triggered spark gap) break down voltage is 120kV with 200ns rise time [7].

Topologies are considered not only as an alternative way to overcome the switch limits but

also in developing flexible and compact systems. Varieties of circuit topologies such as Marx

generators (MG) [8], pulse forming network (PFN) [9], magnetic pulse compressors (MPC)

[10] and multistage Blumlein lines (MBL) [11] have been introduced. These topologies have

been widely used in pulsed power supplies, but complexity, inflexibility and inefficiency are

their main drawbacks [2, 12]. Interest in applying power electronics topologies and techniques

to increase power supply flexibility and reliability is growing fast. In the last decade, research

and studies designate the advantage of using DC-DC converters in pulsed power applications

[12, 13]. However, generation of extra high voltages is still problematic due to components

hold-on voltage limits (such as capacitors, solid-state switches and etc).

Another solution is to combine power supplies. Variety of converter configurations have

been introduced for improving the power supply specifications such as output ripple, high

input voltage, output power ratings, etc [14, 15]. Parallel and series connections are one of the

well known combinations. Designing power supplies based on series or parallel connection are

widely employed for different applications [14, 15].

This paper proposes parallel and series connections of flyback converter modules to

develop power rating and rise time of the pulsed power supply using conventional low voltage

switches. It is to be noted that the proposed idea is applied to the secondary sides of the pulsed

power modules, while the primary sides are connected in parallel due to the current sharing

purpose. The proposed scheme is flexible in increasing the output power and voltage due to its

modular design. Flyback converter is selected due to its unique properties in pulsed power as

is discussed in the next section. Taking the advantage of the parallel connection of the primary

side, it is possible to employ low current rate switches.

The proposed approach is evaluated based on two separate experimentations. In the first

experiment the efficiency of parallel and series configurations theory are evaluated through

two experimental setups. To insure applicability of the proposed modular approach, in terms

of performance and higher number of the modules, generating 20kV output voltage using 10

Chapter 3

137

series flyback modules is presented. The evaluated results indicate the effectiveness of the

proposed method.

3.3. Topology

The proposed approach utilizes flyback converter, which is one of the well known

topologies in the power electronics [16]. Conventional flyback converter is usually preferred

as it is simple, has only one switch and magnetic component, is able to generate high voltage,

can provide multiple outputs, isolation, etc [16, 17]. But when it comes to pulsed power

applications, in addition to the above-mentioned features, it has some more advantages which

make it more suitable. Main features of a flyback converter for pulsed power applications are:

The transformer, in addition to electrical isolation and energy storage also steps down the

reflected voltage across the switch; therefore lower voltage rate switches are needed

unlike other topologies.



As the pulsed power applications mostly have R-C characteristics [12], a current source


Suitable for controlling the energy flow as it acts as both current source and voltage

source.

Below basic principle and operation modes of this converter are briefly described.

The behaviour of a flyback converter can be realized by modelling the transformer with a

simple equivalent circuit consisting of an ideal transformer, magnetizing inductance (Lm) and

leakage inductance (Ll). Fig. 3.1 shows a flyback converter including the simple model of the

transformer connected to a load. In this figure CO is the converter equivalent output

capacitance. This capacitance can get affected in the case of an R-C load. It is to be mentioned

that, to keep the analysis simple, following assumptions are made:

1. The switch and diode are ideal (voltage drop across them is zero).

2. The switch and diode output capacitances are zero.

3. Conduction and switching losses are negligible.

4. The transformer stray capacitances are negligible.

Chapter 3

138

Lm

Vs

C0 Load

1 : n

Ll

Vo

+

-


Basically a flyback converter transfers energy from a source into the transformer

magnetizing inductance when the switch is on, and then transfers the stored energy to the load

while the switch is off. The proposed approach operates in the DCM (Discontinues

Conduction Mode). The operating modes are briefly summarized for 4 different modes

(illustrated in Fig. 3.2) below:

Mode 1: In this mode the switch is in the on-state. As Fig. 3.2 depicts, the current flows

through Lm and, during this stage, the energy is stored in the inductor. This mode lasts

depending on the duty cycle of the PWM (Pulse Width Modulation) signal. The

relationship between Lm and Vs can be expressed as:

s

mm

s

mmsms

TD

IL

TD

ILV

t

iLV

.0.

0

(3-1)

where Vs is the dc supply voltage, Im is the maximum current, D is the duty cycle and Ts

is the switching period.

Mode 2: When the switch is turned off the magnetizing current circulates through the

primary side of the transformer and the diode in the secondary side is turned on. As the

switch is turned off the current flowing through Ll is decreased to zero, which induces

voltage spike across the switch according todt

diLv ll . This voltage may damage the

switch if it exceeds the switch break down voltage level. To overcome this problem a

snubber can provide a path for this current and damps the spike to protect the switch

[18] or Ll can be reduced by employing optimum transformer design. This transient

Chapter 3

139

state is considered as one of the operating modes due to its importance (protecting the

switch) and understanding the procedure of the next operating mode.

Mode 3: The stored current in the magnetising inductance flows fully to the secondary side

of the transformer and charges the output capacitor. At this stage the converter is acting

as a current source. The maximum voltage across the switch at this stage is:

maxmax1

)( osswitch vn

Vv (3-2)

where, vomax is the maximum output voltage level and n is the transformer turns ratio.

Mode 4: As the converter operates in DCM, in this mode all the stored energy in Lm is

completely discharged to the capacitor, causing the diode to get turned off. Here the

converter is a voltage source and a high level of the voltage is applied to the load.

Ll

Lm

Vs

1 : n

LoadC0

Vs Vs

Mode 1

Mode 3 Mode 4

1 : n

C0

Mode 2

Ll

Lm

Vs

Load

Lm Lm

Ll Ll

1 : n 1 : n

C0 C0Load Load

Vo

+

-

Vo

+

-

Vo

+

-

Vo

+

-

Fig. 3.2 Operating modes in a flyback converter.

Chapter 3

140

3.4. Proposed Method

High voltage pulses with fast rise-time are the two main important features of a pulsed

power supply [1-3]. Fig. 3.3 illustrates the proposed method, which employs the advantage of

parallel and series configuration of the pulsed power modules to increase the voltage level and

the rise-time, while using the low-voltage switches. As the figure shows, the parallel and

series connections are only considered for the secondary side of the converter.

Ll

Lm

Vs

1 : n

Co2

Ll

Lm

n : 1

Co1Load

Vo

+

-


(a)

Ll

Lm

Vs

1 : n

Co1

Load

Ll

Lm

1 : n

Co2

Vo

+

-

Converter (2)

Converter (1)

(b)


Chapter 3

141

3.4.1. Single Module

To understand and compare the parallel and series configuration, first a single module

characteristic is described. As mentioned, a flyback converter can operate as a current source.

CoIo Vo

+

-

Load


Therefore, circuit schematic shown in Fig. 3.1 is simplified as in Fig. 3.4. Comparing these

two figures, the estimation of the output voltage can be summarized as below:

First, as described earlier, the magnetizing inductance is charged by the DC power supply.

The charged energy stored in the inductor Lm under ideal situation can be expressed as:

2

2

1mmpri ILE (3-3)

Regarding (3-1) and (3-3) it is possible to control the energy flow in each pulse by limiting

the Im which can be done by varying the duty cycle or the input voltage. This stored energy

will be transferred to the output capacitor when the switch is in the off-state. The capacitive

stored energy is:

2sec

2

1ooVCE (3-4)




o

mmo

C

LIV (3-5)

The rate of rise for the generated voltage is defined as dv/dt. When the switch is turned off,



Chapter 3

142

o

m

o

o

nC

I

C

I

dt

dv

(3-6)



The idea of parallel and series configurations of the pulsed power modules is inspired by

both equations (3-5) and (3-6), as these equations show the effect of stored current level,

magnetising inductance and the output capacitor on the generated output voltage and the rise-

time.

3.4.2. Parallel Modules

Considering Fig. 3.3(a) and the fact that each transformer acts as a current source, the

stored energy in the primary side is doubled when two modules are connected in parallel. This

stored energy will be transferred to the output capacitors; hence considering (3-3) and (3-4)

the output voltage magnitude and its rate of rise are as follows:

op

mmparallelo

C

LIV 2

(3-7)

op

o

Parallel C

I

dt

dv 2

(3-8)

where Cop=Co1+Co2. Equations (3-7) and (3-8) indicate the importance of the output capacitor.

Even if the modules are paralleled, the output capacitance can affect the output voltage level

and rise-time and can keep the performance same as a single module. Therefore, the parallel

configuration is beneficial in pulsed power applications when Cop=Co. This happens in the

case of R-C load when the load capacitance is much bigger than the power supply output

capacitance.

3.4.3. Series Modules

Second alternative is series connection of the pulsed power modules, as illustrated in Fig.

3.3(b). Here, the injected energy to the output is also doubled. But as described, the generated

voltage features also depend on the output capacitance. Taking into account the equations (3-

3) to (3-6), the output voltage magnitude and its rate of rise can be expressed as:

Chapter 3

143

os

mmSerieso

C

LIV 2 (3-9)

os

o

Series C

I

dt

dv

(3-10)

where Cos is o2o1

o2o1

CC

CC

. In series modules, same as in parallel modules, the output voltage

level and rise time improve as the output capacitance decreases. Here the series modules are

beneficial in both level of voltage and its rate of rise when Cos<Co.

In pulsed power applications two different cases may occur. One is when the load

capacitance is lower than the output capacitance of the pulsed power supply. In this situation

series modules exhibit better performance as illustrated in Table 3.1. The second case is when

the load has much higher capacitance, therefore all series, parallel and single modules have

same output capacitance. Under this circumstance the parallel connection has better

performance (see Table 3.2).

Hence, depending on the load capacitance one of the modules is beneficial to use.

Regarding the voltage level and rate of rise as presented in Table 3.1 and 3.2, it is possible to

generate wide range of voltage levels and improve dv/dt by connecting N modules in series or

parallel. This feature increases the flexibility of the pulsed power supply in varied

applications.



dt

dv

Parallel Singleo

o

mm V

C

LI

Singleo

o

dt

dv

C

I

Series Singleo

o

mm NV

C

LNI

Singleo

o

dt

dvN

C

IN



dt

dv

Parallel Singleoo

mm VN

C

LIN

Singleo

o

dt

dvN

C

IN

Series Singleo

o

mm VN

C

LIN

Singleo

o

dt

dv

C

I

Chapter 3

144

In addition, regarding (3-1) and the calculated output voltages in Table 3.1 and 3.2, the output

voltage can be easily adjusted for the required level whether by Vin or D. For example (3-11)

shows the output voltage of series module based on the results presented in Table II as:

.TDCL

VNV

om

ino (3-11)

Usually this adjustment is performed by limiting the current by selecting suitable duty cycle

(D). But Im should be calculated in a way that Im<Isat, where Isat is the current saturation level

of the transformer.

As Fig. 3.3(b) shows, in the series connection each switch withstands its own module

reflected output voltage, while in the parallel connection; each switch should tolerate the

whole reflected output voltage. This fact makes the series connected modules more

appropriate for generating high voltage waveforms using low voltage switches. Therefore, (3-

2) can be rewritten as below for N-series module:

maxmax1

)( osswitch vNn

Vv (3-12)

By comparing Fig. 3.3(a) and (b) it is obvious that the idea of parallel and series

connections of the flyback modules is applied to the secondary side, while the primary sides

for both cases are paralleled. The main advantage of paralleling the primary side of the pulsed

power modules are: current sharing so the proposed idea is applicable for high power

applications, reducing the number of the power supplies to one at the input side and ability to

employ low current rating switches.

3.4.4. Operating Conditions


designing procedure of the pulsed power modules. These are:

1) The output capacitor should be completely discharged during each period; otherwise the

whole stored energy in the magnetising inductor will not transfer to the capacitor.

2) Synchronization of each switch gate signal is required. Delays between each module gate

signal reduce the performance of the system. This is important when higher number of

modules is used.

3) The damping factor (ζ):

Chapter 3

145

As shown in Fig. 3.4, the entire circuit acts as a parallel RLC circuit (the current source is

an inductor). Hence, the output voltage and the rise time are the results of the resonance effect

of this circuit. The output voltage can be expressed as:

tstso eAeAtV 21

21)(

(3-13)


20

22,1 S

(3-14)

here, RC2

1 is known as neper frequency and

LC

1


0)0(

toV and oItCI

)0( are the primary conditions. I0 is the stored current in the

magnetizing inductor (Im=nI0) transferred to the secondary side. The coefficients A1 and A2

are determined by the primary conditions as:

02

02

20

212

LIAA

(3-15)


Underdamped ζ <1 or α < 0 , C

L

R2

1

C

LIA

2

01

(2-16)

Overdamped ζ >1 or α > 0

01 RIA

(2-17)

Equation (3-16) shows that if the circuit is underdamped, the generated voltage amplitude

is independent of the load impedance, which is the situation in which the proposed method is

valid. But under overdamped condition as in (3-17), the output voltage amplitude depends on

the load impedance and the initial current. Hence, in overdamped condition, the parallel

connection has better performance.

In Fig. 3.5, the effect of damping factor on the output voltage and rise time is illustrated. A

decreasing damping factor results in a higher voltage level and faster rise time.

Chapter 3

146


Considering the above mentioned features, this effect should be considered especially in

low impedance applications such as the liquid discharge. Therefore, regarding the mentioned

characteristics, the proposed method is more beneficial for high impedance-capacitive load

such as DBD (Dielectric Barrier Discharge) loads.

3.5. Experimental Results and Discussion

In this section the proposed idea has been evaluated based on practical experimentations. In

the first set of experimental results the idea of parallel and series configurations is evaluated

based on two flyback modules. In the second part, in order to assess the proposed method for

higher number of the modules, the obtained results of a high voltage prototype based on 10

series flyback modules is presented.

3.5.1. Evaluating Parallel and Series Configurations

Two laboratory prototypes for parallel and series connections, based on single module

flyback converter are implemented, to investigate performance of the proposed method

practically. Fig. 3.6 shows the experimental hardware setup for two flyback modules. Here

600V IGBT modules, SK25GB065, are used as power switches. Semikron Skyper 32-pro gate

drive modules are utilized to drive the IGBTs and provide the necessary isolation between the

switching-signal ground and the power ground. Four 1000 V diodes, STTH3010, are

Chapter 3

147

connected in series for each module. A Texas Instrument TMSF28335 DSC (Digital Signal

Controller) is used for PWM signal generation. Two step-up transformer with an UU100 core

3C90 grade material ferrite from Ferroxcube, are designed with N1 = 4 and N2 = 40.

Magnetizing inductance and leakage inductance of each transformer is approximately equal to

152µH and 1.6µH, respectively. Here a 5nF capacitor is placed across the switch to damp the

voltage spikes caused by leakage inductance. The circuit is implemented with Vs = 17V, fs = 1

kHz, 15% duty cycle and resistive load of 20kΩ.

To verify the proposed method and the theoretical analysis, experimental evaluations under

two different conditions (Case 1 and 2) are carried out. Both cases have been conducted

regarding the conditions presented in Table 3.1 and 3.2, respectively.

Fig. 3.6 Hardware setup for two flyback modules.

3.5.1.1. Case1


supplies is compared with the single one. Therefore, the output capacitor of each module is

equal (2.35nF for each module). So the equivalent capacitance of the series and parallel

modules are equal to 1.175nF and 4.7nF, respectively.

Chapter 3

148

Fig. 3.7 depicted the measured results. Fig. 3.7(a) shows the current sharing at the primary

sides. Due to parallel connection of the primary sides, in both parallel and series

configurations, the current is approximately equally shared between the two modules. As

mentioned before, parallel connection at the primary sides makes the proposed method

independent from using high current rate switches.

The voltage pulses, in Fig. 3.7(b), illustrate the better performance of series connection

over the other connections in both voltage level and rate of rise. In series modules the

maximum voltage level and rate of rise are 4.02 kV and 608 V/µs, respectively. While the

parallel and single modules achieved approximately same level of the voltage (2.37 kV) and

rate of rise (304 V/µs). As can be seen, the series modules performance is around 1.7 and 2

times better than the other modules in voltage level and rate of rise, respectively. From the

aforementioned theoretical analysis the performance of the series modules in voltage level

expected to be 2 times better, this difference is due to the resonance and dissipation happening

in the practical circuit. As the rate of charging is in terms of a time constant RC; hence, as

illustrated in the figure, series connection discharging time is shorter than the parallel and

single modules.

The voltage across the switch, shown in Fig. 3.8, in series connection is same as the other

modules, while series module generates higher level of voltage. This shows ability of the

series modules in generating high voltages with low voltage switches (see (3-12)). However,

as the figure shows the resonance appeared across the switch should be considered too. This

issue which mainly causes by stray capacitances and inductances of the transformer can be

reduced by proper transformer design and fabrication.

Chapter 3

149



(a)

(b)

Fig. 3.7 (a) current sharing at the primary side for both parallel and series connections, (b) output voltage

waveform of single, parallel and series modules in Case1 (Co1=Co2=Co).

Chapter 3

150

Fig. 3.8 Voltage across the switches in Case1.

Fig. 3.9 depicts voltage stress across the output diodes. As can be seen the reverse blocking

voltage of the diodes in series connection modules are less than the other ones while it is

operating at higher voltage level. This is due to the fact that each module in series connection

sees the output voltage divided by the number of the modules. This is another advantage of

series connection, because in the parallel connection module’s diode should withstand the

whole generated output voltage. It is also important to consider the current rating of the diodes

for high power applications.

Fig. 3.9 Voltage across the diodes in Case1.

3.5.1.2. Case2

If the load capacitance is much higher than the power supply output capacitance, then all of

the single, parallel and series modules will have the same equivalent output capacitance. In

this case the output capacitance of all modules is considered to be equal to 4.7nF. As depicted

Chapter 3

151

in Fig. 3.10, single module maximum voltage level is 1.72 kV and both parallel and series

modules obtained same voltage levels (2.37kV). But the parallel connection has better rate of

rise (304V/µs) compared to the two other modules (162 V/µs).


Although in this case parallel configuration shows better performance, but as its circuit

components should with-stand higher level of voltages compared to the series connection, the

use of the series modules is more convenient.

3.5.2. High Voltage Modular Power Supply

To insure applicability of the proposed approach for a modular power supply, series

configuration is applied for 10 flyback modules. Fig. 3.11 illustrated the block diagram of the

implemented hardware setup. To decrease the number of switches, depending on the switch

power ratings, a set of modules (here 5) is controlled with one switch. The transferred energy

to the load can be controlled by monitoring the primary stored energy. This is done through

limiting the input current by changing the duty cycle of the PWM signal in (3-1). As Fig. 3.11

shows, a current sensor is used for monitoring the input current. The experimental hardware

setup for 10 series flyback modules is depicted in Fig. 3.12. 1700V IGBT modules

(SKM200GB176D) are used as power switches. Same gate drive modules and controller setup

are used for this experiment as the former one. Also same configurations for the transformers

are used except here the number of the cores per transformer is reduced to one. A HX10-NP is

used as the current transducer (CT).

Regarding the selected components each module is able to generate up to 4kV, which

makes the whole system capable of generating up to 40kV. The circuit is implemented with

Chapter 3

152

fs=1kHz and VS=10V and 10% duty cycle. Each module has a 470pF capacitor (CO) and a

1MΩ high power resistor was connected as a load.

VS

Lo

ad

Control

protocol of

switches

VLoad

+

-

Vo1

+

-

Vo2

+

-CO

CO

CO Vo3

+

-

CO Vo4

+

-

CO Vo5

+

-

Vo6

+

-

Vo7

+

-

Vo8

+

-

Vo9

+

-

Vo10

+

-

CO

CO

CO

CO

CO

Me

asu

rin

g th

e C

urr

en

t

Fig. 3.11 Block diagram of the implemented setup.

Fig. 3.12 Hardware setup for 10-series flyback modules.

Chapter 3

153


Fig. 3.13 shows the measured practical results when the generated voltage is applied to a

resistive load. As can be seen from Fig. 3.13 the voltage pulse of the series modules obtained

the peak amplitude of 20.8kV, while the single module has the peak voltage of 2.15kV. The

measured output voltage shows the rate of voltage rise of 8kV/µs in the series connection. The

peak voltage level and rate of voltage rise indicate approximately the 10 times (number of

series modules) better performance in comparison with a single module.

3.6. Conclusion

This paper demonstrates the advantages of parallel and series configuration of flyback

converters for pulsed power applications. The proposed method aims at increasing the voltage

level and rise time of the generated pulses while emphasizing on the modularity concept. By

employing parallel and series connections it is possible to generate wide range of voltage

levels with improved rate of rise.

In this method the flyback converter topology is employed as it shows beneficial

characteristics especially for pulsed power applications. The proposed method was evaluated

through two different hardware setups. Two prototypes capable of generating up to 4 kV to

prove the proposed parallel and series configurations theory were used as the first experiment.

In the second experiment the idea of modularity and employing higher number of the modules

were evaluated, by implementing a series connection of 10 flyback modules and tested up to

20kV. The experimentations and analysis demonstrated that the proposed scheme is beneficial

Chapter 3

154

for high impedance-capacitive loads. Generally series modules show better performance, as

the parallel modules has more constraints due to its circuit components blocking voltage level.

3.7. References

[1] H. Bluhm, Pulsed Power Systems: Principle and Applications. Berlin Heidelberg: Springer-

Verlag, 2006.






[4] ABB, "High Power Semiconductors," in Short Form Catalogue, ed, 2003.

[5] ABB, "High Power Semiconductors," in Short From Catalogue, ed, 2011.

[6] ABB, "Asymmetric Integrated Gate Commutated Thyristor," ed, Jun. 2010.

[7] P. Elmer, "Triggered Spark Gaps, Ceramic-Metal ", ed, 2001.


Generator for Microplasma Applications," Plasma Science, IEEE Transactions on, vol. 33,

pp. 1205-1209, 2005.

[9] T. G. Engel and W. C. Nunnally, "Design and operation of a sequentially-fired pulse forming

network for non-linear loads," Plasma Science, IEEE Transactions on, vol. 33, pp. 2060-

2065, 2005.




1744-1750, 2006.

[11] D. Durga Praveen Kumar, S. Mitra, K. Senthil, A. Sharma, K. V. Nagesh, S. K. Singh, et al.,


Blumlein," Review of Scientific Instruments, vol. 78, pp. 115107-115107-4, 2007.




[13] S. Zabihi, F. Zare, G. Ledwich, and A. Ghosh, "A novel high voltage pulsed power supply

based on low voltage switch-capacitor units," in Pulsed Power Conference, 2009 IET

European, 2009, pp. 1-4.

[14] A. Cid-Pastor, L. Martinez-Salamero, C. Alonso, R. Leyva, and S. Singer, "Paralleling DC-

DC Switching Converters by Means of Power Gyrators," Power Electronics, IEEE


[15] B. R. Lin, H. K. Chiang, C. C. Chen, C. S. Lin, and A. Chiang, "Analysis and implementation

of soft switching converter with series-connected transformers," Electric Power Applications,

IET, vol. 1, pp. 82-92, 2007.



on, vol. 37, pp. 171-178, 2009.

Chapter 3

155



1189-1193.

[18] M. Peipei, W. Xinke, Y. Jianyou, C. Henglin, and Q. Zhaoming, "Analysis and design

considerations for EMI and losses of RCD snubber in flyback converter," in Applied Power

Electronics Conference and Exposition (APEC), 2010 Twenty-Fifth Annual IEEE, 2010, pp.

642-647.

156













A Flexible Solid-State Pulsed Power Topology

Published in proceedings of: 15th International Power Electronics and Motion Control Conference, EPE-

PEMC 2012 ECCE Europe, Novi Sad, Serbia.


Pooya Davari Proposed the initial design and control strategy. Develop the control strategy

regarding the simulation and data analysis. Wrote the manuscript.

22 Feb 2013

Firuz Zare Proposed the initial idea and supervised the simulation process and writing the

manuscript.

Arindam Ghosh Aided data analysis and writing the paper



Prof. Arindam Ghosh


157

Chapter

4 . A F l e x i b l e S o l i d - S t a t e P u l s e d P o w e r

T o p o l o g y



Presented and Published at: 15th International Power Electronics and Motion Control

Conference, EPE-PEMC 2012 ECCE Europe, Novi Sad, Serbia, Sep. 2012.

4

Chapter 4

158

Abstract — Advances in solid-state switches and power electronics techniques have led to the

development of compact, efficient and more reliable pulsed power systems. This paper

proposes an efficient scheme that utilizes modular switch-capacitor units in obtaining high

voltage levels with fast rise time (dv/dt) using low voltage solid-state switches. The proposed

pulsed power supply has flexibility in terms of controlling energy and generating broad range

of voltage levels. The energy flow can be controlled as the stored energy can be adjusted by a

current source utilized at the first stage of the system. Desirable voltage level can be obtained

by connecting adequate number of switch-capacitor units. Moreover, the proposed topology is

load independent. Therefore it can easily supply wide range of applications especially the low

impedance ones. The effectiveness of the proposed approach is verified by simulations.

4.1. Keywords

Pulsed power supply, Current source, High voltage generator, Power Converter control

4.2. Introduction



Generally, a pulsed power system consists of three main parts: energy storage, pulse generator

and load. The most prominent part of a pulsed power system is the pulse generator, which is

based on utilized switch and topology. Hence, the switch is the connecting device between the

storage and the load. This means that the whole system characteristics such as rise time,

repetition rate, voltage rating, efficiency, cost, life time, etc, depend on the employed switch

specifications.







system [2], [3]. On the other hand, solid-state switches are compact, reliable, cost effective,

and have a long lifetime and repetition rate. The main drawbacks of solid-state switches are

their limited power rating and operation speed [2], [3].

Chapter 4

159


speed) and exploiting power electronics techniques and topologies have led to compact,

efficient and more reliable pulsed power systems. The new developed solid-state switches

such as Insulated-Gate Bipolar Transistor (IGBT) and Integrated Gate-Commutated Thyristor

(IGCT) have high power rating [4], but their lower operation speed comparing with the gas-

state switches and high cost still put limits in the pulsed power supplies.

One way to increase the pulsed power supply performance and cover the switch limits is to

explore alternative circuit topologies. Varieties of circuit topologies such as Marx generators

(MG) [5], pulse forming network (PFN) [6], magnetic pulse compressors (MPC) [7] and

multistage Blumlein lines (MBL) [8] have been introduced. These topologies have been

widely used in pulsed power supplies, but complexity, inflexibility and inefficiency are their

main drawbacks [2], [9]. Interest in applying power electronics topologies and techniques to

increase power supply efficiency and reliability is growing fast. In the last decade, research

and studies designate the advantage of using converters in pulsed power applications [3], [9-

11].

Pulsed power applications present one of the most varied ranges of loads in terms of load

behavior and impedance. Hence, various load conditions adversely affect the power supply

flexibility. Regarding this, it is desirable to have flexible pulsed power supply that can cover

varied range of applications. This flexibility comes in terms of: adjustable output voltage,

controlling the flow of energy and repetition rate. Moreover, the generated voltage pulse

waveform must be load independent [11]. This is quite important especially in the low

impedance applications such as in water discharge.

To achieve all the mentioned features, this paper proposes an efficient scheme that utilizes

modular switch-capacitor units in obtaining high voltage levels with fast rise time (dv/dt)

using low voltage solid-state switches. The proposed pulsed power supply has flexibility in

terms of controlling input energy and generating broad range of voltage levels. The energy

flow can be controlled by adjusting the utilized current source at the first stage of the system

and desirable voltage level can be obtained by connecting adequate number of switched-

capacitor units. The proposed topology is able to operate independently from the load while

preserving high repetition rate. Moreover, the control algorithm is designed in a way that it

can prevent from any possible faults and protect the system from the over-voltage.


Chapter 4

160

S11

S12

S21

S22

Sn1

Sn2

SL

Load

Si SVVS

L

D1

D2

C1

C2

Cn

Posi

tive

buck

-boost

conver

ter

Load Switch

Control

protocol of

switches

Sw

itch

-cap

acit

or

unit

s

Hysteresis

current control

Hysteresis

voltage control

Ov

er-r

ide

sig

nal

Ov

er-r

ide

sig

nal

Voltage

Feedback

Synchronizing

PWM Signal

Fig. 4.1 Proposed pulsed power topology for n level switch-capacitor units.

4.3. Topology

Fig. 4.1 illustrates the proposed method with a brief controlling algorithm scheme. Generally,

as can be seen the proposed topology consists of three different stages. The first stage is a

positive buck-boost converter as a current source and the energy storage section. The next

stage is the switch-capacitor units, which generates the required voltage level and acts as a

pulse generator, and the last stage is the load.

In order to have a flexible pulsed power supply, here the positive buck-boost converter

control the amount of stored energy by employing appropriate duty cycle for Si and SV. The

desirable voltage level can be obtained by connecting adequate number of switch-capacitor

units in series. This feature also makes the system capable of taking the advantage of

employing low voltage switches, as lower voltage rating switches have intrinsically better

behavior [12].

Chapter 4

161

In addition to the common aforementioned three sections, this topology utilizes a load

switch for low impedance applications such as water decontamination in which it disconnects

the load from the pulsed power supply before the required voltage level is obtained. This

makes this topology load independent, because while the switch-capacitor units are charging it

is not connected to the load. In order to preserve a same break down voltage for the load

switch as the other switches, the switching signals need to be controlled precisely.

4.4. Control Strategy and Operating Modes

Effective algorithms are considered to define the turn on and off procedures of the power

switches. Fig. 4.2 and Fig. 4.3 illustrate the operating modes of the proposed topology

regarding the designed control algorithm.

4.4.1. Current & Voltage Control

The output current of the positive buck-boost converter need to be adjusted on a suitable

level. A hysteresis control is used for stabilizing this parameter. This control technique is

selected due to its simplicity, robust performance and stability [13, 14].

The hysteresis control determines the duty cycle of Si by comparing the inductor current

with the predefined reference current. The hysteresis control performs based on the upper and

the lower bands. When the inductor current iL reaches to the upper-band Si turns off and when

it reduces down to the lower-band it turns on. Here the hysteresis bands are selected in such

way that the converter operates in CCM (Continues Conduction Mode). Therefore this

converter operates as a current source for the switch-capacitor unit.

To charge switch-capacitor units equally, the output voltage needs to be adjusted on an

appropriate level. This can be done same as the current control by utilizing the hysteresis

method. Here the Vref is selected as the controllable voltage. When Vref increased to the upper-

band then the SV turns on and when it reduces down to the lower-band it turns off.

Considering the mentioned control procedure the operating modes can be depicted as in

Fig. 4.2. In the first mode, Fig. 4.2a, both Si and SV are in the on-state. During this mode the

inductor L starts to store energy. The required charging time can be calculated as:

S

LLSL

V

iLt

t

iLVV

(4-1)

At this stage the required energy based on the pulsed power application can be estimate as:

Chapter 4

162

2

2

1LLiE (4-2)

Therefore, it is possible to control the energy flow by selecting proper values for iL and L.

Si SVVS D1

C R

VrefL

iL

D2

(a)

Si SVVS

L

D1

D2

C R

Vref

iL

(b)

Si SVVS

L

D1

D2

C R

Vref

iL

(c)

Fig. 4.2 Operating modes for positive buck-boost converter.

In the voltage control algorithm SV turns on when iL drops below the desired level, this

override signal speed up the inductor charging process. This is why in the first mode SV is in

the on-state. This mode last when iL reaches to the upper-band of the current hysteresis

control.

In the second mode, Fig. 4.2b, the inductor is charged to the required level. Hence, Si and

SV will turn off and both diodes D1 and D2 turn on in order to transfer the stored energy to the

output which is the capacitor units.

In the last mode Vref reaches to its upper limit when the capacitor unit is charged up to the

required level. Here, SV turns on as shown in Fig. 4.2c. As the proposed topology is designed

in a way that each switches breakdown voltage should be equal to one switched-capacitor unit,

therefore in order to protect this switch, SV also turns on prior to connecting all switch-

capacitor units in series and supplying the load.

Chapter 4

163

4.4.2. Charging & Load Control

The charging process is synchronized with a PWM (Pulse Width Modulation) signal, which

also determines the repetition rate of the pulsed power supply. Here the procedure is explained

for two switched-capacitor units (Fig. 4.3) which can be extended to n units. The procedure is

controlled by Sn1 and Sn2, which n is the capacitor unit number.

When the PWM signal is high, the charging process begins with the first switch-capacitor

unit. At this mode (Fig. 4.3a), S22 turns on and the stored energy form the current source starts

charging the capacitor of the first unit via anti-parallel body diode of S11. When the capacitor

voltage reaches to the required level the second mode (Fig. 4.3b) starts by turning S12 on and

S22 off.

When the last unit charged up (here the second unit), SL and SV should be turned on prior to

applying all units voltages to the load (turning on the high-side switches). The reason is to

protect these switches from over voltage, as each switch blocking voltage is considered to be

equal to one capacitor unit voltage. As can be seen from Fig. 4.3c in the third mode the load is

connected to the last unit and the current source is disconnected as SV is in the on-state.

Depending on the load impedance and its time constant, , in the worst scenario case a

low impedance load may fully discharge the last unit.

After SL and SV turned on, the forth mode occurred by turning on the high side switches of

the capacitor units. At this moment, as Fig. 4.3d depicted, all of the capacitors connected in

series and a high voltage is applied to the load. Due to the last unit discharging in the previous

mode in the worst case the output voltage is:

Cout VnV )1( (4-3)

where n is the number of switched-capacitor units and VC is the voltage of each unit.

Regarding (4-3) the voltage drop issue can be solved by increasing the number of the units.

Finally, when the PWM signal is low all of the high-side and low-side switches turn off and

on, respectively and the voltage across the load drops to zero (Fig. 4.3e).

The reason that the last unit should be connected to the load prior to the rest of the units is

due to the presence of SL. To protect SL this switch needs to turn on before supplying the high

voltage to the load. Before supplying the load the low-side switches are in on-state (the output

is short circuit). Practically a switch can turns on when it has been biased enough. Hence, the

last unit gets connected to provide a voltage across SL, so it can turn on.

Chapter 4

164

The load switch (SL) is considered in the proposed topology to make load independent. In

the other word, SL causes high impedance across the capacitor units when the load has low

impedance so the load impedance (especially in low impedance applications) doesn’t affect

the system. Hence, for high impedance loads there is no need to place SL and therefore there is

no need to connect last unit prior to connection of the other units.

The break-down voltage of the employed components is equal to one capacitor unit except

D2. Therefore, the control algorithm should design in a way to prevent from any overvoltage

across the components. In practical implementation, for generating high level of voltages, D2

can be several series connected diodes.

S11

S12

SL

LoadC1

S21

S22

C2

S11

S12

SL

LoadC1

S21

S22

C2

S11

S12

SL

LoadC1

S21

S22

C2

S11

S12

SL

LoadC1

S21

S22

C2

S11

S12

SL

LoadC1

S21

S22

C2

(a) (b)

(c) (d) (e)

Fig. 4.3 The charging and supplying the load operating modes.

Regarding the mentioned charging and supplying the load process two different control

algorithms can be employed. The main difference between these two algorithms is the way

that the voltage feedback is provided.

One possible way to control the whole process is to provide individual voltage feedback

from each capacitor unit. Hence, the charging process of each unit can be started by

monitoring the previous stage charging level (Vch). Finally, when the unit’s capacitors charged

to the required level (VR) then it can be applied to the load. Fig. 4.4 illustrated the algorithm

flowchart. The advantage of this algorithm is the ability of preventing from any possible faults

Chapter 4

165

and indicating the failed unit as each unit’s voltage is measured. But the main drawback is

having multiple voltage feedback which increases the hardware implementation complexity.

The second potential way for the control algorithm is based on the measured voltage across

the output. The simplicity is the main advantage of this algorithm comparing with the previous

one, as it only employs one voltage feedback. Fig. 4.5 shows the second algorithm flowchart.

As can be seen after charging each unit’s capacitor, low-side switches should turn on in order

to short circuit the output. Hence, at each charging stage the measured output voltage

corresponds to only one unit. By short circuiting the output after each charging stage and using

a flag (h in Fig. 4.5) it is possible to control the charging and supplying the load process.

SK1: off

SK2: onK = 1,..., n

h =1

Start

If Sync

PWM=1

Yes

Yes

No

Sh1 & Sh2: off

Sg1: off

Sg2: on

g = 1,..., n – h

SL: off

If Vch=VR

NoYesIf VcK>VR

K =1,..., n

NoIf h = n

Yes

No

h = h + 1

Yes

SL & SV: on

Sn1: on

Sn2: off

SK1: off

SK2: on

K = 1,..., n-1 Delay

SK1: on

SK2: off

K = 1,..., n

If Sync

PWM=0

No

Charging unit h

SK1: off

SK2: on

Si: off

Fault detection

Supplying the Load

Fig. 4.4 Charging and supplying the load control algorithm flowchart (based on individual voltage feedback of

each unit).

Chapter 4

166

SK1: off

SK2: onK = 1,..., n

h =1

Start

If Sync

PWM=1

Yes

Yes

No

Sh1 & Sh2: off

Sg1: off

Sg2: on

g = 1,..., n – h

SL: off

If Vout=VR

NoYesIf Vout>VR

K =1,..., n

NoIf h = n

Yes

No

h = h + 1

Yes

SL & SV: on

Sn1: on

Sn2: off

SK1: off

SK2: on

K = 1,..., n-1 Delay

SK1: on

SK2: off

K = 1,..., n

If Sync

PWM=0

No

Charging unit h

SK1: off

SK2: on

Si: off

Fault detection

SK1: off

SK2: on

K = 1,..., n

Supplying the Load

Fig. 4.5 Charging and supplying the load control algorithm flowchart (based on only one voltage feedback).

The main advantages of the second algorithm are the ability to prevent from any fault (as the

output voltage is measured) and convenient practical implementation.

One of the important issues is to protect the circuit against the failures. These failures

happen if switch/switches stop operating, which can leads to over-voltage. As both described

algorithms monitors output voltage so they have the ability to prevent from any faults by

stopping the charging process. As an example, Fig. 4.4 and 4.5 illustrate one way of detecting

system failures.

Another issue needs to be pointed out is the duty cycle of the PWM signal, which should

adjusted in a way that it covers the whole charging process. To calculate the required pulse

width, let’s consider an ideal system. In an ideal situation the current source is charging each

capacitor unit with a constant current. Therefore without considering any losses:

Chapter 4

167

LC it

vC

dt

dvCi

(4-4)

here ∆v is equal to the voltage level that each switched-capacitor units need to charge up.

Considering the time require for charging the capacitor, ∆t, and the number of the units the

minimum pulse width can be expressed as:

100

T

tD

i

vCnt

L

(4-5)

where n is the number of the units, D is the duty cycle and T is the PWM signal period.

4.5. Simulation Results and Analysis

In this section the proposed topology with three switched-capacitor units (n=3 in Fig. 4.1)

is simulated in order to verify the performance of the proposed method. The software that was

used for simulation was MATLAB 7.10. Table 4.1 shows the parameters value used in

simulation corresponding to Fig. 4.1. The upper and lower bands for adjusting the current are

selected as 20.5A and 19.5A, respectively. The charging level is selected as 1000V, and to

control the voltage of each unit on 1000V the bands are selected with 2V variations and fS is

the frequency of the synchronizing PWM signal. The delay time as mentioned in Fig. 4.4 and

4.5 is selected as 1µs.

The simulation results and analysis are presented for three distinguished cases. In Case 1

the output voltage for three different loads are depicted. The switching signals and switched-

capacitor unit voltages are considered for one cycle in Case 2. Finally, the voltage stresses

across the critical components are depicted in Case 3.

4.5.1. Case1

Here the effects of different loads impedance on the generated output voltage are depicted

in Fig. 4.6. For high impedance loads the considered delay for turning on the SL and SV

doesn’t affect the output voltage level, but as the load impedance decreases the voltage drop

across the output voltage increases. This is due to the last unit capacitor discharging based on

the For the low impedance applications this issue can be solved by increasing the

number of the switched-capacitor units.

Chapter 4

168

4.5.2. Case2

In this case the switching signals for the charging process and the generated output voltage

are demonstrated for 1kΩ load. Fig. 4.7 completely demonstrates the charging and supplying

processes. As depicted during the charging process all of the high-side switches are in the off-

state. As can be seen, the last unit S31 turns on prior to the other switches. Finally because the

PWM signal becomes low, the output voltage drop to zero before the load fully discharges the

capacitors.


VS L C1,2,3 D fS iL VC1,2,3

100V 1mH 10nF %0.5 1kHz 20A 1000V

Fig. 4.6 The effect of the load impedance on the generated output voltage.

4.5.3. Case3

As mentioned before, in order to benefit from low voltage switches the break down voltage

of switches are selected to be equal to one capacitor unit voltage. Therefore, it is quite

important to turn on and off SL and SV effectively, otherwise they should tolerate the whole

output voltage. As Fig. 4.8 shows, following the proposed switching strategy the voltage

Chapter 4

169

across these two switches never exceeded from 1 kV. The only component in this topology

which must handle the whole output voltage is D2.

Fig. 4.7 Voltage waveforms with relative switching signal patterns.

Fig. 4.8 Voltage across the critical components.

Chapter 4

170

4.6. Conclusion

This paper demonstrates a flexible high voltage pulsed power topology. Various voltage

levels can be obtained by connecting different numbers of switched-capacitor units. The

energy flow can be controlled via a positive buck-boost converter. The repetition rate can be

adjusted using a synchronizing PWM signal. Moreover, the proposed topology is load

independent, as it is disconnected from the load during the charging process. This feature

makes the proposed method suitable for supplying wide range of applications specially the one

with low impedance such as water discharge.

The proposed method is evaluated under different circumstances and obtained simulation

results indicate the effectiveness of the proposed approach.

4.7. References

[1] Hansjoachim Bluhm, Pulsed Power System: Principle and Applications, Berlin Heidelberg,

Springer-Verlag, 2006.

[2] E. Schamiloglu, R.J. Barker, M. Gundersen, and A.A. Neuber, "Modern Pulsed Power: Charlie

Martin and Beyond," Proceedings of the IEEE , vol.92, no.7, pp. 1014- 1020, July 2004.



IEEE Transactions on, vol.14, no.5, pp.1051-1064, October 2007.

[4] ABB, High power semiconductors, Short form catalogue, 2003.


Generator for Microplasma Applications," Plasma Science, IEEE Transactions on , vol.33, no.4,

pp.1205- 1209, Aug. 2005.

[6] T.G. Engel and W.C. Nunnally, "Design and operation of a sequentially-fired pulse forming

network for non-linear loads," Plasma Science, IEEE Transactions on, vol.33, no.6, pp. 2060-

2065, Dec. 2005.

[7] Choi. Jaegu, T. Yamaguchi, K. Yamamoto, T. Namihira, T. Sakugawa, S. Katsuki, and H.

Akiyama, "Feasibility Studies of EMTP Simulation for the Design of the Pulsed-P ower

Generator Using MPC and BPFN for Water Treatments," Plasma Science, IEEE Transactions on,

vol.34, no.5, pp.1744-1750, Oct. 2006.

[8] D. Durga Praveen Kumar, S. Mitra, K. Senthil, A. Sharma, K. V. Nagesh, S. K. Singh, J. Mondal,

A. Roy, and D. P. Chakravarthy, "Characterization and analysis of a pulse power system based on

Marx generator and Blumlein," Review of Scientific Instruments, vol.78, no.11, pp.115107-

115107-4, Nov 2007.


topology based on positive buck-boost converters concept," Dielectrics and Electrical Insulation,

IEEE Transactions on, vol.17, no.6, pp.1901-1911, Dec. 2010.


Parallel and Series Configurations of Flyback Converters for Pulsed Power Applications," Plasma

Science, IEEE Transactions on, 2012.

Chapter 4

171

[11] L. Redondo and J. F. Silva, "26 - Solid State Pulsed Power Electronics," in Power Electronics

Handbook (Third Edition), ed Boston: Butterworth-Heinemann, 2011, pp. 669-707.

[12] Honggang Sheng; Wei Shen; Hongfang Wang; Dianbo Fu; Yunqing Pei; Xu Yang; Fei Wang;

Boroyevich, D.; Lee, F.C.; Tipton, C.W.; , "Design and Implementation of a High Power Density

Three-Level Parallel Resonant Converter for Capacitor Charging Pulsed-Power Supply," Plasma

Science, IEEE Transactions on , vol.39, no.4, pp.1131-1140, April 2011.

[13] F. Zare, G. Ledwich, "A hysteresis current control for single-phase multilevel voltage source

inverters: PLD implementation," Power Electronics, IEEE Transactions on, vol.17, no.5, pp. 731-

738, Sep 2002.

[14] A.A. Boora, F. Zare, G. Ledwich, A. Ghosh, " A general approach to control a Positive Buck- Boost converter to achieve robustness against input voltage fluctuations and load changes,"

Power Electronics Specialists Conference, PESC 2008, pp.2011-2017, 15-19 June 2008.

172













Effect of Pulsed Power on Particle Matter in Diesel Engine Exhaust Using a DBD Plasma Reactor

Submitted to: IEEE Transactions of Plasma Science


Meisam Babaie Designed and developed the mechanical setup. Planned and conducted

experimental verification and wrote the paper.

Pooya Davari Proposed the initial idea and designed and implemented the pulsed power supply.

Planned and conducted experimental verification and wrote the paper.

22 Feb 2013

Firuz Zare

Proposed the initial idea and evaluation process. Supervised the data analysis,

experimentation and writing the paper.

MD Mostafizur Rahman Aided experimentation process

Hassan Rahimzadeh Aided experimentation process

Zoran Ristovski Aided data analysis, supervised experimentation process and writing the paper

Richard Brown Aided data analysis, supervised experimentation process and writing the paper



Prof. Arindam Ghosh


173

Chapter

5 . E f f e c t o f P u l s e d P o w e r o n P a r t i c l e M a t t e r

i n D i e s e l E n g i n e E x h a u s t U s i n g a D B D

P l a s m a R e a c t o r

School of Chemistry, Physics and Mechanical Engineering, Queensland University of Technology,




+ Danfoss Power Electronics, Graasten DK-6300, Denmark

Amirkabir University of Technology, Iran

Published in: IEEE Transactions on Plasma Science, Vol.41, No.8, pp. 2349-2358, Aug.

2013.

5

Chapter 5

174

Abstract— Non-thermal plasma (NTP) treatment of exhaust gas is a promising technology

for both nitrogen oxides (NOX) and particulate matter (PM) reduction by introducing plasma

into the exhaust gases. This study considers the effect of NTP on PM mass reduction, PM size

distribution and PM removal efficiency. The experiments have been performed on real exhaust

gases from a diesel engine. The NTP is generated by applying high voltage pulses using a

pulsed power supply across a dielectric barrier discharge (DBD) reactor. The effects of the

applied high voltage pulses up to 19.44 kVpp with repetition rate of 10 kHz are investigated.

In this paper, it is shown that PM removal and PM size distribution need to be considered both

together, as it is possible to achieve high PM removal efficiency with undesirable increase in

the number of small particles. Regarding these two important factors, in this research, 17

kVpp voltage level is determined to be an optimum point for the given configuration.

Moreover, particles deposition on the surface of the DBD reactor was found to be a significant

phenomenon which should be considered in all plasma PM removal tests.

5.1. Index Terms

Particle size distribution, Particle mass reduction, Diesel exhaust gas, Dielectric barrier

discharge (DBD), Pulsed power, Push-pull converter

5.2. Introduction

There is a continuous increase in the number of diesel engines in both stationary and

mobile application due to the lower operating cost, higher thermal efficiency, longer durability

as well as lower hydrocarbons (HC) and carbon monoxide (CO) emissions [1]. However NOX

and particulate matter (PM) emissions still remain the two main environmental concerns in

diesel engine applications. Studies focused on risk assessment have showed that high outdoor

NOX concentration observed in residential areas contributes to increased respiratory and

cardiovascular diseases and mortality [2]. Moreover, the health effects of diesel particulate

matter have been an area of concern for many years, due to both the chemical composition and

the particle size distribution [3]. The small particles are inhalable and penetrate deep into the

lungs where they are able to enter the bloodstream and even reach the brain [4, 5].

Up to now, several technologies have been applied for NOX and particulate treatment of

diesel engines. In recent years, application of non-thermal plasma (NTP) in exhaust gas

treatment has gained lot of interests [6-9]. NTP treatment of exhaust gas is a promising

technology for both NOX and PM reduction by introducing plasma inside the exhaust gases.

Chapter 5

175

In the non-thermal plasma, electrons have a kinetic energy higher than the energy

corresponding to the random motion of the background gas molecules. The intent of using

non-thermal plasma is to selectively transfer the input electrical energy to the electrons which

would generate free radicals through collisions and promote the desired chemical changes in

the exhaust gas. These reactions can be accomplished at a fraction of the energy which is

required in a thermal plasma system [10]. NOX, unburned hydrocarbons, CO, and PM will be

oxidized due to oxidation processes which happen by introducing plasma in the exhaust gas.

Applying pulsed power is one of the efficient ways to generate NTP. Pulsed power is the

rapid release of stored energy in the form of electrical pulses into a load, which can result in

delivery of large amounts of instantaneous power over a short period of time. Recently, solid-

state pulsed power has gained more interest as it is compact, reliable, has a long lifetime and

high repetition rate. In the last decade, research and studies established the advantage of using

power electronics topologies in pulsed power applications [11-13]. In this research a bipolar

pulsed power supply based on push-pull topology is implemented.

Diesel particulate matter (DPM) consist mostly of carbonaceous soot with minor

components of volatile organic fraction (VOF) from unburned fuel, lubricating oil, inorganic

compounds such as ash and sulphur compounds and metals including zinc from lubricating

oil [38]. DPM are the cause of a series of adverse effects on environment [39] and human

health [40-42]. Particulate formation begins with nucleation in the engine cylinder and

dilution tunnel, and is followed thereafter by agglomeration [43]. Most of the diesel

particulate matter mass is in the accumulation mode, whereas in terms of particle number,

most particles are found in the nucleation mode. More than 90% of diesel exhaust-derived

PM is smaller than 1 μm in diameter [44]. Most of the mass is in the 0.1–1.0 μm

“accumulation” size fraction, while most of the particle numbers are in the <0.1 μm “nano-

particle” fraction [45, 46]. Ultrafine particles have an aerodynamic diameter less than 100 nm

and are emitted in high number by compression ignition engines. Whilst ultrafine particles

do not contribute much to the total mass of particulate matter emitted from an engine, they

contribute greatly to the total number of particles. The particle size distribution of particulate

matter from compression ignition engines has become of increased concern since a study by

the Health Effects Institute demonstrated an increased number of nanoparticles emitted from

a 1991 Cummins engine, despite a reduction in overall particle mass, relative to an older

1988 Cummins engine [47].

Ultrafine particles can penetrate deep into the lungs where they are able to enter the

bloodstream and even reach the brain [4]. The respiratory health effects (in particular asthma)

Chapter 5

176

from particle emissions correlate strongly with particle number, rather than particle mass

emissions [48] . In 2014 the Euro VI regulation will be implemented and the number of

particles emitted by compression ignition engines (in addition to a new particle mass limit)

will be regulated. This demonstrates that particle number emissions are becoming a very

prominent issue in engine design and research [49, 50].

The main concern of this paper is to analyse the effect of pulsed power on PM mass

reduction and PM size distribution considering the pulsed power effects on ultra-fine particles

emitted from real diesel engine exhaust gas. In this research, a pulsed power supply based on

the push-pull inverter is developed to generate up to 19.44 kVpp across the DBD load. The

experiments were conducted at different voltage levels with fixed repetition rate of 10 kHz.

PM mass reduction, PM removal efficiency and PM size distribution are investigated by

evaluating the results obtained.

5.3. Experimental method

5.3.1. Experimental Setup

A schematic diagram of the experimental setup is shown in Fig. 5.1. Experiments were

conducted on a modern turbo-charged 6-cylinder Cummins diesel engine (ISBe22031) at the

Queensland University of Technology (QUT) Biofuel Engine Research Facility (BERF). The

engine has a capacity of 5.9l, a bore of 102 mm, a stroke length of 120 mm, a compression

ratio of 17.3:1 and maximum power of 162 kW at 2500 rpm. Particle number distributions

are measured with a scanning mobility particle sizer (SMPS) consisting of a TSI 3080

classifier, which pre-selects particles within a narrow mobility (and hence size) range and a

TSI 3025 condensation particle counter (CPC) which grows particles (via condensation) to

optically detectable sizes. The SMPS software increases the classifier voltage in a pre-

determined manner so that particles within a 10-500 nm size range are pre-selected and

subsequently counted using the CPC. The software also integrates the particle number

distribution to enable calculation of the total number of particles emitted by the engine at

each test mode. Gaseous emissions are measured with CAI 600 series gas analyses. CO2,

NOX and CO concentrations can be measured by this gas analyser, whereas particulate mass

emissions are measured with a TSI 8530 Dust-Track II.

Chapter 5

177

As depicted in Fig. 5.1, three way valves to control the exhaust gas path ways are

employed. With this configuration, it is possible to measure both gaseous emissions and

particles before entering the reactor and after leaving it by changing the three way valve

directions. CO2 was used as a tracer gas in order to calculate the dilution ratio. CO2 was

measured from the dilution tunnel with dilution ratios being calculated using the following

equation:

(5-1)

Laboratory background CO2 measurements were made before the commencement of each

test session. Every concentration measured after dilution should be modified by using a

dilution ratio.

Fig. 5.1 Schematic diagram of plasma treatment system developed at QUT engine lab.

Chapter 5

178

5.3.2. DBD Reactor

A conventional dielectric barrier discharge reactor was designed for the experiments. Fig.

5.2 shows a schematic of the reactor. As illustrated in Fig. 5.2, it consists of two concentric

quartz tubes. Both tubes are 400 mm long and have a wall thickness of 1.5 mm. The outside

diameters of inner and outer quartz tubes are 20 mm and 25 mm, respectively. Exhaust gas

passes through the gap between these two quartz tubes. Based on pre-designed geometry, the

discharge gap is 1 mm. The DBD is connected to the pulsed power supply using internal and

external electrodes. The internal electrode is a copper cylinder and the external electrode is

made by a copper mesh that wraps the exterior part of the DBD. The electrodes are placed in

the middle of the DBD load with the length of 100 mm. Both tubes are fixed by two Teflon

caps at each end. Exhaust enters the reactor at the angle of 45 degree and flows throughout the

gap and leaves the reactor with the same angle.

(a)

(b)

Fig. 5.2 DBD reactor in Solidworks: a) Schematic view, b) Cross-sectional view.

5.3.3. Bipolar Pulsed Power Supply

Fig. 5.3 shows a circuit schematic diagram of the pulsed power supply. As illustrated, it is

based on the push-pull inverter topology. The push-pull inverter contains two switches that are

driven with respect to ground. This is the main advantage of the inverter. This topology uses a

centre-tapped transformer which is excited in both directions. A step up transformer is used to

boost the voltage and achieve galvanic isolation.

Chapter 5

179

The main reason to use a push-pull topology is to generate bipolar output voltage.

Applying voltage periodically build-up charges across the electrodes, which can results in

arcing. In order to sustained non-thermal plasma and prevent from arcing, bipolar pulse

generation can employed for clearing charges [50]. Employing lower number of switches is

another advantage of push-pull inverter. The two switches S1 and S2 are switched alternately







resonance it is possible to obtain high frequency semi-sinusoidal waveform. A typical

measured output voltage of the employed pulsed power supply is depicted in Fig. 5.4.

S2

S1

Vin

DB

DNA

NB

NS

CS

CS

NA = NB << NS


The first portion of the output voltage waveform is the resonant circuit dominated by the

magnetizing inductance of the transformer and the capacitances of the transformer and DBD.

The period of this signal is approximately 11.2 µs. The second one is the resonance

happening during the switches off-state between the leakage inductance and the capacitances

of the transformer and DBD. The period of this signal is equal to 88.8µs. As can be seen from

Chapter 5

180

the figure the repetition rate is set to 10 kHz. It is quite important to generate symmetrical

waveform regarding to clearing charge purpose and avoiding transformer saturation.

Fig. 5.5 shows the experimental hardware setup for the pulsed power supply. Here 1200 V



switching-signal ground and the power ground. A Texas Instrument TMS320F28335 DSC

(Digital Signal Controller) is used for PWM signal generation. A centre-tapped step-up

transformer with an UU100 core 3C90 grade material ferrite from Ferroxcube, are designed

with NA = NB = 5 and NS = 293. Here a 470 pF capacitor (CS) is placed across each switch to

protect them against the voltage spikes. The output voltage is measured and captured using a

Pintek DP-22Kpro differential probe and RIGOL DS1204B oscilloscope, respectively.

Fig. 5.4 Typical measured output voltage of the employed pulsed power supply.

Fig. 5.5 Electrical hardware setup with the DBD load.

Chapter 5

181

5.4. Results and Discussion

5.4.1. Plasma effect on PM Size Distribution

The effect of plasma on emission treatment is considered for varied experiments based on

the aforementioned Cummins diesel engine. In all experiments engine speed and load are kept

constant at 40 kW (25 %load) and 2000 rpm, respectively. A portion of raw exhaust gas

directly from an iso-kinetic sampling port of the tailpipe was diluted with air and passed

through the reactor. Emissions concentration is measured before and after the applying pulse

to study the plasma effects. In addition, the median particle diameter which is another useful

parameter to study the effect of plasma technique is also measured. It is to be noted that all

illustrated results in each experiment have been obtained as an average over three consecutive

measurements.

The experiments were made by applying output voltage from 10 kVpp up to 19.44 kVpp.

However, the first effect of plasma was appeared at 15 kVpp following with optimum

operation at 17kVpp and finally high amount of small particle generation at 19.44 kVpp.

Hence, the measurements are reported in this section regarding to the mentioned three applied

voltage levels. Fig. 5.6 illustrates the measured results. The applied output voltages across the

DBD load for three different voltage levels of 15 kVpp, 17 kVpp, and 19.44 kVpp are

depicted in Fig. 5.6a. The measured output voltages show the rate of voltage rise of

2840V/µs.

The measured load currents are illustrated in Fig. 5.6b, which shows many narrow pulsed

current spikes occurring in each half-cycle of the applied voltage. The measured current at

19.44 kVpp comparing with the other applied voltages shows higher number of the micro-

discharges in the gap with much higher amplitude. This is due to the fact that the applied

voltage has reached to the value of the breakdown voltage, which depends on the gap

distance, dielectric material, repetition rate, and etc. Controlling the amplitude of current

discharges is quite important as it can directly affect the plasma reaction which is discussed

further. Fig. 5.6c shows the voltage stress across the switch during the switching transition.

As can be seen, due to employing a centre-tapped transformer, the peak-to-peak voltage

stress across the switch in a push-pull inverter is approximately two times of the input

voltage.

Chapter 5

182

(a)

(b)

(c)

Fig. 5.6 Measured electrical parameters: a) output voltages across the DBD load, b) output current, and c)

voltage stress across the switch.

Chapter 5

183

Fig. 5.7 DBD image.

Fig. 5.8 V-Q cyclogram of the DBD load as a basis of power consumption calculation.

In these experiments, the output voltage amplitude is controlled by changing the input DC

voltage between 72.4 V, 84.4 V, and 94.8 V. Under same situation, Fig. 5.7 shows an image

of DBD recorded at 19.44 kVpp. As can be seen, NTP is clearly occurring between the two

electrodes.

To measure the power consumption of the DBD load, the energy transferred to the DBD

load has been calculated by employing the Lissajous (V−Q) diagram [51, 52].To measure Q a

4nF capacitor is placed in series with the DBD reactor. Thus, by measuring the voltage across

the capacitor and multiplying it by its capacitance value it is possible to calculate Q. The

energy consumed by the DBD reactor for one cycle is calculated from the area of V−Q curve

for different experiments, where V is the measured voltage across the DBD reactor (see Fig.

5.8). Hence, by considering the employed repetition rate (frequency) it is possible to calculate

the average consumed power by the DBD load. It is to be noted that the series connected

capacitor is selected large enough to not to affect the DBD reactor capacitance. The relevant

equations are:

Chapter 5

184

dvtQW )( (5-2)

)(

)()(

tdV

tdQtC (5-3)

By substituting (3) in (2):

2

2

1)(CVdQ

C

tQW (5-4)

Therefore, considering (5-2) to (5-4) the averaged consumed power can be calculated as

below:

2

2

1)( CVfdvtQfWfPA (5-5)

The illustrated data in Fig. 5.8 clearly indicate that the DBD reactor power consumption

correspondingly increases with the applied voltage level. This can be also realized from (5-5),

which shows the relation between the power and the applied voltage. The calculated averaged

power consumption (PA) for the applied voltages of 15 kVpp, 17 kVpp, and 19.44 kVpp are

27.37 W, 36.54 W, and 55.17 W respectively. The higher averaged power at 19.44 kVpp can

be realized through the measured discharged current as it has occurred at higher amplitude

and higher number of the micro-discharges.

In the first experiment, the maximum voltage level (19.44 kVpp) is applied. To estimate

the deposition rate on the reactor surface, emissions in the reactor inlet (reactor inlet no pulse)

and reactor outlet (reactor outlet no pulse) without applying any pulse voltage are measured.

Finally, the pulsed power supply is applied across the DBD and the emissions in reactor

outlet are measured (reactor outlet with pulse). The same process is employed in all following

experiments.

Fig. 5.9 illustrates the particle size distribution at 19.44 kVpp. There is a considerable

amount of PM deposition on reactor surface which is likely related to small gap between the

tubes (1 mm). The median diameter in the reactor inlet is 70 nm, while in the reactor outlet is

decreased to 66 nm. This shows that larger particles deposited more on the reactor surface.

By applying pulsed power, as shown in this figure, the median diameter decreases

remarkably to 35 nm. This implies that lots of big particles have been oxidized or broken to

small particles by producing plasma inside the exhaust gasses at this voltage level.

Chapter 5

185

Fig. 5.9 Particle Size Distribution (2000rpm, 25%Load, 19.44 kVpp).

The peak value of particle number at the reactor inlet is around ⁄ .

This value declines to ⁄ at the reactor outlet due to deposition. By

applying pulsed power the graph peaks to 4.5 ⁄ which is approximately

four times bigger than the particle number at reactor outlet without any pulse. The number of

particles with diameter of less than 70 nm in the reactor outlet with applying pulse is higher

than the particle numbers at the reactor outlet without any pulse. This effect increases even

more at particle diameters less than 50 nm. At this level, the particle number at reactor outlet

surpasses the number of particles at reactor inlet. These findings imply that the 19.44 kVpp

pulse power at 10 kHz, increases the number of small particles considerably. The origin and

nature of these particles is still not clear and will be of interests in future investigations.

However, the effect of 19.44 kVpp on particle size can be understood regarding to the high

level of micro-discharges in the discharged current as depicted in Fig. 5.6b.

The effect of voltage level with 17 kVpp is considered in the second experiment. The

results obtained have been summarized in Fig. 5.10. The figure shows the median diameter

changes from 70 nm at reactor inlet to 78 nm at reactor outlet without any pulse, and then falls

to 75 nm at reactor outlet with applying pulse. This shows that the larger particles are

deposited and also removed by plasma selectivity compared to smaller particles. As can be

Chapter 5

186

seen, at this voltage level (17 kVpp) the number of small particles has not increased. This is an

important feature when compared with the previous experiment (19.44 kVpp).

The last experiment is conducted by applying 15 kVpp pulses. The measured results are

depicted in Fig. 5.11, which shows no growth in the number of small particles as well as

second experiment (17 kVpp). The maximum of PM concentration took place at around 71

nm. There is a small difference between PM concentrations with and without applying plasma.

Therefore, this level of voltage can remove particles at the same rate of deposition.

Comparing the values of median diameter in the reactor inlet and outlet shows that the

smaller particles are more likely to be deposited inside the reactor under the no pulse

condition. On the other hand, by applying the pulse the median diameter is in same range of

reactor inlet. There is no increase in the number of small particles at this voltage level. This

trend in median diameter variation is almost in complete agreement with the voltage of 17

kVpp.

Careful comparison of the reactor outlet particle size distribution, when there is no pulsed

power, indicates that the distribution in Fig. 5.9 shows a reduction in particle median

diameter, whereas Fig. 5.10 shows a slight increase in particle median diameter. Experiments

for Fig. 5.10 were conducted approximately 30 min after that of Fig. 5.9. Therefore, there is a

possibility that wall initial deposition occurred with larger particles and the later experiments

for Fig. 5.10 favoured slightly smaller particles due to the larger surface area of the wall and

the attraction of particles to deposited particles rather than the quartz wall alone.


Chapter 5

187


5.4.2. PM removal efficiency

PM removal can be calculated based on the following equation:

100)(

ionconcentratPMinlet

pulsewithionconcentratPMoutletionconcentratPMinletremovalPM (5-6)

Where the PM concentration unit is (particle/cm3) and PM removal is calculated for all

PM diameters.

Fig. 5.12 illustrates the PM removal at 19.44 kVpp. PM deposition on the reactor surface

increases with the PM size and gets to the maximum value of 70% removal at around 80 nm.

But below 80nm PM removal decreases again. For most of the particle sizes, PM deposition

on the reactor surface is more than 40%. When a 19.44 kVpp voltage is applied to the

reactor, PM removal efficiency reaches the value of 90% for larger particles. Removal

efficiency for particulate matter less than 80 nm is less than PM removal without applying

any pulse. This indicates undesirable operation of the plasma within this particle size range.

Also there is no removal for particles smaller than 50 nm. There is a high possibility that this

incense in particle numbers can be related to the following two factors: firstly, fragmentation

of larger particles by electron impact reactions or incomplete oxidation and secondly,

oxidation of gaseous exhaust emissions to particles by plasma generated ozone.

Chapter 5

188

Fig. 5.12 PM removal as a function of PM size (2000rpm, 25%Load, 19.44 kVpp)


Fig. 5.13 shows the PM removal at 17 kVpp which shows that PM removal by plasma is

more effective than deposition removal. This means that at this level of voltage, all deposited

particles and also some large particles inside the flow can be removed or oxidized. For

particles smaller than 35 nm, PM removal percentage by deposition is higher than PM

Chapter 5

189

removal by plasma. Apparently, smaller particles cannot be removed by plasma within this

range (<35nm). However another possibility can be dissociation of larger particles to smaller

ones by electron impact reactions.

PM removal efficiency at 15 kVpp is illustrated in Fig. 5.14. As can be seen the PM

removal for both graphs increases to an optimum value then decreases. The maximum PM

removals in reactor outlet with and without the pulse are 61% and 69% respectively. For

particles larger than the 60 nm diameter, PM removal when applying pulsed power is slightly

higher than PM removal without any pulse. However, for particles with smaller diameters,

these values are in the same ranges.

Regarding the results obtained, it can be concluded that the voltage level has an important

role in the size dependent removal efficiency. At 15 kVpp the particle size distribution has

been affected slightly. The PM removal without producing small particles can be improved

as the voltage increases to 17 kVpp. The increase in the number of small particles has been

noticed when the voltage level goes up to 19.44 kVpp, while larger particles have been

reduced considerably. By taking into account all the above mentioned features, the

experiment with 17 kVpp voltage shows better efficiency in terms of particle size distribution

for the given configuration.


Chapter 5

190

5.4.3. Plasma Effect on PM Mass Reduction

After studying the effect of different voltages on particle size distribution, in this section, the

effect of plasma on particle mass reduction is considered. In a similar way to the previous

sections three different voltage levels have been applied to the DBD load. All results obtained

have been summarized in Table 5.1. PM mass concentrations in the reactor inlet were 4.56

(mg/m3), 4.26 (mg/m

3) and 5.14 (mg/m

3) respecting to the variation of engine operating

conditions for the three tests conducted. These values decreased to 2.84(mg/m3), 3.68 (mg/m

3)

and 4.37 (mg/m3) at reactor outlet in the no pulse condition, respectively. This shows a

significant particle deposition inside the reactor. When the pulsed power is applied, plasma

PM removal occurred. This causes the PM mass concentration reduction of 43.9 %, 38.6% and

27.1% at 19.44 kVpp, 17 kVpp and 15 kVpp respectively.

The maximum PM mass reduction has been obtained when the voltage level is 19.44 kVpp.

However, according to the particle size distribution measurements, this voltage level increases

the number of small particles, which is not a desirable feature. The 17 kVpp applied voltage

shows a more suitable performance with good mass reduction of around 40% without any

increase in ultra-fine particle numbers. The 15 kVpp voltage level is found to be almost the

threshold breakdown voltage for the given configuration, below which no significant PM mass

reduction, PM removal, and PM size distribution which have not been affected too much.

Table 5.1 PM Mass Reduction at Different Voltage Levels

Applied Voltage

Measurement

19.44 kVpp 17 kVpp 15 kVpp

Reactor Inlet PM

Concentration (

)

4.56 4.26 5.14

Reactor Outlet No Pulse PM

Concentration(

)

2.84 3.68 4.37

Reactor Outlet By-Pulse PM

Concentration (

)

2.56 2.62 3.74

Plasma PM Removal

Efficiency (%)

43.9 38.6 27.1

Chapter 5

191

5.5. Conclusion

In this study the effect of non-thermal plasma obtained by applying high voltage pulses on

PM size distribution and PM mass reduction were investigated. It was found that NTP plasma

not only affects the PM mass concentration, but also changes the PM size distribution. At

very high voltage levels (here 19.44 kVpp), NTP was very effective for PM mass reduction.

However, PM mass reduction is not the only concern. It became clear that at high voltage

levels the number of ultra-fine particles increases significantly. Regarding the negative health

effects of tiny particles, the performance of plasma at such a high voltage levels is not

desirable. Considering the PM mass reduction and PM size distribution simultaneously, an

optimum voltage level of 17 kVpp at 10 kHz was found for the given configuration and

operating condition. Moreover the wall attachments of particulates are another important

parameter which should be considered in all experiments. It was found that wall attachments

are variable even without introducing any plasma. Therefore, particle deposition insides the

reactor and its effect on plasma PM removal should be considered with more details in future.

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http://www.dieselnet.com/tech/dpm.html

195













Analysing DBD Plasma Lamp Intensity versus Power Consumption Using a Push-Pull Pulsed Power

Supply

Submitted to: 15th

European Conference on Power Electronics and Applications, France, Sep 2013.


Pooya Davari Proposed the initial idea. Designed and conducted simulation and data analysis.



22 Feb 2013



Arindam Ghosh Aided data analysis and writing the manuscript



Prof. Arindam Ghosh


196

Chapter

6 . A n a l y s i n g D B D P l a s m a L a m p I n t e n s i t y

v e r s u s P o w e r C o n s u m p t i o n U s i n g a P u s h -

P u l l P u l s e d P o w e r S u p p l y



† Danfoss Power Electronics, Graasten DK-6300, Denmark

Accepted in: 15th

European Conference on Power Electronics and Applications, France, Sep

2013.

6

Chapter 6

197

6.1. Acknowledgement

We thank Dr. Robert Carman of the Department of Physics and Astronomy at Macquarie

University, Sydney, Australia, for the helpful advice, and loan of the XeCl lamp which was

originally supplied by Prof. G. Zissis, LAPLACE Laboratory, Universite Paul Sabatier,

Toulouse, France.

Abstract- In this paper characteristic of a DBD (Dielectric Barrier Discharge) plasma lamp is

investigated based on the lamp intensity and power consumption. A pulsed power supply

with controllable parameters based on a push-pull converter is developed for lamp excitation

at different voltage levels and repetition rate. The experimentations were conducted for 28

different operating points with the frequency range of 2 kHz to 15 kHz at output voltage

levels of between 7.4 kVpp up to 13 kVpp. The obtained results show the feasibility of finding

an optimum operation point due to nonlinear behaviour of the DBD lamp.

6.2. Keywords

Pulsed power supply, Plasma, DBD Lamp, High voltage pulse.

6.3. Introduction

Dielectric barrier discharge (DBD) is a promising method in producing non-thermal

plasma, which is widely used in a variety of industrial applications. Recently, DBD lamps

have gained much attention as they are mercury free, easily scalable and simple to construct

[1-4]. DBD lamps are mostly filled with rare gases and rare gas-halides, which can provide

efficient scheme for generating incoherent UV (Ultra Violet) and VUV (Vacuum UV)

radiation [3]. The range of applications is broad owing to the variability of output wavelength

(88–310 nm) with the choice of gas fill.

DBD Lamps can be operated with continuous excitation or with pulsed excitation. Because

a pulsed discharge can operate at much higher peak voltages and peak currents for the same

average power as in a dc glow discharge, higher instantaneous sputtering, ionization and

excitation can be expected and hence better efficiencies [5, 6]. Recently, solid-state pulsed

power has gained more interest as it is compact, reliable, has a long lifetime and high

repetition rate. In the last decade, research and studies established the advantage of using

power electronics topologies in developing pulsed power supplies for variety of applications

Chapter 6

198

[6-10]. Therefore, a solid-state pulsed power supply can be a suitable choice for exciting a

DBD lamp as it provides controllable variables such as duty cycle, frequency, generated

voltage, and etc.

One of the important features of a DBD lamp is the light intensity. The light intensity is

quite important in sensitive applications such as in therapy of skin diseases and for

disinfecting the skin surface. Hence, increasing or decreasing generated UV intensity can

harm or not efficiently affect the under treatment area respectively. The light intensity can be

controlled by factors such as frequency (pulse repetition rate) and applied voltage. However,

another important issue which needs to take into account is the lamp power consumption [2].

Increment in power consumption not only reduces the system efficiency but also can lead to

short operational lifetime of the lamp. Thus, finding the relation between the lamp intensity

and power consumption in order to select a proper operating point is vital.

In this paper light intensity versus power consumption of a Xenon-filled coaxial DBD lamp

is analysed at different voltage level and repetition rates. The DBD lamp is excited using a

push-pull based pulsed power supply with the frequency range of 2 kHz to15 kHz and at

output voltage starting from 7.4kVpp up to 13kVpp. The light intensity is measured using a

UV detector and the power consumption is measured based on Lissajous V-Q diagram [2, 3]

at 28 different operating points. The obtained results not only depicted the performance of the

implemented pulsed power supply in driving the DBD lamp, but also the DBD lamp

illustrated a non-linear characteristic which can be advantageous in selecting an optimum

operating point.

6.4. Experimental Setup

6.4.1. DBD Lamp

Here a DBD lamp, as shown in Fig 6.1, is selected which is due to their particular interest as

they possess high efficiency [3]. The lamp tube has a coaxial geometry with a double

dielectric barrier, and is fabricated from UV grade fused silica tubing. The external electrodes

are wire/wire mesh yielding an active region of ~60mm length with a discharge gap of ~9mm

between the inner and outer dielectrics. The gas fill (sealed off) is a Xenon buffer at ~100mb

pressure, seeded with a low partial pressure (~1torr) of halogen (Cl) to yield XeCl excimers

upon discharge excitation which emit in the UV at 308m. Typical output irradiances from the

lamp are 1-10mW per cm2.

Chapter 6

199

6.4.2. Pulsed Power Supply

Fig 6.2 depicted a circuit schematic diagram of the developed pulsed power supply. As

illustrated, it is based on a push-pull inverter topology. The push-pull inverter contains two

switches that are driven with respect to ground. This is the main advantage of the inverter.

This topology uses a centre-tapped transformer which is excited in both directions. A step up

transformer is used to boost the voltage and achieve galvanic isolation.

Fig. 6.1 Dielectric barrier discharge lamp

In order to sustained NTP and prevent from arcing, bipolar pulse generation is employed

for clearing charges [11].The two switches S1 and S2 are switched alternately with a

controlled duty ratio to convert input DC voltage into high frequency AC voltage suitable for

exciting the DBD load. Hence, the generated output voltage is bipolar.

Adding a DBD load turns the push-pull inverter into a resonant stage with an

approximately sinusoidal output. The frequency of the semi-sinusoidal shape signal is

determined by an L-C circuit comprising of the transformer inductance and capacitances of

DBD and the transformer. The repetition rate can be used to adjust the power and by

optimizing the resonance it is possible to obtain high frequency semi-sinusoidal waveform.

The pulsed power supply was developed to provide complete control over output voltage

and repetition rate by means of regulating the input voltage and the duty cycle of power

switches. Fig. 6.3 shows the experimental hardware setup for the pulsed power supply. Here

1200V IGBT modules, SK75GB123, are used as power switches. Semikron Skyper 32-pro

gate drive modules are utilized to drive the IGBTs and provide the necessary isolation

between the switching-signal ground and the power ground. A Texas Instrument

TMS320F28335 DSC (Digital Signal Controller) is used for PWM signal generation. A

Chapter 6

200

centre-tapped step-up transformer with an UU100 core 3C90 grade material ferrite from

Ferroxcube, are designed with NA = NB = 5 and NS = 293. Here a 470 pF capacitor (CS) is

placed across each switch to protect them against the voltage spikes. To calculate the power

consumption a 4nF capacitor (Cm) is connected in series with the DBD lamp. Cm is selected

large enough to not to affect the DBD lamp capacitance. The output voltage is measured and

captured using a Pintek DP-22Kpro differential probe and RIGOL DS1204B oscilloscope,

respectively.

S2

S1

Vin DB

D L

am

p

NA

NB

NS

CS

CS

NA = NB << NS

Cm



An example of a measured output voltage of the employed pulsed power supply at the output

voltage of 13kVpp is depicted in Fig. 6.4. The depicted bipolar voltage waveform is generated

by switching S1 and S2 alternately (see Fig. 6.4a). This part (the first portion) of the output

voltage waveform is the resonant circuit dominated by the magnetizing inductance of the

transformer and the capacitances of the transformer and DBD. The period of this signal is

approximately 11.5 µs. The second part is the resonance happening during the switches off-

state between the leakage inductance and the capacitances of the transformer and DBD.

Chapter 6

201

(a)

(b)

Fig. 6.4 Output voltage of the employed pulsed power supply: a) at 10 kHz, b) typical measured output voltages

at 10 kHz and 5 kHz.

As the depicted signal (Fig. 6.4a) was generated at 10 kHz (repetition rate), therefore the

period of the second portion is equal to 88.5µs. It is to be noted that for all the applied

voltages and repetition rates the bipolar pulses were generated with a fixed duration of 11.5

µs (see Fig. 6.4b). This implies the importance of repetition rate which differs in four types of

applied pulses and can results in delivering varied power level to the DBD lamp.

6.4.3. Measurements

The lamp intensity was measured using a UV-photodetector (see Fig 6.3). The photo-detector

is equipped with JIC157 which has a spectral range of 210...390 nm. The averaged intensity

was considered for evaluation regarding to the area of the captured intensity waveform and

employed frequency.

Chapter 6

202

Fig. 6.5 V-Q cyclogram of the DBD lamp as a basis of power consumption calculation.

To measure the power consumption of the DBD lamp, the energy transferred to the DBD

lamp has been calculated by applying the Lissajous (V −Q) diagram [2, 3]. To measure Q one

capacitor (Cm) is placed in series with the DBD lamp, as depicted in Fig 6.2. Thus, by

measuring the voltage across Cm and multiplying it by Cm value it is possible to calculate Q.

The energy consumed by the plasma for one cycle is calculated from the area of V−Q curve

for different experiments (see Fig 6.5). Hence, by considering the employed repetition rate

(frequency) it is possible to calculate the average consumed power by the DBD lamp. Due to

presence of noise during measurement the captured data is filtered before power consumption

estimation stage (see Fig 6.5). The relevant equations are:

∮ (6-1)

(6-2)

By substituting (6-2) in (6-1):

∮

(6-3)

Therefore, considering (6-1) to (6-3) the averaged consumed power can be calculated as

below:

∮

(6-4)

Chapter 6

203

6.5. Results and Discussion

To analysis the lamp characteristics the lamp intensity versus the lamp power consumption

is considered at different voltage level and operating frequency. Input voltage (Vin) and

repetition rate (fr) were assigned as controlling parameters to determine the plasma lamp

characteristics. The high voltage pulses were applied to the plasma lamp at seven different

voltage levels and four different repetition rates of 2 kHz, 5 kHz, 10 kHz, and 15 kHz (see

Table 6.1). The 28 operating points are selected based on combination of varied operating

frequencies and voltages to evaluate the behaviour of the DBD lamp at variety of power

levels and intensities. The starting point was selected as Vin = 36 V as the first sustainable

plasma was achieved at this point. The measured results are illustrated as averaged intensity

versus averaged power consumption (see Fig. 6.6). It is to be noted that the results are

normalized for a better comparison.

Fig. 6.6a depicted the applied voltage versus power consumption. The illustrated data

clearly indicate that the DBD lamp power consumption correspondingly increases with the

applied voltage level and the repetition rate. This can be also realized from (6-4), which

shows the relation between the power, repetition rate and the applied voltage. However, as

the power is proportional to the applied voltage squared, the applied voltage is much more

effective than the repetition rate on the power level.

Table 6.1 Considered voltages and frequencies for different experiments

Frequency

Output Voltage Level

2kHz

5kHz

10kHz

15kHz

Vin= 36V D11 D21 D31 D41

Vin= 40V D12 D22 D32 D42

Vin= 44V D13 D23 D33 D43

Vin= 48V D14 D24 D34 D44

Vin= 52V D15 D25 D35 D45

Vin= 56V D16 D26 D36 D46

Vin= 60V D17 D27 D37 D47

Chapter 6

204

(a)

(b)

Fig. 6.6 Comparing different operation points regarding to the varied applied voltage levels and repetition rates,

a) applied voltage versus plasma lamp power consumption, b) applied voltage versus plasma lamp intensity.

Fig. 6.7 The DBD lamp light intensity measured for varied applied voltage levels.

Despite the power consumption which is always desirable to be as low as possible, it is the

lamp intensity which is the main goal. Obtaining high intensity at low power consumption in

plasma lamp applications is one of the major concerns. The obtained results regarding to the

applied voltage versus the lamp intensity are illustrated in Fig. 6.6b. It is obvious that

Chapter 6

205

increasing the applied voltage results higher intensity (see Figs 6.6b and 6.7). However, the

obtained results indicate that not necessarily the higher intensity the high power consumption

is. In the other word, the intensity doesn’t have a linear relation with the power consumption.

The measured results in Fig. 6.6 depicted the possibility of achieving same or even higher

intensity at lower power consumption by increasing the repetition rate. For instance,

comparing D44 with D36 shows that, D44 has higher intensity than D36 while it has same

power consumption. Such behaviour can be realized by comparing D32 with D27 as well.

This can be due to the different slope of change regarding to the applied voltage and

consumed power.

Regarding to the aforementioned facts, it is feasible to find an optimum operation point

where the required intensity can be achieved at lower power consumption. This can be

obtained by operating at lower voltage levels but with the cost of higher repetition rate.

Reducing the generated voltage level while obtaining the required performance is always

desirable due to high voltage insulation and power switches limited breakdown voltage

issues.

6.6. Conclusion

In this paper, the behaviour of a DBD plasma lamp was analysed with respect to its

intensity and power consumption. A pulsed power supply based on the push-pull topology

was developed with controllable parameters to trigger the DBD lamp over a wide range of

operating conditions. The feasibility of optimum operation point due to non-linear

characteristics of the plasma lamp was concluded based on the results obtained. To ensure

that the following conclusions apply for different power levels of the DBD lamp, all

experiments have been conducted on 28 different operating points by combining different

repetition frequencies and applied voltages.

6.7. References






Chapter 6

206



[4] U. N. Pal, P. Gulati, N. Kumar, M. Kumar, M. S. Tyagi, B. L. Meena, et al., "Analysis of

Discharge Parameters in Xenon-Filled Coaxial DBD Tube," Plasma Science, IEEE





[6] L. Redondo and J. F. Silva, "Solid State Pulsed Power Electronics," in POWER

ELECTRONICS HANDBOOK DEVICES, CIRCUITS,ANDAPPLICATIONS, M. H.Rashid,

Ed., Third Edition ed: Elsevier, 2011.

[7] P. Davari, F. Zare, and A. Ghosh, "Parallel and series configurations of flyback converter for

pulsed power applications," in Industrial Electronics and Applications (ICIEA), 2012 7th

IEEE Conference on, 2012, pp. 1517-1522.

[8] P. Davari, F. Zare, and A. Ghosh, "Flexible Solid-State Pulsed Power Topology," presented at

the 15th International Power Electronics and Motion Control Conference, EPE-PEMC 2012

ECCE Europe, Novi Sad, Serbia, Sep 2012.






on, vol. 37, pp. 171-178, 2009.

[11] R. A. Scholl, "Power Supplies for Pulsed Plasma Technologies: State-Of-The-Art And

Outlook,"" Advanced Energy Industries, Inc, pp. 1-8, 1999.

207













Power Electronic Converters for High Power Ultrasound Transducers

Published in proceedings of: The 7th IEEE Conference on Industrial Electronics and Applications, ICIEA

2012, Singapore.


Negareh Ghasemi Proposed the initial idea and designed and implemented the pulsed power supply.

Planned and conducted experimental verification and wrote the manuscript

Firuz Zare Proposed the initial idea and evaluation process. Supervised the data analysis,

experimentation and writing the manuscript

Pooya Davari Aided data analysis, experimental verification, and writing the manuscript

22 Feb 2013

Christian Langton Aided data analysis and writing the manuscript

Peter Weber Provided the piezoelectric transducer and general information about the ultrasound

applications and writing the manuscript

Arindam Ghosh Aided data analysis and writing the manuscript



Prof. Arindam Ghosh


208

Chapter

7 . P o w e r E l e c t r o n i c C o n v e r t e r s f o r H i g h

P o w e r U l t r a s o u n d T r a n s d u c e r s



Science and Engineering Faculty, Queensland University of Technology, GPO BOX 2434,

Brisbane, Australia

Ultrasound-System-Development, Fraunhofer Institute for Biomedical Engineering, Germany

Presented and Published at: The 7th


Applications, ICIEA 2012, Singapore, July 2012.

7

Chapter 7

209

Abstract- Piezoelectric transducers convert electrical energy to mechanical energy and play a

great role in ultrasound systems. Ultrasound power transducer performance is strongly related

to the applied electrical excitation. To have a suitable excitation for maximum energy

conversion, it is required to analyze the effects of input signal waveform, medium and input

signal distortion on the characteristics of a high power ultrasound system (including

ultrasound transducer). In this research, different input voltage signals are generated using a

single-phase power inverter and a linear power amplifier to excite a high power ultrasound

transducer in different mediums (water and oil) in order to study the characteristics of the

system. We have also considered and analyzed the effect of power converter output voltage

distortions on the performance of the high power ultrasound transducer using a passive filter.

7.1. Keywords

Component; Power Converter, Piezoelectric Transducer Excitation

7.2. Introduction

Much research has been conducted on piezoelectric behavior since 1880, the year that Pierre

and Jacque Curie discovered the phenomenon of piezoelectricity. The critical behavior of a

piezoelectric device is encapsulated in its resonant frequencies and the most efficient way to

find the critical piezoelectric specifications is to analyze its impedance frequency response [1].

IEEE Standard on Piezoelectricity introduced the basic equivalent circuit model characterizing

a piezoelectric ceramic near the resonant frequency which is known as Van Dyke Model. This

model is often adapted to model electromechanical resonance characteristics of crystal

oscillators.

The Van Dyke Model is a parallel connection of a series RLC representing mechanical

damping, mass, and elastic compliance and a capacitor representing the electrostatic

capacitance between the two parallel ceramic plates [2]. When a piezoelectric ceramic is

mounted to a mechanical structure, a loaded piezoelectric ceramic experiences multiple

resonances, a circuit model (Fig. 7.1 (a)) for a wide frequency range with multiple resonant

frequencies can be employed to model the behavior of a loaded piezoelectric ceramic.

Ultrasound systems are used in different industrial and medical applications. According to

various applications, an ultrasound system can be used in low (1-100 W) and high (0.1- 50

kW) power and frequency ranges. For instance, in biomedical applications a high frequency

Chapter 7

210

ultrasound system is used for diagnosis (low power) or therapeutic application (high power).

In order to generate ultrasound wave, piezoelectric transducer is a key part of the ultrasound

system which converts electrical to mechanical energy. A most important issue in exciting a

high power ultrasound transducer is quality and shape of the electrical signal which drives the

power transducer [3-9]. It is important to generate a high quality power signal at its resonant

frequency with low distortion to attain the highest energy conversion. Different methods are

introduced to generate a suitable signal to drive a power transducer such as radio-frequency

linear amplifiers and switched mode power converters [6].

A main advantage of switched mode power converters compare to power amplifiers is its

high efficiency at high power operation. Fig. 7.1 (b) and Fig. 7.1 (c) show a power converter

connected to a piezoelectric transducer as a load and the output voltage levels of a power

converter respectively. Multilevel converters are suitable power converters to drive power

transducers due to their attractive ability to generate a high quality power waveform with low

harmonic distortion and voltage stress [10-12].

In order to drive a power transducer with an appropriate signal, it is essential to study and

analyze the impedance of a piezoelectric transducer at different frequency and power ranges.

Usually, a network analyzer is used to measure piezoelectric transducer impedance and its

resonant frequencies in frequency domain. It is not possible to analyze and study the

characteristics of a high power transducer using a network analyzer due to the fact that a

network analyzer operates at low power[5]. Therefore, some tests have been carried out in this

research to study the performance and behavior (linear or nonlinear) of ultrasound system at

high power range which are presented in the next sections.

Chapter 7

211

(a)

(b)

(c)

Fig. 7.1 (a) Van Dyke Model, (b) a power converter, (c) output voltage of power converter.

7.3. Experimental Procedure

In this research work we have used a power ultrasound transducer which has some resonant

frequencies below 100 kHz and high energy conversion happens at those resonant frequencies.

Table 7.1 shows a summary of all test conditions which have been carried out at different

mediums and input voltages. In the first two tests, sinusoidal voltage waveforms are generated

by a linear power amplifier to drive the piezoelectric transducers. In these two tests, nonlinear

characteristics of the high power ultrasound system is analyzed using superposition method.

Chapter 7

212

In the other tests, a high power single phase inverter is used to generate high voltage and

high frequency signals (square wave) to analyze behavior of ultrasound system when it is

driven by non-sinusoidal signals.

In test 1 and test 2, a power amplifier (OPA 459) is connected to a step-up transformer to

generate a high voltage signal to excite a high power transducer as shown in Fig. 7.2. Two

transducers (same type) are placed in a container (exactly opposite to each other) and one

transducer is excited by the electrical signal as a transmitter and the other one converts the

generated ultrasound signal to electrical energy at the other side of the container as a receiver.

Since the performance of the piezoelectric transducer is highly dependent on its excitation,

generating a proper sinusoidal signal with low order harmonics is required. If the frequency of

the excitation signal is same as the resonant frequency of the transducer, the highest energy

conversion will be attained. A signal generator and a high power amplifier (OPA 459) are used

to generate a high power signal (up to 8 A at 30 Volts) but a high frequency step-up

transformer is used to increase the output voltage level. The piezoelectric has some resonant

frequencies around 39 kHz. We have generated two sinusoidal signals at 39 kHz and 61 kHz

in two different tests. In test 1, the magnitudes of the sinusoidal signals are adjusted at 15 V

and 30 V and we have excited the first transducer at 39 kHz and 61 kHz separately and have

measured the output voltage of the second transducer in time domain. In order to check the

quality of the input and output voltages at 15 V and 30 V, the signals are shown in frequency

domain (Fig. 7.3).

The test results show that the output voltage measured by the second transducer is not

proportional to the input voltage. This test verifies that the ultrasound system has nonlinear

behavior at its resonant frequencies. In order to study the performance of the ultrasound

system more, a superposition law is used as a key factor. Based on this principle, if the

ultrasound system has a linear characteristics, the responses of the system with two sinusoidal

signals (as follow) should be same:

a) Separately excited by two signals at 39 kHz and 61 kHz and adding the output signals

b) Simultaneously excited by two signals at 39 kHz and 61 kHz

Chapter 7

213

Table 7.1 Test conditions and setups

Test # Medium Excitation System Input Voltage

Magnitude (peak)

Frequency of Input

Signal (s)

Test 1 water

Signal Generator, a

power amplifier and a

high frequency

transformer

15 V and 30 V 39 kHz and 61 kHz

Test 2 water

Signal Generator, a

power amplifier and a

high frequency

transformer

15 V + 30 V

(time domain) 39 kHz and 61 kHz

Test 3 oil three-level inverter

50 V, 100 V, 200 V

and

300 V

39 kHz

Test 4 oil

three-level inverter and

a tube between two

transducers

50 V, 100 V, 200 V

and

300 V

39 kHz

Test 5 oil three-level inverter and

a filter

50 V, 100 V, 200 V

and

300 V

39 kHz

Test 6 water three-level inverter

50 V, 100 V, 200 V

and

300 V

39 kHz

Fig. 7.2 A block diagram of a lab prototype for test 1 and test 2.

Chapter 7

214

(a)

(b)

(c)

(d)

Fig. 7.3 (a) input signals at 39 kHz (b) output signals at 39 kHz (c) input signals at 61 kHz (d) output signals at

61 kHz

Therefore, in test 2, the same transducer is excited by two signals at 39 kHz and 61 kHz

which are added together at the input side of the power amplifier. We expect the output

voltages of the ultrasound system in two different tests should be exactly same as each other if

the system has a linear characteristics. The test result for each voltage level is shown in Fig.

7.4 (a). In order to compare this test results with the previous one, we have added the output

voltage results of test 1 at 39 kHz and 61 kHz in time domain and the results are shown in Fig.

7.4(b). The output voltage of each test at 15 V and 30 V are shown in Fig. 7.4 (c) and Fig. 7.4

(d) and it is clear that the output voltages (separately and simultaneously excited) are not

same. A difference between these test results shows the ultrasound system has a nonlinear

characteristics when the input voltage and power are increased.

Chapter 7

215

(a)

(b)

(c)

(d)

Fig. 7.4 (a) Test 1: summation of two output signals (at 39 kHz & 61 kHz) for Vin=15V and Vin=30 V

(b) Test 2: two output signals for Vin=15V and Vin=30 V. (c) Comparing the results of test 1 and test 2 for

Vin=15 V. (d) Comparing the results of test 1 and test 2 for Vin=30 V.

The results of Fig. 7.4 show that the ultrasound system does not obey the superposition

principle and it has nonlinear behavior as the output voltages of the two tests at 30 V are not

same. In order to study the nonlinearity of the ultrasound system at higher power and different

mediums, different tests have been carried out using a single phase inverter, generating a

square wave uni-polar voltage waveform. A laboratory setup of this configuration is shown in

Fig. 7.5.

A coupling box of laboratory setup is filled of oil or water for different tests in this research.

A block diagram of the setup of test 3 is shown in Fig. 7.6. In this test, a power converter

generates a square wave signal (uni-polar modulation) at different voltage levels (50V, 100V,

200V and 300V) and at 39 kHz including some harmonics. In order to compare the quality of

all input voltages, we have normalized all input voltage at 50 Volts (dividing all input voltages

by factors of 1, 2, 4 and 6, respectively) and the input voltage waveforms are shown in

frequency domain (Fig. 7.7 (a)). Then, we have measured the output voltage of the second

transducer at different excitation voltages and the results are shown in Fig. 7.7 (b). Similar to

Chapter 7

216

the input voltages, we have divided the output voltages by the same factors (1, 2, 4 and 6,

respectively) in order to compare the output voltages. The results show that the ultrasound

system has nonlinear characteristics at different voltage levels and the output voltages are not

same when they are normalized.

Fig. 7.5 The experimental setup.

Fig. 7.6 A block diagram of test 3.

Chapter 7

217

(a)

(b)

Fig. 7.7 The results of test 3 (a) input signals and (b) output signals.

According to the applied input voltages, an ultrasound wave is generated and transferred to

the second transducer. But some of the generated wave will be attenuated due to interaction

with inhomogeneous material. The velocity and attenuation of generated wave are highly

dependent on the mechanical and structural properties of the medium [13, 14]. To study the

effect of attenuation of generated ultrasound wave, a tube is placed between the two

transducers in which ultrasound wave is guided from the first transducer to the second

transducer. It is expected that the tube reduces the propagation of generated wave and thus

increases the intensity of the wave at the second transducer. A block diagram of this

configuration is illustrated in Fig. 7.8.

Similar to the previous test, we have normalized all input and output voltages in order to

check the quality of the input voltages and compare the output voltages in frequency domain.

As is shown in Fig. 7.9 the output voltage magnitudes are increased compare to the previous

Chapter 7

218

test result (Fig. 7.7 (b)) but there are still differences in the output voltage magnitudes due to

the nonlinear behaviour of the system at those frequencies.

The output voltage of the power inverter generates voltage stress (dv/dt) across the

piezoelectric transducer. This voltage with the capacitive characteristics of the piezoelectric

transducer can generate significant current spikes which increases losses and high frequency

noise. Since the input voltage distortion can tend to deteriorate the output voltage quality and

can increase the power dissipation, a 990 μH inductor as a filter is placed between the power

converter and the transducer to reduce the input voltage distortion across the first transducer.

Fig. 7.10 shows a block diagram of the setup.

The added filter reduces the amplitude of the output voltage in frequency domain (Fig. 7.11)

compared to the test results shown in Fig. 7.7 but this filter has not a significant effect on the

characteristics of the ultrasound system and it has still a nonlinear characteristics at those

frequencies which is shown in Fig. 7.11 (b).

In order to study the effects of medium on the ultrasound system characteristics, we have

performed a new test similar to test 3 but with water (instead of oil). A block diagram of the

setup is shown in Fig. 7.12.


Chapter 7

219

(a)

(b)



Similar to the previous test results, when the input and output voltages are normalized at 50

V, there are still significant differences between the output voltages at different frequencies

shown in Fig. 7.13. The nonlinear behavior of a high power ultrasound system is obvious in

this figure due to mismatch of the responses of the ultrasound system to the normalized input

voltages. The nonlinearity of the ultrasound system has been shown in different cases where a

high power ultrasound transducer was excited by a sinusoidal and a pulse voltage waveform.

Chapter 7

220

(a)

(b)



Chapter 7

221

(a)

(b)


7.4. Conclusion

In this research a high power transducer is driven by different input signals and the

performance of ultrasound system is studied and analyzed in several frequencies. According to

the first and the second test results, it is obvious that the ultrasound system has nonlinear

behavior at high voltage and high power. In order to analyze the effects of a) input signal

waveform, b) medium and c) input signal distortion on characteristics of an ultrasound system,

several tests with different setups are carried out. According to the test results of this study and

research work, it is concluded that a high power ultrasound system has nonlinear

characteristics with respect to the input voltage magnitude. The results verify that the

nonlinear characteristics of a high power ultrasound system exists in different medium. It

means that a high power ultrasound system may be saturated when the input voltage

magnitude is increased and the piezoelectric ceramic plates cannot vibrate proportional to the

input voltage.

Chapter 7

222

7.5. References

[1] H. Walter, L. Karl, W. Wolfram, Piezoelectricity. New York: Springer, 2008.

[2] N. E. R. Centre, "Charactrestic of Piezoeletric Transducer," 2001-2011.

[3] S. Ozeri, D. Shmilovitz, "High frequency resonant inverter for excitation of piezoelectric

devices," in IEEE Power Electronics Specialists Conf., Jun 2008.

[4] S. B. Yaakov, N. Krihely "Modeling and driving piezoelectric resonant blade elements," in in

Nineteenth Annual IEEEApplied Power Electronics Conf. , 2004, pp. 1733-1739.

[5] A. A. Vives. (2008). Piezoelectric Transducers and Applications (2 ed.). Available:

http://QUT.eblib.com.au/patron/FullRecord.aspx?p=372502

[6] T. S. Chu, G. T. Clement, "A Harmonic Cancellation Technique for an Ultrasound

Transducer Excited by a Switched-Mode Power Converter," IEEE TRANS. ON

ULTRASONICS, FEROELECTRICS, AND FREQUENCY CONTROL, vol. 55, Feb 2008.

[7] T. Suzuki, H. Ikeda, H. Yoshida, S. Shinohara, "Megasonic transducer drive utilizing

MOSFET DC-to-RF inverter with output power of 600 W at 1 MHz," IEEE Trans. on

Industrial Electronics, vol. 46, pp. 1159-1173, 1999.

[8] K. Agbossou, J. L. Dion, S. Carignan, M. Abdelkrim, A. Cheriti, "Class D amplifier for a

power piezoelectric load," IEEE Trans. on Ultrasonics, Ferroelectrics and Frequency

Control, vol. 47, pp. 1036-1041, 2000.

[9] P. Fabijanski, R. Lagoda, "Series resonant converter with sandwich-type piezoelectric

ceramic transducers," in Proc. IEEE International Industrial TechnologyConf., 1996, pp.

252-256.

[10] A. Nami, F. Zare, A. Ghosh, "Asymmetrical DC link voltage configuration for a diode-

clamped inverter," IEEJ Transactions on Industry Applications (Denki Gakkai Ronbunshi. D,

Sangyo Oyo Bumonshi), vol. 130, pp. 195-206, 2010.

[11] A. Nami, F. Zare, A. Ghosh, F. Blaabjerg, "A Hybrid Cascade Converter Topology With

Series-Connected Symmetrical and Asymmetrical Diode-Clamped H-Bridge Cells," IEEE

Transactions on Power Electronics, vol. 26, pp. 51-65, 2011.

[12] A. A. Boora, A. Nami, F. Zare, A. Ghosh, F. Blaabjerg, "Voltage-Sharing Converter to

Supply Single-Phase Asymmetrical Four-Level Diode-Clamped Inverter With High Power

Factor Loads," IEEE Transactions on Power Electronics, , vol. 25, pp. 2507-2520, October

2010.

[13] M. Willatzen, "Ultrasound transducer modeling-received voltage signals and the use of half-

wavelength window layers with acoustic coupling layers," IEEE Trans. on Ultrasonics,

Ferroelectrics and Frequency Control, vol. 46, pp. 1164-1174, 1999.

[14] A. Lanata, E. P. Scilingo, R. Francesconi, D. De Rossi, , "Performance Analysis and Early

Validation of a Bi-modal Ultrasound Transducer," in in 28th Annual International IEEE

Medicine and Biology Society Engineering Conf., 2006, pp. 1858-1861.

http://qut.eblib.com.au/patron/FullRecord.aspx?p=372502

223













Improving the Efficiency of High Power Piezoelectric Transducers for Industrial Applications

Published in: IET Science, Measurement, and Technology, vol.6, no.4, pp.213-221, July 2012.


Pooya Davari Proposed the initial idea. Designed and conducted simulation and data analysis.



22 Feb 2013

Negareh Ghasemi Aided data analysis, experimental verifications and writing the manuscript



Peter O’Shea Aided data analysis, designing the algorithm and writing the manuscript

Arindam Ghosh Proposed the initial idea. Designed and conducted simulation and data analysis.





Prof. Arindam Ghosh


224

Chapter

8 . I m p r o v i n g t h e E f f i c i e n c y o f H i g h P o w e r

P i e z o e l e c t r i c T r a n s d u c e r s f o r I n d u s t r i a l




Published in: IET Science, Measurement, and Technology, vol.6, no.4, pp.213-221, July

2012.

8

Chapter 8

225

Abstract: Most high power ultrasound applications are driven by 2-level inverters. However,

the broad spectral content of the 2-level pulse results in undesired harmonics which can

decrease the performance of the system significantly. On the other hand, it is crucial to excite

the piezoelectric devices at their main resonant frequency in order to have maximum energy

conversion. Therefore a high quality, low distorted power signal is needed to excite the high

power piezoelectric transducer at its resonant frequency. This paper proposes an efficient

approach to develop the performance of high power ultrasonic applications using multilevel

inverters along with a frequency estimation algorithm. In this method, the resonant frequencies

are estimated based on relative minimums of the piezoelectric impedance frequency response.

The algorithm follows the resonant frequency variation and adapts the multilevel inverter

reference frequency to drive an ultrasound transducer at high power. Extensive simulation and

experimental results indicate the effectiveness of the proposed approach.

8.1. Introduction

Wide spread research has been conducted on piezoelectric transducer applications since

1880, the year that Pierre and Jacque Curie discovered the phenomenon of piezoelectricity [1,

2]. Piezoelectric transducers convert electric power to acoustic power and vice versa, with

most of the applications to date being low power ones. In the last decade, however, high

power ultrasound applications have gained significant importance [3-8]. These types of

applications have great potential in chemical and bio-technology processing, specifically for

enhancing chemical reaction kinetics and new reaction pathways. Such enhancements allow

changing the production from batch processing to continuous flow processing, thereby

reducing investment and operational costs.

Improvement of ultrasound systems has very significant environmental implications,

particularly in the area of renewable energy (bio-mass and bio-fuel), waste-water treatment

and biomedical applications. High power ultrasound technology is already being used to

address these areas of need, but to a very limited extent due to the very inefficient nature of the

state of the art in energy conversion.

The critical behavior of a piezoelectric device is encapsulated in its resonance frequencies

due to its maximum transmission performance at these frequencies [1, 2, 9-11]. Hence, ideal

scenario is to have a sinusoidal excitation signal at the resonant frequency. But at high power

and high voltage, which is the focus area of this paper, generating pure sinusoidal signals is

not possible.

Chapter 8

226

The most efficient way to generate power signals is to use a 2-level power inverter.

Specifically, switch mode inverters are used for piezoelectric high power applications due to

their high power density, efficiency, low cost and size compared to conventional linear power

supplies [5, 12, 13]. However, the harmonics present in the output waveform produce

undesired side bands which are not suitable in many applications. Moreover, they also cause

unnecessary power dissipation which reduces the efficiency of the power converter [8, 14].

On the other hand, for ultrasound applications which operate at high fundamental frequency

power converters based on a PWM strategy with high switching frequency index (fs/fo) cannot

be a practical solution. In such applications, the maximum possible fundamental frequency of

a converter is restricted by the switching transients of each power switching event and the

number of switching events per cycle.








Fig. 8.1 depicts typical voltage waveforms and their harmonic spectra. It can be seen that the

harmonics are depressed for multilevel inverters such that they can be easily filtered out from

the frequency range of interest.

To improve the performance of the piezoelectric transducer for high power applications, in

addition to the multilevel converter, the device needs to be excited at its resonant frequency.

Piezoelectric devices typically have multiple resonant frequencies, but only the major resonant

frequency is generally targeted for excitation in practice. Structural and environmental

changes of a piezoelectric system can affect variations in the resonant frequencies [2, 13].

Therefore, it is important to estimate the main resonant frequency in order to maintain

efficient system operation. Therefore, the multilevel converter needs to be adapted with a

suitable frequency estimation algorithm.

Chapter 8

227

Fig. 8.1 Effect of using multi-level waveform in harmonic elimination.

The most effective way to find the resonant frequencies of a piezoelectric transducer is by

evaluating its impedance frequency response [19-28]. A minimum in the impedance response

corresponds to a resonant frequency, fr. The impedance frequency response is the ratio of the

voltage spectrum to the current spectrum. To calculate the piezoelectric impedance response, a

voltage source needs to be applied to the device as an excitation signal and current needs to be

measured simultaneously. In order to obtain the response of the device for a specific range of

frequencies, the excitation signal should cover the entire frequency range. The idea of

calculating the piezoelectric impedance is inspired by the general concept of performing

system identification (i.e. finding the system transfer function [29]). It is therefore possible to

benefit from the existing knowledge base of excitation signals for system identification.

Considering the above mentioned features, a broad-band excitation signal is the most

appropriate candidate [29, 30].

Chapter 8

228

DC/AC

Inverter

AC

Current Sensor

Piezoelectric

Frequency

Estimation

Fig. 8.2 Exploiting online resonance frequency estimation in high power applications of piezoelectric devices.

This paper advocates the advantage of using multilevel inverters along with an efficient

frequency estimation algorithm (as depicted in Fig. 8.2) to develop the performance of the

high power ultrasonic applications. In the proposed algorithm, a 1 kHz rectangular pulse with

10% duty cycle is applied to the device and the current is measured as the response. Then the

captured data is cropped to retain only one cycle of the applied input voltage and

corresponding output current. In the next step, FFT (Fast Fourier Transform) is applied to both

the signals and the impedance response is found as a ratio of the voltage to current transforms.

Finally, the relative minimum values are estimated and sorted according to both impedance

derivative and impedance magnitude. The FFT is used because it can be computed relatively

efficiently (with order NlogN operations), thus enabling real-time operations. Finally, the

multilevel inverter reference frequency is updated with the estimated resonance frequency.

The proposed method has been evaluated in simulations (using an electrical circuit model) and

experimentally (using two piezoelectric devices). The results obtained indicate the efficiency

and high performance of the proposed method.

8.2. Methodology and Approach to Estimate Resonance

Frequency

8.2.1. Excitation Signal

Piezoelectric devices have different resonance frequencies which are sometimes described as

vibration modes. To identify these frequencies, the excitation signal needs to be wide-band. In

the proposed approach, a sequence of rectangular pulses is applied to the device. The pulse

widths are 0.1ms and the pulses are reapplied every 1ms. It is quite easy to generate the

Chapter 8

229

selected pulse stream and the pulses occur fast enough to enable the system to create regular

updates of the resonance frequency online.

Fig. 8.3 (a) shows the frequency response of a typical rectangular pulse. As can be seen the

energy in the spectrum is well spread, but varies considerably in intensity as a function of

frequency. The frequency response of the output current will be moderated by this input

spectrum since it is not possible to study the piezoelectric characteristic by just looking at the

response to the rectangular pulse. One needs to compute the impedance response by forming

the quotient of the input and output responses.

Fig. 8.3(b) and (c) show a sample of the captured voltage and current after applying the

proposed excitation signal. As can be seen, an imperfect pulse is obtained in a practical

situation. This imperfection actually proves to an advantage because the practical spectrum

does not suffer from the problem of having zero energy at some frequency components (see

Fig. 8.3 (d)).

8.2.2. Estimating Impedance

The response due to the excitation signal (also known as the ‘residual vibrations’) is given

by:

n

k

kkk tatfSinAty1

)exp()2()( (8-1)

where, n is the number of resonance frequencies, Ak is the amplitude, fk is the frequency and

ak is the damping coefficient of the thk resonance frequency.

A current sensor is used to capture the response (residual vibrations) of the device. The

current and voltage across the piezoelectric are captured simultaneously. To get the best

results, one cycle of the excitation pulse, along with the corresponding current response, needs

to be extracted from the captured signals. The starting point of the voltage waveform is

specified on the leading edge of the pulse. Then, from a knowledge of the sampling rate and

the length of the signal (which is 1ms) the end point is determined. Based on these starting and

end points the voltage and current waveforms are cropped from the captured signals (see Fig.

8.3 (b) and (c) for sample captured signals).

Chapter 8

230

(a)

(b)

(c)

(d)

Fig. 8.3 a) Ideal pulse frequency response, b) Captured voltage c) Captured response (current), d) Practical pulse

frequency response.

Chapter 8

231

After cropping, the FFT of the both signals are calculated as follows:

1,...,1,0,)()(

1

1

2

NkenxkX

N

n

N

nki

(8-2)

where x(n) and X(k) are the discrete inputs and outputs respectively. To find the power

spectrum of the voltage and current signals the FFT outputs are multiplied by their conjugates

as per equation (8-3) and finally the impedance is calculated based on equation (8-4).

N

XXPx

.

(8-3)

NN

FffP

fPfP s

i

vz /)

2:0(,

)(

)()( (8-4)

where Fs denotes the sampling frequency and N is the number of FFT points.

8.2.3. Extracting Resonant Frequencies

The resonant frequencies correspond to the local (or relative) minimums of the piezoelectric

impedance. The relative minimums of a function are the points where that the slope of the

tangent changes from – to +. Fig. 8. 4(a) shows a power spectrum, )( fPzof an impedance and

the change of slope in the resonant frequency. At point A, )( fPz is equal to zero. Moreover,

immediately to the left of this point at point B the slope is negative while at point C the slope

is positive.

Motivated by the above, the derivative of )( fPz is calculated and all the points where )( fPz

changes sign from – to + are extracted. As the final step, at the extracted frequencies, a sorting

is performed based on the magnitude of )( fPz. The main resonant frequency is the one with the

lowest magnitude.

The procedure described above is depicted as an algorithm flowchart in Fig. 8.4(b). Within

that flowchart R is a constant defining the number of resonant frequencies needing to be

extracted. The algorithm can be used both for offline and online systems. It is important to

note that, repeating the proposed algorithm, while applying a repetitive pulse and averaging,

results in an increase of the estimation accuracy.

Chapter 8

232

(a)

Input

captured

signals

Finding

starting and

end points

Cropping the

signals

Saving as

relative

minimum

Sorting using

amplitude

Yes

No

Finding the

minimum

amplitude

Number of extracted

frequencies = R

No

Yes

(b)

Fig. 8.4 Extracting resonant frequencies: a) change of slope at relative minimum (resonant frequency), b)

flowchart of the proposed algorithm.

Chapter 8

233

8.2.4. Noise Issues

In practical systems the presence of noise is inevitable. Fortunately, at resonant frequencies,

the impedance of the device is minimum and current increases significantly. The signal-to-

noise ratio is therefore relatively high. Noise is nonetheless an issue, and noise removal is

therefore required.

Several approaches have been introduced for noise reduction. One of the most common

ways to reduce the noise is to use an anti-aliasing filter [31]. This type of filter is used before

sampling to limit the bandwidth of a signal. A simple anti-aliasing filtering is proposed for use

on the analog signal before sampling. This is in keeping with the goal to devise a simple

method which is easy to implement and fast enough to follow the resonant frequency

variations. In addition to a low-pass filter, it is also important to choose the sampling

frequency in accordance with the Nyquist theorem [30] and to ensure that there is a good

coupling at sampling points.

It is recommended that a simple RC low-pass filter be used between the current sensor and

the capturing device. The cut-off frequency of the filter is selected according to:

ff

cCR

f2

1 (8-5)

8.3. Simulation and Experiment

In this section, in order to show the performance of the proposed method in frequency

estimation, the proposed algorithm is evaluated via both simulations and experimentation. The

effect of using a multilevel inverter is shown in the last part as a practical evaluation using

ultrasound interface. The proposed frequency estimation algorithm is compared with the three

key types of alternative impedance analysing methods. These alternative methods are:

a) The traditional method: In this method several single frequency sine waves from 30 kHz

to 80 kHz have been applied separately to the piezoelectric. The frequency step size was

set to 1 kHz.

b) The Network Analyser method: An R&S ZVL3 vector network analyser has been used

to obtain the impedance frequency response of a piezoelectric device in the frequency

range of 30 kHz to 80 kHz.

Chapter 8

234

c) The unit step and white noise excitation method: In order to compare the proposed

method with different wide-band excitation signals, the impedance frequency behaviour

has been obtained based on step response and white noise.

Step Pulse: From system identification theory, it is known that the Fourier transform

of the impulse response (h(t)) of a system gives the system transfer function. Since

generating an impulse ( )(t ) at high power is highly impractical a step (u(t)) is

preferable. Here a 1ms step is applied to the device and as discussed in the previous

section the impedance response can be obtained by dividing the Fourier transform of

the output into the Fourier transform of the input.

White Noise: One of the most popular excitation signals used in system identification

is white noise [29, 30]. As white noise theoretically has a constant amount of energy

per frequency band, it is possible to simply look at the captured current to find the

resonant frequencies. To account for the fact that white noise spectra are not always

perfectly flat in practice, the impedance response is calculated in the same way as the

other two methods – by dividing the Fourier transform of the output into Fourier

transform of the input.

The software that was used for simulation and analysis of experimental results was Matlab

7.10. The sampling frequency in both simulation and experimental results was 2 MHz and the

number of FFT points, N, was set to 1024. Here two different kinds of piezoelectric devices

were used (Type A and Type B). Type A has three dominant resonant frequencies while Type

B has just one. For the simulation and experimental evaluations the piezoelectric devices were

immersed in housing containing water.

8.3.1. Simulation Results

In order to simulate and compare the proposed method with other methods, a circuit model

of the piezoelectric device was needed. The electrical circuit model [26] of a piezoelectric

transducer is shown in Fig. 8.5(a). The parameters in the electrical circuit model were

determined for the experimental piezoelectric device by performing measurements on a

network analyser. These values are presented in Table 8.1.

Chapter 8

235

Rp

Rs

R1

C0

C1

L1

R2

C2

L2

R3

C3

L3

(a)

(b)

Fig. 8.5 Simulation results for piezoelectric Type A: a) circuit model, b) impedance response.

The accuracy of the circuit model was further evaluated by comparing simulated and

measured impedance responses from a network analyser. Fig. 8.5(b) shows the impedance

responses of the piezoelectric device obtained from real measurements and from simulation.

As can be seen from the figure only three resonant frequencies have been modelled. With the

model established the next step was to compare the proposed method with the alternatives via

simulations. To this end the resonant frequencies for all methods were calculated based on the

model in Fig. 8.5(a) and are presented in Table 8.2.

Chapter 8

236

Table ‎8.1 Values of the Components in the Electrical Circuit Model

Non-resonant part First resonant mode (70.19 kHz)

C0(F) Rs )( Rp )( R1 )( L1(H) C1(F)

9103 10 5101 492.3 410665 1110006.9

Second resonant mode (48.6 kHz) Third resonant mode (38.81 kHz)

R2 )( L2(H) C2(F) R3 )( L3(H) C3(F)

640.0795 410834 0110184.2 582.744 410483 01103.414

Table ‎8.2 Estimated Resonant Frequencies in Simulation

Method


Impedance Measurement 70.19 48.6 38.81


White Noise 70.31 48.83 39.06

Proposed 70.31 48.83 39.06

8.3.2. Experimental Results

Two different piezoelectric devices Type A and Type B were considered for experimental

evaluation. As was the case with the simulation testing, the impedance of both devices was

obtained in the frequency range of 30 kHz to 80 kHz. For the proposed method a power

converter was used for generating pulses. For the traditional and broadband excitation

methods a G5100A function waveform generator was used as the signal generator source and

the signals were amplified using an OPA549. Fig. 8.6 shows the experimental setup for the

proposed method. All signals were captured using a RIGOL DS1204B oscilloscope. As the

frequency was between 30 kHz and 80 kHz, the cut-off frequency of the filter was set to 100

kHz. Hence, according to equation (8-5), capacitor and resistor values of Cf =1nF and Rf

=1.59K respectively were selected for the low pass filter. The measured impedance

frequency behaviour of the Type A piezoelectric device for all methods is shown in Fig.

8.7(a). Table 8.3 also shows the extracted frequencies for the three strongest resonant

frequencies.

Chapter 8

237

Fig. 8.7(b) shows the impedance frequency response of the second piezoelectric device. The

estimated frequencies for the first three resonant frequencies are shown in Table 8.4. As can

be seen from the measured results, the white noise method did not result in accurate estimation

of Fr2 and Fr3 compared to the other methods. The main reason is that in each experiment the

level of power in white noise changes randomly over time, and may even go to zero at

particular frequencies. Therefore for resonances which are not especially strong (as was the

case for Fr2 and Fr3 in the Type B device) the results are unreliable.

It should be noted that the electrical circuit model of the piezoelectric device is a simple

model which is not able to perfectly model the piezoelectric device nonlinearity, time

variations and high frequency behaviour. That is why there are small differences between the

simulation and test results. The advantages and drawbacks of the various methods are

summarized in Table 8.5. The proposed method offers simplicity and high performance in

addition to its ability to be used for online systems.

Fig. 8.6 Experimental setup for the proposed method.

Chapter 8

238

(a)

(b)

Fig. 8.7 Experimental results obtained for the piezoelectric devices impedance response: (a) Type A, (b)

Type B.

Table ‎8.3 Estimated resonant frequencies of the Type A piezoelectric device

Method




White Noise 70.31 48.83 39.06


Proposed 70.31 48.83 39.06

Chapter 8

239

Table ‎8.4 Estimated resonant frequencies of the Type B piezoelectric device

Method




White Noise 78.13 44.92 68.36


Proposed 78.13 46.88 66.41

Table ‎8.5 Survey on advantages and drawbacks of mentioned methods

Method Advantages Drawbacks

Traditional Easy to implement

Low expense

Direct frequency output

Circuit stability required

Labour intensive

Not applicable for online process

Network Analyzer Provides complete impedance

frequency response

High accuracy due to calibration

High expense

Not applicable for online process

Not applicable on high power converter

Slow due to the complex measurement

procedure

Impulse Response

(Step Excitation)

Provides impedance frequency

response

Easy to generate with power

converter

Can be generated at high power

Can be applied for online process

Slower than the proposed method (because the

applied step is longer than the pulse).

Sensitive to the noise

White Noise

Excitation

Provides impedance frequency

response

Can be applied for online process

Not easy to generate especially at high power

Not possible to generate with power converter


Proposed Provides impedance frequency

response

Easy to generate with power

converter


Can be applied for online processes


Chapter 8

240

8.3.3. Ultrasound Interface

The performance of the proposed frequency estimation algorithm was evaluated in the

previous sections, but as already mentioned, exciting the device at its resonant frequency is not

enough to achieve maximum power conversion. The excitation signal harmonics also need to

be considered.

In this section the advantage of exciting a piezoelectric transducer using a multi-level

waveform at the resonant frequency compared with a uni-polar waveform is illustrated. For

the comparison, one multi-level waveform and one uni-polar waveform were generated with

peak to peak voltage of 120V at 39 kHz (Type A device resonant frequency).

To perform the evaluation, one pair of the Type A piezoelectric transducers was placed face

to face as sender and receiver. For the first experiment, a unipolar pulse was applied to one of

the piezoelectric devices and the voltage across the other one was captured. Fig. 8.8(a) shows

the applied voltages. To show their influence on piezoelectric devices the frequency responses

of applied and captured voltages are illustrated in Fig. 8.8 (b). As can be seen from Fig. 8.8(b)

the captured response contains several harmonics, due to the excitation signal having much

energy away from the fundamental frequency.

For the next test, a multi-level waveform was applied to the piezoelectric device. As can be

seen from Fig. 8.8(c), harmonic levels are attenuated significantly. This was due to the use of a

multi-level waveform which damped the harmonics in the vast area around the fundamental

frequency.

Comparing the Fig. 8.8(b) with Fig. 8.8(c), illustrates that the maximum energy is achieved

at the resonant frequency. Higher efficiency is obtained for multi-level signal. In particular the

method is quite effective for reducing the harmonic content. In addition, the presence of

harmonics not only adversely affects frequency sensitive applications, but also causes an

increase in temperature and an increase in power loss. Moreover, when using filters for

attenuating remaining harmonics the filter cost and size decreases when multi-level topology

is employed.

Chapter 8

241

(a)

(d)

Input voltage (unipolar)

(b)

Input voltage (multilevel)

(e)

Output voltage (unipolar)

(c)

Output voltage (multilevel)

(f)

Fig. 8.8 Obtained results for the ultrasound interface: (a) applied uni-polar pulse at 39 kHz in time domain, (b)

frequency response of the applied uni-polar pulse (input signal), (c) frequency response of the output signal when

the input is a uni-polar pulse, (d) applied multi-level pulse at 39 kHz in time domain, (e) frequency response of the

applied multi-level pulse (input signal), (f) ) frequency response of the output signal when the input is a multi-level

pulse.

8.4. Conclusions

In this paper, a new method is proposed for improving the efficiency of high power

ultrasound applications. It has been seen that the resonant frequencies vary under different

Chapter 8

242

system conditions. Moreover, the advantage of exciting the piezoelectric device at the exact

resonant frequency using multi-level waveform generation has been illustrated. An algorithm

which extracts the resonant frequencies and updates the system is therefore needed. The

proposed method has been compared with different methods and excitation signals. The

comparative results have been conducted based on experimentation and simulation. The

results obtained demonstrated the high performance of the proposed method.

8.5. References

[1] W. Heywang, K. Lubitz, and W. Wersing, Piezoelectricity. New York: Springer, 2008.

[2] C. Steinem and A. Janshoff, Piezoelectric Sensors. New York: Springer, 2007.

[3] T. Hall and C. Cain, "A Low Cost Compact 512 Channel Therapeutic Ultrasound System For

Transcutaneous Ultrasound Surgery," AIP Conference Proceedings, vol. 829, pp. 445-449,

05/08/ 2006.

[4] Y. Kui, K. Uchino, X. Yuan, D. Shuxiang, and L. Leong Chew, "Compact piezoelectric

stacked actuators for high power applications," Ultrasonics, Ferroelectrics and Frequency


[5] R. Li, N. Frohleke, and J. Bocker, "DESIGN AND IMPLEMENTATION OF A POWER

INVERTER FOR A HIGH POOWER PEIZOELECTRIC BRAKE ACTUATOR IN

AIRCRAFTS," presented at the 9th Brazilian Power Electronics Conference, 2003.

[6] S. Priya, "High power universal piezoelectric transformer," Ultrasonics, Ferroelectrics and


[7] A. M. Sanchez, M. Sanz, R. Prieto, J. A. Oliver, P. Alou, and J. A. Cobos, "Design of

Piezoelectric Transformers for Power Converters by Means of Analytical and Numerical

Methods," Industrial Electronics, IEEE Transactions on, vol. 55, pp. 79-88, 2008.

[8] C. Volosencu, "Methods for Parameter Estimation and Frequency Control of Piezoelectric

Transducers, Automation Control - Theory and Practice," in Automation Control - Theory and

Practice, A. D. Rodić, Ed., ed: InTech, 2009, pp. 115-136.

[9] E. Dallago and A. Danioni, "Resonance frequency tracking control for piezoelectric

transformer," Electronics Letters, vol. 37, pp. 1317-1318, 2001.

[10] D. Huijuan, W. Jian, Z. Hui, and Z. Guangyu, "Measurement of a piezoelectric transducer's

mechanical resonant frequency based on residual vibration signals," presented at the

Information and Automation (ICIA), 2010 IEEE International Conference on, 2010.

[11] T. Saar, O. Martens, M. Reidla, and A. Ronk, "Chirp-based impedance spectroscopy of piezo-

sensors," presented at the Electronics Conference (BEC), 2010.

[12] K. Agbossou, J. L. Dion, S. Carignan, M. Abdelkrim, and A. Cheriti, "Class D amplifier for a

power piezoelectric load," Ultrasonics, Ferroelectrics and Frequency Control, IEEE


[13] C. Kauczor and N. Frohleke, "Inverter topologies for ultrasonic piezoelectric transducers with

high mechanical Q-factor," in Power Electronics Specialists Conference, 2004. PESC 04.

2004 IEEE 35th Annual, 2004, pp. 2736-2741 Vol.4.

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[14] T. Sai Chun and G. T. Clement, "A harmonic cancellation technique for an ultrasound

transducer excited by a switched-mode power converter," Ultrasonics, Ferroelectrics and


[15] A. Nami and F. Zare, "Multilevel Converters in Renewable Energy Systems," in Renewable

Energy, T. J. Hammons, Ed., ed: InTech, 2009, pp. 271-296.

[16] J. Rodriguez, L. Jih-Sheng, and P. Fang Zheng, "Multilevel inverters: a survey of topologies,

controls, and applications," Industrial Electronics, IEEE Transactions on, vol. 49, pp. 724-

738, 2002.

[17] F. Zare and G. Ledwich, "A hysteresis current control for single-phase multilevel voltage

source inverters: PLD implementation," Power Electronics, IEEE Transactions on, vol. 17,

pp. 731-738, 2002.

[18] F. Zare and G. Ledwich, "A New Predictive Current Control Technique for Multilevel

Converters," presented at the TENCON IEEE Region 10 Conference, 2006.

[19] J. George K Lewis, George K Lewis, Sr, and William Olbricht, "Cost-effective broad-band

electrical impedance spectroscopy measurement circuit and signal analysis for piezo-materials

and ultrasound transducers," MEASUREMENT SCIENCE AND TECHNOLOGY, vol. 19, pp.

1-13, 2008.

[20] I. Getman and S. Lopatin, "Matching of series and parallel resonance frequencies for

ultrasonic piezoelectric transducers," presented at the Proc. Int. Symp. Applications of

Ferroelectrics.

[21] J. S. Kim, K. Choi, and I. Yu, "A new method of determining the equivalent circuit

parameters of piezoelectric resonators and analysis of the piezoelectric loading effect,"

Ultrasonics, Ferroelectrics and Frequency Control, IEEE Transactions on, vol. 40, pp. 424-

426, 1993.

[22] K. Kin Wing, H. L. W. Chan, and C. L. Choy, "Evaluation of the material parameters of

piezoelectric materials by various methods," Ultrasonics, Ferroelectrics and Frequency


[23] Y. Y. Lim, S. Bhalla, and C. K. Soh, "Structural identification and damage diagnosis using

self-sensing piezo-impedance transducers," Smart Materials and Structures, vol. 15, pp. 987-

995, 2006.

[24] A. L. Lopez-Sanchez and L. W. Schmerr, "Determination of an ultrasonic transducer's

sensitivity and impedance in a pulse-echo setup," Ultrasonics, Ferroelectrics and Frequency


[25] V. Loyau and G. Feuillard, "Relationship between electrical impedance of a transducer and its

electroacoustic behavior: Measurement without primary source," Journal of Applied Physics,

vol. 100, pp. 034909-034909-7, 2006.

[26] G. Mingjie and L. Wei-Hsin, "Studies on the circuit models of piezoelectric ceramics,"

presented at the Proc. Int. Conf. Information Acquisition, 2004.

[27] T. S. O. Märtens, M. Min, R. Land, M. Reidla, "Fast Impedance Spectroscopy of

Piezosensors for Structural Health Monitoring," ELECTRONICS AND ELECTRICAL

ENGINEERING, vol. 7, pp. 31-34, 2010.

[28] L. W. Schmerr, A. Lopez-Sanchez, and R. Huang, "Complete ultrasonic transducer

characterization and its use for models and measurements," Ultrasonics, vol. 44, pp. e753-

e757, 2006.

[29] R. Pintelon and J. Schoukens, "Basic Choices in System Identification," in System

Identification: A Frequency Domain Approach, ed: IEEE Press and John Wiley, 2001, pp.

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Chapter 8

244

[30] P. Davari and H. Hassanpour, "Designing a new robust on-line secondary path modeling

technique for feedforward active noise control systems," Signal Processing, vol. 89, pp. 1195-

1204, 2009.

[31] B. C. Baker, "Anti-aliasing, analog filters for data acquisition systems," in Application Note

no. AN699, Microchip Technology Inc., 1999.

245

Chapter

9 . C o n c l u s i o n s a n d F u r t h e r R e s e a r c h

9

Chapter 9

246

9.1. Conclusions

In the past decades, high power converters have penetrated into more and more applications,

playing as key role in industry. Regarding this, covering wide range of all related areas is not

practical, therefore in this thesis two main areas were considered. The main intention of this

research project was pulsed power technology. Investigating pulsed power area aimed at

applying power electronics techniques in order to develop solid-state pulsed power supplies

to meet application demands. On the other hand, high power ultrasound application regarding

to the high frequency issues was aimed as the second aspect of this PhD research. Three main

research objectives were identified as below:

Improving solid-state pulsed power supplies regarding to the low power rating and

operating speed limits of the current switching devices.

Considering load characteristics and effect of pulsed power supply

Increasing the efficiency of high power ultrasound applications

Considering the mentioned objectives, this dissertation set out to investigate power

converters at system and application levels for the two areas of pulsed power and high power

ultrasound. At system level different topologies were proposed to improve the system

efficiency regarding to the existing device’s limits and load demands, and at application level

the performance of the developed method was evaluated using real-world applications. In

each investigated area, the studies were started at system level and ended up at application

stage.

9.1.1. Improving solid-state pulsed power supplies regarding to the

low power rating and operating speed limits o f the current

switching devices

For the pulsed power area, employing different combination of flyback converter was

considered as the first step. The flyback converter was selected as it can generate high level

of the output voltages using a low input voltage, provides isolation and gives the ability of

controlling the input energy. To take advantage of current and voltage sharing, two flyback

configurations were considered. The proposed idea depicted the ability to increase the

performance of a pulsed power supply in terms of output voltage level and rate of rise based

on the simulation results. The preliminary concept of parallel and series configurations are

introduced in Chapter 2.

Chapter 9

247

To evaluate the proposed methods, laboratory prototypes were implemented with ability of

generating up to 4kV. The experimentations demonstrated the performance of the proposed

method in boosting up the voltage level and rate of rise comparing with the single module.

The further analysis and experimentations are depicted in Chapter 3. Analysing varied

operating conditions and effect of the load, illustrated that this technique can be utilized in

high impedance applications with the advantage of generating both DC and AC voltages.

The idea along with the simulation results are published in a conference paper entitled

“Parallel and Series Configurations of Flyback Converter for Pulsed Power Applications” at

The 7th

IEEE Conference on Industrial Electronics and Applications in July 2012 in

Singapore. The hardware implementation and experimental results are presented as a part of a

journal paper entitled “High Voltage Modular Power Supply Using Parallel and Series

Configurations of Flyback Converter for Pulsed Power Applications” published in IEEE

Transaction on Plasma Science Journal, Oct 2012.

The achieved outcomes illustrated that the series configuration is more practical. However,

two series modules were not enough to provide required voltage level and rate of rise. To

investigate the possibility of extending the proposed idea and benefiting from modularity

concept, a laboratory prototype based on 10 series flyback modules was implemented. This

research work is presented in Chapter 3. The capability of generating up to 40kV with

improved rate of rise (ten times comparing to a single module) completely proved the

effective performance of the proposed method. The analysis and experimentations were

published as a part of a journal paper entitled “High Voltage Modular Power Supply Using

Parallel and Series Configurations of Flyback Converter for Pulsed Power Applications”

published in IEEE Transaction on Plasma Science Journal, Oct 2012.

To preserve the modularity concept while making the pulsed power supply load independent

inspired the idea of employing switch-capacitor units supplying by a current source. The

ability of supplying the low impedance loads at high voltage levels and fast rise time made

the proposed topology capable of covering wide range of applications. Chapter 4 presented

extended analyses of this research. Moreover, employing a smart controlling algorithm made

the proposed method to prevent from any possible faults and easy diagnosis.

The proposed method was published and presented entitled “A Flexible Solid-State Pulsed

Power Topology” at the 15th

International Power Electronics and Motion Control Conference

and Exposition, Novi Sad, Serbia 2012.

Chapter 9

248

9.1.2. Considering load characteristics and effect of pulsed power

supply

To explore load characteristics and evaluate the designed power supply three real-world

applications were considered. The first application was investigating pulsed power supply for

exhaust gas treatment. In this regard DBD load characteristics were considered and based on

that a push-pull based power supply was implemented. Moreover, the effect of plasma

regarding to the generated voltage level on particle matters were studied. The measured

results depicted the availability of optimum operating point where gas treatment can be

performed while small size particle production is prevented. Chapter 5 presented extended

description about this work. The pulsed power supply take advantage of resonance

phenomena happening between the converter and the DBD capacitance to generate high level

of voltages. This idea made the developed pulsed power supply suitable for high impedance

capacitive loads. This study entitled “Effect of Pulsed Power on Particle matter in Diesel

Engine Exhaust Using a DBD Plasma Reactor” is published in IEEE Transactions on

Plasma Science Journal, Aug 2013.

To study another type of application, exciting plasma lamp was selected. As the plasma

lamp is a DBD load, the same push-pull pulsed power supply applied for the exhaust gas

treatment was considered here. The obtained results are discussed in Chapter 6. Controllable

parameters of the developed pulsed power supply (adjustable output voltage and repetition

rate) make it suitable for plasma lamp applications. The main reason is due to possibility of

exciting the plasma lamp at variety of power consumption and intensity levels by combining

different operating voltages and frequencies. Moreover, the measured results showed

nonlinear behaviour of plasma lamp which provides the feasibility of selecting an optimum

operating point.

This work entitled “Analysing DBD Plasma Lamp Intensity versus Power Consumption

Using a Push-Pull Pulsed Power Supply” is accepted in 15th

European Conference on Power

Electronics and Applications to be held in Lille, France, 2013.

Chapter 9

249

9.1.3. Increasing the efficiency of high power ultrasound applications

The efficiency of piezoelectric transducer can increases by generating a proper excitation

signal regarding to its resonance frequencies and operating conditions. Therefore, to indentify

the properties of the required excitation signal, as the first step, characteristics of a high

power piezoelectric transducer was investigated under varied situations. The complete

analysis of this research is presented in Chapter 7. The extensive experimentations under

varied mediums, power levels and voltage waveforms illustrated the nonlinear behaviour of

the piezoelectric transducer under different excited frequencies. This shows that performance

of a high power ultrasound system not necessarily increases with the increment in the input

power level.

This work was published and presented entitled “Power Electronic Converters for High

Power Ultrasound Transducers” at The 7th


Applications in July 2012 in Singapore.

Regarding to the obtained results from studying the piezoelectric transducer behaviour,

generating an excitation signal using multilevel converters found to be advantageous. In

addition, due to effect of environmental situation on resonance frequency variations

employing an adaptive frequency detection algorithm found to be beneficial. Further

description of this study is provided in Chapter 8. The proposed adaptive algorithm not only

follows the resonance frequency variations but also update the power converter in order to

generate the proper excitation signal.

This work entitled “Improving the Efficiency of High Power Piezoelectric Transducers for

Industrial Applications” was published in IET Science, Measurement and Technology

Journal, in Feb 2012.

Chapter 9

250

9.1.4. Summary of advantages and drawbacks of proposed topologies

and methods

Parallel Configuration of Flyback Converter for Pulsed Power Applications

Ll

Lm

Vs

1 : n

Co2

Ll

Lm

n : 1

Co1Load

Vo

+

-


Advantages Improves rate of rise

Requires low input voltage

Provides insulation between the load side and the input side

Ability to control the energy flow

Provides both AC and DC output voltage

Requires low power rating switches as it provides current sharing at

the input side

Drawbacks Load dependent

Requires careful high voltage insulation consideration for the

transformer

The components at the load side should tolerate the whole generated

voltage

Chapter 9

251

Series Configuration of Flyback Converter for Pulsed Power Applications

Ll

Lm

Vs

1 : n

Co1

Load

Ll

Lm

1 : n

Co2

Vo

+

-

Converter (2)

Converter (1)

Advantages Improves rate of rise





Extendable to higher number of modules

Requires low power rating switches as it provides:

Current sharing at the input side

Low reflected output voltage across the switches



transformer

Chapter 9

252

Modular Pulsed Power Supply Based on Series Configuration of Flyback Converter

VS

Lo

ad

Control

protocol of

switches

VLoad

+

-

Vo1

+

-

Vo2

+

-CO

CO

CO Vo3

+

-

CO Vo4

+

-

CO Vo5

+

-

Vo6

+

-

Vo7

+

-

Vo8

+

-

Vo9

+

-

Vo10

+

-

CO

CO

CO

CO

CO

Me

asu

rin

g th

e C

urr

en

t

Advantages Improves rate of rise and output voltage level


Easy diagnostic as it is modular




Requires low number of switches

Based on low power rating switches as it provides current and voltage

sharing

Drawbacks Only applicable for high impedance applications

Requires good synchronization between the gating signal

Chapter 9

253

Flexible Pulsed Power Topology Based on Switch-Capacitor Units

S11

S12

S21

S22

Sn1

Sn2

SL

Load

Si SVVS

L

D1

D2

C1

C2

Cn

Posi

tive

buck

-boost

conver

ter

Load Switch

Control

protocol of

switches

Sw

itch

-cap

acit

or

unit

s

Hysteresis

current control

Hysteresis

voltage control

Ov

er-r

ide

sig

nal

Ov

er-r

ide

sig

nal

Voltage

Feedback

Synchronizing

PWM Signal

Advantages Flexible in terms of providing:

Adjustable output voltage

Controllable repetition rate

Controlling the energy flow

Load independent (suitable for any kind of application regardless

of the load impedance)

Provides high dv/dt

Requires low voltage components

Insulates the load side from the input side

Transformer-less

Provides protection and fault detection

Easy diagnostic as it is modular

Drawbacks D2 should be stack of series diodes as it should tolerate the whole

generated voltage

Chapter 9

254

Utilizing Push-Pull Converter for Exhaust Gas Treatment

S2

S1

Vin

DB

D

NA

NB

NS

CS

CS

NA = NB << NS

Advantages Ability to generate high level of the voltages using low voltage

switches

Requires only two switches

Provides easily controllable parameters (output voltage and repetition

rate)

High dv/dt

Compact

Insulates the load side from the input side

Suitable for DBD loads as it can discharge the load

Effective for exhaust gas treatment



transformer

Chapter 9

255

Developing a Pulsed Power Supply for Plasma Lamp Applications

S2

S1

Vin DB

D L

am

p

NA

NB

NS

CS

CS

NA = NB << NS

Cm

Advantages Ability to generate high level of the voltages using low voltage

switches

Requires only two switches

Effective for plasma lamp applications as it provides easily

controllable parameters (output voltage and repetition rate) in order to

control plasma lamp intensity and power consumption

High dv/dt

Compact

Suitable for DBD loads as it can discharge the load



transformer

Chapter 9

256

Improving the Efficiency of High Power Piezoelectric Transducers

DC/AC

Inverter

AC

Current Sensor

Piezoelectric

Frequency

Estimation

Advantages Detects piezoelectric transducer resonance frequency variations and

updates power converter fundamental frequency

Provides complete impedance frequency response

Easy to generate with power converter


Can be applied for online processes

Fast computation due to low complexity

Cost effective

Drawbacks Required filter as it is noise sensitive

9.2. Further Research

This research study has focused on developing high frequency high power converters with

intention of applying for pulsed power and high power ultrasound applications. Moreover, the

experiments and investigated applications also yielded a large amount of valuable data

concerning converter development under different pulse conditions and load configurations.

Regarding this suggestions for further research work are discussed in six specific areas.

Chapter 9

257

Hardware implementation of the proposed flexible switch-capacitor unit based

pulsed power supply

To investigate the performance of the proposed flexible topology (Chapter 4) performance on

different applications, a hardware implementation is required. The implemented hardware

setup can cover wide range of applications especially liquid discharge applications (low

impedance load). This is quite important as recently, due to ability for employing them for

biomass production, liquid discharge applications have gained lot of interest.

Implement and investigate a flexible pulsed power supply with bipolar modulation

The proposed flexible method shows variety of advantages regarding to the pulsed power

applications. However the developed topology is capable of generating unipolar voltages,

while recent studies shows the advantages of using bipolar voltage waveform. This is due to

the fact that with bipolar modulation it is possible to generate high peak to peak value across

the load while using switches with the same break down voltage as in unipolar modulation.

Hence, studying bipolar modulation using switch-capacitor units is recommended for the

future research.

Investigating the effect of electrodes shape and different geometries on the pulsed

power performance

Optimizing the designed reactor by considering the electrodes shape and applied geometry is

effective on the generated plasma performance. This can reduces the required voltage level

and rate of rise as the reactor capacitance can adversely affect the pulsed power supply.

Investigating the effect of different geometry and electrodes shape combinations on electric

field distribution using Finite Element simulations is suggested as the first step. As the

second step the by employing the implemented hardware setup it is possible to optimize the

required reactor setup.

Study the effect of negative and positive discharges on different applications

Plasma discharge can be positive or negative. This is determined by the polarity of the

voltage on the electrode with a high potential gradient. The characteristics of positive and

negative plasma are totally unlike. This difference causes varied phenomena, regarding to the

proposed and implemented pulsed power supplies it is advantageous to study the effect of

positive and negative discharges in different applications.

http://en.wikipedia.org/wiki/Potential_gradient

Chapter 9

258

Pulsed Power Applications

Regarding to the presented information and experimentation of pulsed power applications

and considering the recent interests in bio-applications investigating the following

applications can be quite beneficial. This study not only can improve and evaluate the

performance of the designed pulsed power supplies, but also there is a high possibility of

commercialization due to recent interests and investments in these types of applications:

1-Plasma Gasification

Gasification is a process which relates to conversion of organic or fossil based carbonaceous

materials into syngas. In other word, it is a process of converting waste into gaseous fuel. In

recent years plasma gasification has gained lot of interest. However, increasing the system

performance and efficiency using power electronics techniques is one of the active areas.

QUT has conducted a cleantech project based on plasma gasification. One of the goals in this

undergoing project is finding an optimum operating point with respect to power consumption

point of view and process outcome.

2-Biomedical applications

The generated UV, ozone, and active radicals due to the presence of plasma found to be

extremely effective in killing harmful bacteria and generally sterilization. Applying pulsed

power supplies can increase compactness and reduces the time required for the process.

Investigating different types of plasma discharges, effect of AC or DC voltages, and effect of

dv/dt can be advantageous in order to provide a protocol for biomedical applications.

3- Liquid discharge applications

Production of biomass has recently gained lot of interest due to the intention of switching

from fossil fuels to biofuels. Hence, investigating an efficient method to ease pre-processing

stage of converting biological matter into energy products is extremely important. For the

pulsed power supply applications this process involves within the liquid discharge

applications. As improvements of solid-state topologies for liquid discharge applications

were quite slow (low impedance application drawbacks), there is a high research potential in

this field.

Chapter 9

259

4- Biofuel treatment

Recent applications show a high interest of using biofuels instead of diesel. Investigating the

effect of plasma on particle matter distribution and mass is an interesting research, as the

diesel exhaust gas treatment has been studied in this research.

Applying current source based converters using adaptive algorithm for high power

piezoelectric transducer applications

The extensive experiments conducted on the piezoelectric transducer showed capacitive

behaviour of the piezoelectric device. Therefore, applying current source based converter is

more efficient and can reduce lot of spikes generated across the output. In addition, the

efficiency of the ultrasound system can be improved by employing a resonance frequency

detection algorithm. Hence, developing a current source converter along with an adaptive

algorithm is recommended as a potential future work in this field.

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