Final_29_09_v3 (1)

In partial fulfilment of the requirements for the degree of

MASTER OF SCIENCE

in MICRO-ELECTRONICS

at the Delft University of Technology, to be defended publicly on Wednesday September 24, 2014 at 15:00hrs

Supervisors : 1) Prof. Ir. Dr. C.J.M Chris Verhoeven,

Associate Professor,

TU Delft

2) Ir. Waldemar Lubbers,

RF & E head,

Innovative Solutions In Space B.V, Delft

Thesis Committee : 1) Prof. Ir. Dr. C.J.M Chris Verhoeven (TU Delft)

2) Ir. Waldemar Lubbers (ISIS BV)

3) Prof. Dr. Nick van der Meijs (TU Delft)

4) Ir. P.P.(Prem) Sundaramoorthy (TU Delft)

An electronic version of this thesis is available at http://repository.tudelft.nl/

The work in this thesis was supported by ISISpace B.V. Their cooperation is hereby gratefullyacknowledged.

Copyright c©All rights reserved.

[Dedication]

I dedicate this work to my mom, dad, vikku and my “Small Wall friends”….

Acknowledgement I would like to express my sincere gratitude to the management of Innovative Solutions

In Space B.V for allowing me to work on such a fascinating research topic for my Master’s

thesis. They have provided me with some of the best possible resources to carry out my work

smoothly.

I would like to thank my university supervisor Prof. Chris and company supervisor Waldemar

for providing me the support, guidance and encouragement through the entire period of my

thesis. I am sure their guidance will go a long way in my career as a researcher.

The RF&E team at ISISpace B.V provided a great environment to exchange ideas. I

would like to thank Arancha, Eelco, Johan, Javier, Stefano, Wouter2 and Waldemar for making

me feel a part of the RF&E team and provide guidance and support and some very critical times

of my thesis.

I would like to express my gratitude to Ernst Habekotte from CATENA

microelectronics B.V for supporting this thesis at a crucial time by providing the LINC

development boards to understand the design and introducing me to the Mixed signal group at

TU Eindhoeven. I am grateful Prof. Baltus, P.G.M and Rainier from the mixed signal group at

TU Eindhoeven for allowing me to use the measurement setup there and providing an un-

conditional support through my entire stay.

I would like to thank my parent back in India who always had the confidence in me, no

matter what and always would be there for me as an emotional support. A very special thanks

to SKYPE and Facebook, I have not missed my home country!

My twin brother has played a very important part during my studies, he has been my

pillar of support ever since I came to Netherlands. I have almost never missed home.

It will not be complete without thanking my friends here in Delft. I surely did make a

lots of friends, they have supported me through the entire two years of my studies. A special

thanks to the “Small Wall” group of friends: Harshitha, Abhimanyu, Adithya, Arun, Arul,

Manjunath, Phani, Sriram and Sumedh. You guys were simply amazing. I would like to thank

Arun for helping me out in making the cover page of this thesis.

-Vishu

Preface

The fast growing interest in nano-satellite development and the use of nano-satellite for complex missions has led to an increase in the downlink data-rate. The availability of free bands (amateur frequency bands: VHF and UHF) is getting limited and more challenging to do the frequency coordination. This can be attributed to the surge in number of nano-satellite being launched. There has been 430% increase in the number of satellites launched in 2013 compared to 2012. This trend has led the nano-satellite developers to start looking into higher frequency bands and S-band has been a popular choice. But, going by the trend, frequency coordination in the S-band can also get challenging. Thus, there is a need to develop a nano-satellite transmitter that uses spectrally efficient modulation scheme, at the same time it has to be power efficient and be compatible with some of the existing communication standards. This work deals with developing the requirements on a nano-satellite transmitter based on mission case study, look at the need for efficiency and linearity enhancement technique for the transmitter, select the most appropriate architecture for nano-satellite application and show a proof of concept using a prototype. Using system engineering approach the efficiency and linearity enhancement technique that was chosen was LINC architecture. ADS simulations were performed to understand different configurations of LINC and finally, measurements were performed on the chosen configuration to characterize its efficiency and linearity. 16-QAM, 16-APSK, 32-QAM, 32-APSK and 64-QAM modulation schemes were implemented and tested. The best EVM was obtained for 16-APSK with 20 degrees compensation stub in the Chireix combiner which was 0.27% and the best efficiency was obtained for 16-APSK modulation scheme which was implemented without any compensation stubs. Based on the lessons learnt during simulations and measurements, recommendations are provided to improve the LINC configuration and improve the reliability of the measurement setup.

Visweswaran Karunanithi

Delft, September 2014

11

MSc Thesis, Visweswaran Karunanithi

Contents

[Dedication] ............................................................................................................................... 5

Acknowledgement....................................................................................................................... 7

Preface ....................................................................................................................................... 9

Table of figures ........................................................................................................................ 15

1. Introduction...................................................................................................................... 20

1.1. Current trends in nano-satellites ...................................................................................... 21

1.2. Conclusion of the analysis ................................................................................................ 24

1.3. Problem statement............................................................................................................. 25

2. Nano-Satellite transmitters for the future ........................................................................ 28

2.1. CCSDS and ECSS recommendations ................................................................................ 28

2.1.1. Frequency band allocation & constraints ........................................................... 29

2.1.2. Transmitter spurious emission and harmonic levels............................................ 31

2.1.3. Recommended modulation schemes ..................................................................... 31

2.1.4. Conclusion of the CCSDS and ECSS recommendations...................................... 32

2.2. Nano-satellite mission case study ..................................................................................... 33

2.2.1. Spacecraft housekeeping data downlink requirement ......................................... 34

2.2.2. Payload downlink data-rate requirement for a EES service ............................... 34

2.2.3. Downlink data-rate requirement for nano-satellite providing data services ...... 36

2.3. Link budget analysis..................................................................................................... 38

2.4. Summary of the case study and conclusion.................................................................. 40

3. High efficiency transmitter architectures ........................................................................ 43

3.1. Overview ........................................................................................................................... 43

3.1.1. Characterization of linearity and efficiency .............................................................. 45

3.2. Doherty architecture .................................................................................................... 47

3.2.1. Working Principle ................................................................................................ 47

3.2.2. Advantages and dis-advantages of Doherty architecture .................................... 49

3.3. Kahn/EER (Envelope Elimination and Restoration) ................................................... 49

3.3.1. Working principle of Kahn/EER architecture...................................................... 49

3.3.2. Advantages and dis-advantages of Kahn architecture ........................................ 50

3.4. Envelope Tracking (ET) architecture .......................................................................... 50

3.4.1. Working principle of ET....................................................................................... 50

3.4.2. Advantages and disadvantages of ET architecture .............................................. 51

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3.5. Switched Capacitor Digital Power Amplifiers SCDPA ............................................... 51

3.5.1. Working principle of SCDPAs ............................................................................. 51

3.5.2. Advantages and dis-advantages of SCDPAs........................................................ 52

3.6. LINC (Linear amplification using non-linear components) architecture ......................... 53

3.6.1. Working principle ...................................................................................................... 53

3.6.2. Advantages and dis-advantages of LINC architecture .............................................. 54

3.7. Trade-off analysis ............................................................................................................. 55

3.8. Conclusion ........................................................................................................................ 56

4. LINC Architecture ............................................................................................................ 57

4.1. Class of operation ............................................................................................................. 57

4.1.1. Class-F implementation ............................................................................................. 58

4.2. Different power combining techniques ........................................................................ 62

4.3. Experimental setup............................................................................................................ 72

4.3.1. Single RF source measurement setup ........................................................................ 72

4.3.2. Modifications done to the original setup ................................................................... 76

4.4. Catena LINC/Chireix PA architecture.............................................................................. 77

5. Measurement Results ....................................................................................................... 81

5.1. Modulation schemes tested ............................................................................................... 81

5.2. Performance characterization: EVM ................................................................................ 82

5.2.1. 16-QAM modulation scheme................................................................................ 82

5.2.2. 16-APSK modulation scheme ............................................................................... 84


5.2.4. 32-APSK modulation scheme ............................................................................... 88


5.2.6. Summary of EVM and ACLP measurement. ........................................................ 90

5.3. Performance characterization: LINC/ Chireix efficiency................................................. 91

6. Conclusion and Recommendations for future work ......................................................... 93

6.1 Recommendations on communication standards. .............................................................. 93

6.2 Recommendations for a LINC architecture ....................................................................... 94

6.2.1. PA cell .................................................................................................................. 94

6.2.2. Combiner architecture ......................................................................................... 95

6.3 Recommendations for the experimental setup ................................................................... 95

6.4 Future work ........................................................................................................................ 96

Nomenclature ........................................................................................................................... 99

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Bibliography .......................................................................................................................... 101

Table of figures Figure 1. Number of nano-satellite launches.......................................................................... 21

Figure 2. (Left) Mission types between 2003 and 2012, (right) Mission types in 2013. ........ 22

Figure 3. (Left) data-rates used by nano-satellite mission between 2003 and 2012 (right)

data-rates used by nano-satellite missions in 2013. ................................................................ 22

Figure 4. (Left) popular frequency bands used between 2003 and 2012. (Right) Frequency

bands used by nano-satellite missions in 2013. ....................................................................... 23

Figure 5. Maximum allowable bandwidth vs symbol rate. ..................................................... 30

Figure 6. Spectral emission limits........................................................................................... 31

Figure 7. Pictorial representation of the relation between downlink data-rate, service

provided and type of payload. .................................................................................................. 33

Figure 8. Pictorial representation of VDES []. ...................................................................... 36

Figure 9. Link budget calculation of spacecraft housekeeping data downlink. ..................... 38

Figure 10. Link margin for different elevation angles. ............................................................ 38

Figure 11. Link budget for an optical payload data downlink. .............................................. 39

Figure 12. Link margin vs elevation angles for optical payload downlink. ............................ 39

Figure 13. Link budget for VED-Sat downlink. ...................................................................... 40

Figure 14. Link margin vs elevation for VDE-Sat. ................................................................. 40

Figure 15. Block-diagram of a transmitter with digital modulation. ..................................... 43

Figure 16. Linearity vs Efficiency trade-off for conventional PA classes. ............................. 44

Figure 17. Efficiency and linearity enhancement techniques. ................................................ 45

Figure 18. PDF of 16-APSK modulation scheme, implemented using an SRRC filter with a

roll-off of 0.5(4000 sps) ........................................................................................................... 46

Figure 19. Doherty PA architecture. ....................................................................................... 47

Figure 20. Schematic representation of Doherty. .................................................................... 47

Figure 21. Input power vs Output power in linear scale [11] ................................................ 48

Figure 22. Block diagram representation of a modern implementation of the Kahn

architecture. ............................................................................................................................. 50

Figure 23. Block diagram of ET architecture.[11]................................................................. 51

Figure 24. Block diagram representation of SCDPA. ............................................................ 52

Figure 25. LINC block diagram.............................................................................................. 53

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Figure 26. 16-APSK decomposition into constant envelope signal. ....................................... 54

Figure 27. Output spectrum of 16-QAM for different classes of operation. [21]................... 58

Figure 28. Functional representation of Class-F PA ............................................................. 59

Figure 29. Class-F harmonic matching circuit. ..................................................................... 60

Figure 30. Load impedance by the drain of the transistor at different harmonic frequencies.

.................................................................................................................................................. 60

Figure 31. IV characteristics of CGH27015F. ....................................................................... 61

Figure 32. Drain voltage and current waveform for a Class-F PA implemented using

CGH27015F GaN HEMT. ....................................................................................................... 61

Figure 33. RF power combining techniques for LINC. .......................................................... 62

Figure 34. WPC as a power combining network for LINC .................................................... 63

Figure 35. Plot of the two out-of-phase signals, combiner voltage waveform and efficiency of

WPC as a function of outphasing angle. .................................................................................. 63

Figure 36. Improvement of isolation by introducing a quadrature coupler between the PA

and WPC. ................................................................................................................................. 64

Figure 37. Coupling between the two ports. ........................................................................... 64

Figure 38. Efficiency and voltage waveforms when a quadrature coupler is introduced

between the PA and combiner.................................................................................................. 65

Figure 39. Schematic representation of Chireix combiner. .................................................... 65

Figure 40. Chireix combiner as seen at the inputs when fed out-of-phase signals. ............... 66

Figure 41. Chireix combiner design on FR4 using CPW. ...................................................... 67

Figure 42. S-parameter simulation of the Chireix combiner with compensation stubs placed

at 10 deg. .................................................................................................................................. 67

Figure 43. Efficiency vs out-phasing angle comparison between Chireix combiner and WPC.

.................................................................................................................................................. 68

Figure 44. Comparison of coupling between Chireix combiner and WPC. ............................ 68

Figure 45. Radiation pattern of crossed dipole on a 3U cube-sat fed with an out-phasing

angle of 45 degrees. ................................................................................................................. 69

Figure 46. Radiation pattern of the crossed dipole when fed with signals with 0 degree

outphasing angles. ................................................................................................................... 70

Figure 47. Radiation pattern of crossed dipole fed with signals with 90 degrees out-phasing

angle. ........................................................................................................................................ 70

Figure 48. Complete measurement setup................................................................................ 72

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Figure 49. Signal flow diagram. ............................................................................................. 73

Figure 50. Initial setup for calibration. .................................................................................. 74

Figure 51. Screenshot of calibration measurement. ................................................................ 74

Figure 52. Setup to measure the non-linearity caused by mini-circuits PAs.......................... 75

Figure 53. Picture of the measurement setup. ........................................................................ 75

Figure 54. Variation in the constant envelope signal due to the previous SCS algorithm. .... 76

Figure 55. Constant envelope signals after the modified SCS algorithm. ............................... 77

Figure 56. LINC/Chireix PA from CATENA Microelectronics. ............................................. 78

Figure 57. Schematic representation of the LINC/Chireix PCB from CATENA .................... 79

Figure 58. Constellation diagrams of 16-QAM, 16-APSK, 32-QAM, 32-APSK and 64-QAM.

.................................................................................................................................................. 81

Figure 59. Received constellation diagram of 16-APSK when no compensation stubs (0

degrees compensation)............................................................................................................. 82

Figure 60. Output spectrum of amplified 16-QAM. ................................................................. 83

Figure 61. 16-QAM constellation diagram with 20 degrees compensation stub. .................. 83

Figure 62. Output spectrum of amplified 16-QAM with stubs................................................ 84

Figure 63. Constellation diagram of 16-APSK modulation without compensation stubs. ..... 84

Figure 64. Output spectrum of 16-APSK modulation without compensation stubs. .............. 85

Figure 65. 16-APSK constellation diagram with a compensation stub of 20 degrees. .......... 85

Figure 66. Output power spectrum of 16-APSK with a compensation stub of 20 degrees..... 86

Figure 67. 32-QAM without compensation stubs. .................................................................. 86

Figure 68. Output power spectrum of 32-QAM without compensation.................................. 87

Figure 69. Constellation diagram of 32-QAM with compensation stub at 20 degrees. ......... 87

Figure 70. Output power spectrum of 32-QAM with a compensation at 20 degrees. ............. 88

Figure 71. Constellation diagram of 32-APSK with 20 degrees compensation stubs. ........... 88

Figure 72. Output power spectrum of 32-APSK modulated signal with a compensation stub

at 20 degrees. ........................................................................................................................... 89

Figure 73. Constellation diagram of 64-QAM with 20 degree compensation in the

LINC/Chireix amplifier. ........................................................................................................... 89

Figure 74. Output power spectrum of 64-QAM amplified by LINC/Chireix PA with 20

degrees compensation stubs. .................................................................................................... 90

Figure 75. Harmonic matching circuit for the Class-F PA on CATENA LINC board. ........... 94

Figure 76. Proposed measurement setup................................................................................ 95

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Figure 77. Hybrid-Chireix combiner. ..................................................................................... 96

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1 1. Introduction

he popularity of nano-satellites among universities and space research organizations has grown quite rapidly over the last decade. Nano-satellite are a class of small satellites that weigh less than 10 Kilograms, having almost all of the subsystems present in a larger satellite and capable of mimicking larger satellites with simpler payloads. The interest for nano-satellites among universities started

in 1999 through a CubeSat1 project that began at California Polytechnic State University (Cal Poly) and Stanford University’s Space Systems Development Laboratory (SSDL). These projects were mainly intended to be used by the Universities as a tool to learn satellite development and understand the technology of developing hardware that can sustain the harsh space environment. These missions mainly made use of Commercially Of-The-Shelf (COTS) components which are industrial grade and easily available. This reduced the development cost and time of the mission. On the other hand, it reduced the life-time of a mission and also a high probability of failure. The failures and lessons learnt from past nano-satellite missions has helped in coming-up with mature designs that help in longer mission duration. An example of this is Delfi-C32 that was developed by a team from Technical University of Delft and launched on 28th April 2008. This mission is a good example of a COTS based but robust design that has survived multiple solar events and is still operational. The focus of nano-satellite missions has slowly started to shift from simple technology demonstration missions to more complex industry driven technology demonstration, science, military and government missions. This clearly shows a paradigm shift in the way nano-satellites are being perceived and the confidence in implementing more and more complex missions on nano-satellites. One of the most important reason for this is the standardization in shape, satellite-bus subsystems such as antenna systems, electrical power systems, On-board computer, etc. This has helped the satellite developers to concentrate mainly on payload development and integrate the payload with the satellite bus components and subsystems that are available on the CubeSat market. They are developed by companies such as Innovative Solutions In Space BV, GomSpace, etc. As the complexity of nano-satellite missions grow, the amount of data generated by the spacecraft (payload data and telemetry) has a direct influence on the downlink data-rate of the communication link. The subsequent sections in Chapter-1 give an overview of the current 1 CubeSats is a standardization introduced by Calpoly that signifies a satellite that is in the shape of a cube. A single unit of CubeSat is a cube of 10 × 10 × 10 = 1000 cm3 by volume and a mass lesser than 1.33 kg, also called 1U. This standardization has mainly helped in using the same type of deployment system for various classes such as 1U, 2U and 3U. 2 http://www.delfispace.nl/delfi-c3

T

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trends in nano-satellites, the paradigm shift in the nano-satellite mission types, how this paradigm shift has caused the missions to start using high data rate transmitters and conclude with the problem statement of this thesis work. In Chapter-2 a few mission cases are considered to quantize the requirements posed by complex missions on the transmitter design with the help of link budget calculations, state the CCSDS (Consultative Committee for Space Data Systems) and ECSS (European Cooperation on Space Standardization) recommendations that have to be followed in-order to design a transmitter that is compatible with a large number of ground stations around the world that follow these standards and conclude with recommendations. Chapter-3 will elaborate about the challenges in developing a high efficiency transmitter that support the modulation schemes recommended in CCSDS space link protocol over ETSI DVB.S2, some of the common efficiency and linearity enhancement techniques and conclude with a trade-off that was performed on these techniques to find a suitable candidate for nano-satellite transmitters. Chapter-4 deals with the LINC architecture which is one of the efficiency and linearity enhancement techniques, where the design methodology is discussed with the help of simulation results followed by an explanation of the measurement setup used to validate the performance of the LINC architecture. Chapter-5 discusses the measurement results and Chapter-6 provides the conclusions and future work.

1.1. Current trends in nano-satellites

In-order to understand the current trends in nano-satellite missions, a database of the nano-satellite missions that were launched between 2003 and November 2013 was made and various analysis were performed on this data. A total of 174 nano-satellites missions were launched during this period, of which 183 transmitters were flown. The focus of the information collected from these missions were: mission objective, year of launch, number of transmitters flown, communication mode (half-duplex or full duplex), transmitter data-rate, modulation schemes, transmit power and status of the mission. The database created can be found in

Figure 1. Number of nano-satellite launches.

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[Appendix]. Figure 1 shows the number of nano-satellites launched corresponding to the year. It can be seen that there is a steep increase in the number of missions launched in 2013.

There is an increase of 437% from 2012 to 2013. These numbers were compared with the Nano/microsatellite market assessment study that was later performed by SpaceWorks 3 in Feb 2014. The analysis converge and based on this analysis, the projections for 2014 show an increase of 2 – 3 times higher than the number of launches in 2013.

Based on the mission objectives and the payload flown, the nano-satellite missions were categorized into: education/technology demonstration, science/technology demonstration, military, Industry/technology demonstration, education/science, remote sensing, military/technology demonstration, technology demonstration, education and science missions. The following Pie chart shows the percentage of missions that belong to each of the category:

Figure 2. (Left) Mission types between 2003 and 2012, (right) Mission types in 2013.

From Figure 2, it can be seen that 36% of the missions in 2013 comprise of industrial technology demonstration, remote sensing, government/military missions compared to only 13% between 2003 and 2012. This change in trend shows that nano-satellites have started to be perceived as a serious contender to replace the bigger satellites. As the mission type has a

3 http://www.sei.aero/eng/papers/uploads/archive/SpaceWorks_Nano_Microsatellite_Market_Assessment_January_2014.pdf

Figure 3. (Left) data-rates used by nano-satellite mission between 2003 and 2012 (right) data-rates used by nano-satellite missions in 2013.

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direct influence on the downlink data-rate, it can be seen in Figure 3 the change in trend in terms of popular data-rates used by nano-satellites between this period: It can be seen that 43% of the missions used downlink data-rates below 1200 bps between 2003 and 2012, compared to only 23% of them missions using this option in the year 2013. Looking at the higher data-rate, only 9% of the mission used data-rates greater than 100 kbps between 2003 and 2012, compared to 24% of the mission in 2013 used a data-rate greater than 100 kbps. Due to the limited availability of frequency bandwidth in some of the conventionally used frequency bands such as VHF and UHF, as the downlink data-rate requirement increases, there is also a need to go to the higher frequency bands such as, S-Band, X-band and Ku-band where the larger bandwidths are available. The Figure 4 shows the trends in the frequency bands used by the nano-satellites.

It can be seen that the number of missions using S-band and X-band downlink has slightly increased in 2013. The amateur bands in UHF and VHF are a very popular choice for downlink. The main reason for this can be attributed to space heritage and the popularity of this band among radio amateurs around the world. This approach proves to be a win-win situation for both the satellite developers in receiving the spacecraft from all around the world, as a result increasing the reliability of a mission and at the same time benefit the amateur radio community to perform small experiments as a return favour. OSCARs (Orbiting Satellites Carrying Amateur Radios) in nano-satellites has gained popularity for the same reason. A good example is FUNcube4 which was launched in November 2013 and has received 256.5 MB of telemetry data so far (as of July 2014) from amateur radio operators around the world. This mission uses a simple 1200 bps BPSK downlink. With the support of the amateur radio operators, the telemetry of the satellite is available almost real-time, through the entire orbit.

4 FUNcube-1 (AO73) http://funcube.org.uk/

Figure 4. (Left) popular frequency bands used between 2003 and 2012. (Right) Frequency bands used by nano -satellite missions in 2013.

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1.2. Conclusion of the analysis

Based on the study done so far to determine the current trend and to see if there is really a need to start investigating the need for a high data-rate transmitter for a nano-satellite mission, the following conclusions were derived:

- First and foremost, the analysis done in the previous section, converges with the 2014 Nano/microsatellite Market Assessment that was conducted by SpaceWorks in February 2014. This market assessment also goes on to project a 2 – 3 times increase in the number of nano-satellite that will be launched in 2014, thus between 160 and 240, and more than half of the missions will be for Earth Observation/Remote Sensing.

- Apart from the total number of launches, nano-satellite cluster launches has also started to gain popularity. A Minotaur-1 rocket put 29 satellites into orbit in a single launch on 19 th November 2013 and DNEPR placed 32 satellites in a single launch on 21st November 2013. This has made frequency coordination really challenging as most of the nano-satellites preferred VHF and UHF bands which are very narrow bands and during initial days of the mission after the satellite separates from the launch adaptor, the satellites are very close to each other. This can result in interference issues. This would mean that satellites will have to start choosing frequency bands where a large bandwidth is available to avoid interfering with other satellites.

- As the complexity of the missions increase, the amount of data needed to be downlinked also increases. This has made the nano-satellite developers to start using higher data-rates, and as a result, start using S and X band.

- It was seen that a redundant low speed downlink transmitter (operating in amateur bands) were flown alongside high data-rate transmitter in various missions. The use of amateur frequency bands is likely to continue till a reliable standard is followed for high speed downlinks and ground stations around the world upgrade to support such missions.

- Spectrally efficient modulation scheme is the need of the hour. Considering the number of launches forecasted, there is a need to start investigating these modulation schemes for nano-satellite missions.

- It is wise to adopt some of the existing recommendations laid down by CCSDS and ECSS. The main advantage is, there are ground stations around the world that already support these standards and the recommendations are popular among the bigger satellites.

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1.3. Problem statement Based on the market study and preliminary analysis performed, it is evident that there is a need to develop high speed transmitters for nano-satellites that implement spectrally efficient modulation scheme and are complaint with ground station standards followed around the world (especially ESA ground stations). Using this as the motivation, recommendations stated in CCSDS and ECSS standards have to be studied and design constraints have to be derived. CCSDS also provides recommendations for spectrally efficient modulations schemes. It has to be investigated if these modulation schemes suit nano-satellite applications. Mission case study needs to be done using some of the on-going and future projects to see if there is really a need to implement such complex modulation schemes in nano-satellites. Implementation challenges needs to be studied and the need to implement efficiency and linearity enhancement technique in the transmitter design has to be investigated. If efficiency and linearity enhancement technique needs to be implemented, using system engineering approach, the most appropriate architecture for nano-satellite application needs to be chosen. Once the architecture is chosen, a prototype needs to be designed to show the proof of concept. The following diagram shows the work-flow of this thesis:

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2 2. Nano-Satellite transmitters for

the future ano-satellites in the past were mainly thought of as practical learning tools by the universities to teach engineering students about designing space hardware and encourage the interest in pursuing a career in the space industry. The main advantage of nano-satellites over the bigger satellites is: very short development

time and cheaper access to space. This attracted industries to start collaborating with the universities and started using nano-satellites as a test-bed to quickly test and space-qualify satellite hardware. A good example for this is Delfi-C3 which carried Thin Film Solar Cells (TFSC) from Dutch space and two Autonomous Wireless Sun Sensors (AWSS) from TNO as a part of its technology demonstration payloads [1]. Gradually over the period, a lot of standardization has lead the nano-satellite developers to start coming-up with more complex payloads. This change has posed the need to look at high data-rate transmitters that can meet the needs of mission requirements. This chapter starts with an overview of the recommendations laid by CCSDS and ECSS for space-to-earth links for satellites that fall under category-A5. Following the recommendations, a few nano-satellite mission cases are studied to see how the recommendations fit, link budget analysis is performed to derive the design requirements of the transmitter. This chapter concludes with some recommendations for the roadmap for nano-satellite transmitter design.

2.1. CCSDS and ECSS recommendations

CCSDS was founded in 1982 by major space agencies of the world, it is a multi-national forum for development of communications and data systems standards for spaceflight6. The CCSDS recommendations cover a broad spectrum: Space Internetworking Services, System engineering, Mission Ops & Information Management services, Cross Support Services, Spacecraft Onboard Interface Services and Space Link Services. The scope of this thesis limits to the following recommendations stated in Space Link Services: Radio Frequency and modulation systems (Blue book, CCSDS 401.0-B) [2], Bandwidth-efficient modulations (Green book, CCSDS 413.0-G-2) [3] and CCSDS Space link protocols over ETSI DVB-S2 standard (Blue book, CCSDS 131.3-B-1) [4].

5 Category-A are those satellites which orbit at an altitude of below 2 x 106 Km, Category-B are those satellites which orbit at an altitude above 2 x 106 Km (This category also includes deep space probes) 6 http://public.ccsds.org/default.aspx

N

http://public.ccsds.org/default.aspx

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2.1.1.Frequency band allocation & constraints

The frequency band allocation is done based on the service/operation provided by the satellites. The different services are: Space Research (SR), Earth Exploration Satellite Service (EES), Amateur Service and Space Operation (SO). The Table 1 gives the list of recommended frequency bands corresponding to the service it can provide.

Table 1. Frequency allocation corresponding to service provided.

Frequency bands (MHz) Service7

(L-Band) 1215 - 1240 SR, EES (L-Band) 1240 - 1300 SR, EES, Amateur (L-Band) 1525 -1535 SO, EES (S-Band) 2200 - 2290 SR, SO, EES (X-Band) 8025 - 8400 EES (X-Band) 8450 - 8500 SR

(Ka-Band) 25500 - 27000 SR, EES (Ka-Band) 37000 - 38000 SR

The constraints in using each of the frequency bands is as follows:

- L-Band: o This band shall not be used for feeder links of any service.

- S-Band (2200 – 2290 MHz): o The maximum occupied bandwidth for spacecraft in this band shall not exceed

6MHz. o The link shall be active only during the period when the satellite is in the

visibility cone of the ground station. (This will help frequency re-use) o The reliability of the device used to switch-off the transmitter shall

commensurate with mission lifetime.

7 Purely science missions such as radio-telescopes fall under the category of SR, Engineering missions such as communication satellites fall under the category of SO and earth observation missions such as Remote sensing fall under the category of EES.

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- X-Band (8025 – 8400 MHz): The constraints specified for S-Band applies for this band as-well, except for the bandwidth constraints.

- X-Band (8400 – 8450 MHz): o The maximum allowable bandwidth for a down-link in this band shall not

exceed the mask specified in Figure 5. - Ka-Band (25.5 to 27 GHz & 37 to 38 GHz): There were no constraints formulated for

this band but there has to be an agreement with the frequency coordinator prior to using this band.

- All the bands, up to X-band shall have a frequency stability not less than ± 2 × 10-5 (20ppm) under all conditions for the entire lifetime of the mission and for Ka-band, a frequency stability of ± 0.02 ppm/oC within the temperature range +10 oC to +40 oC.

The maximum allowed bandwidth corresponding to various data-rates is shown in the table below:

Table 2. Maximum allowed bandwidth corresponding to data-rate and frequency bands.

Frequency

band (MHz)

Operation Occupied Bandwidth

2200

– 2

290

[Dow

nlin

k]

And

84

50 –

850

0 [D

ownl

ink]

Telemetry (Rs < 10ksps)

300 kHz

Telemetry (10 ksps <Rs < 60 ksps)

1200 kHz or 30 × Rs (smaller of the two)

Telemetry (60 ksps < Rs < 2Msps)

1200 kHz or 12 × Rs, whichever is larger up to 6 MHz at 2 GHz and 10 MHz at 8 GHz.

Telemetry (Rs > 2Msps)

1.1 x Rs up to 6 MHz at S-band and 10MHz at 8GHz.

There are no bandwidth constraints stated for the frequency bands in Ka-band.

Figure 5. Maximum allowable bandwidth vs symbol rate.

31


2.1.2.Transmitter spurious emission and harmonic levels

The emitted spectrum for all the space science services shall adhere to the spectral emission mask shown in Figure 6. The total power contained in any single spurious emission shall not exceed -60 dBc. This constraint is valid for frequency bands from VHF to Ka-band. In the figure, the blue line indicated the boundary limits for data rate below 2 Msps, it has a slope of

43 dB per decade and the green line indicates the limit for data-rate above 2 Msps which has a slope of 60 db per decade. 2.1.3.Recommended modulation schemes

The occupied bandwidth is directly dependent on the data rate, the modulation schemes are recommended based on the data-rate is shown in Table 3

Table 3. Recommended modulation schemes.

Dat

a rat

e < 2

M

sps

When using Sub-carrier: - PCM/PM/Bi-phase. - PCM/PM/NRZ.

Suppressed carrier modulation (medium rate <2Msps) - BPSK - QPSK - OQPSK.

Dat

a ra

te >

2

Msp

s

Preferred modulation schemes when data rates are larger than 2Msps:

- GMSK (Pre-coding necessary) - Baseband filtered OQPSK - TCM 8PSK

Figure 6. Spectral emission limits

32


CCSDS space link protocols over ETSI DVB.S2 recommendations:

- QPSK - OQPSK - 16-APSK - 32-APSK - 64-APSK.

The downlink data-rates are categorized into high-speed downlink ( > 2Msps) and low-speed data-rate (<2 Msps). The modulation schemes such as PCM/PM/Bi-phase and NRZ are allowed only for data-rates below 2 Msps. The modulation schemes recommended for data rates below 2 Msps are all constant envelope signals and does not influence the performance of a HPA (High Power Amplifier) to a large extent when compared to non-constant envelope modulation schemes. In the case of digital modulation techniques such as BPSK, QPSK and OQPSK, although there is no inherent amplitude variation, but the use of SRRC (Squire Root Raised Cosine) filter in-order to reduce the ISI (Inter Symbol Interference) causes an amplitude variation, affecting the performance of the HPA. The non-linearity effects caused by OQPSK is lesser compared to QPSK at larger SRRC roll-offs [5]. For data-rates above 2 Msps, the modulation schemes proposed by CCSDS space link protocols over ETSI (European Telecommunication Standards Institute) DVB-S2 (Digital Video Broadcast – Satellite) include QPSK and OQPSK which are constant envelope schemes and a few additional modulation schemes such as 16-APSK, 32-APSK and 64-APSK. APSK stands for Amplitude Phase Shift Keying. As the name suggests, both phase and amplitude of the carrier are modulated in accordance with the message signal. It is similar to QAM (Quadrature Amplitude Modulation) but the PAPR of APSK modulation scheme is lesser than that of QAM. An in-depth analysis of PAPR of these modulation schemes and its effects on the performance of the PA will be discussed in later sections. 2.1.4.Conclusion of the CCSDS and ECSS recommendations

The recommendations stated in CCSDS and ECSS are a good starting point for standardization in the communication system for nano-satellites. The standards have already been implemented on the bigger satellites and ground stations around the world have the infrastructure compatible with CCSDS standards. Some quantitative analysis needs to be performed to determine if the modulation schemes recommended for data-rates larger than 2 Msps is required for nano-satellites with the help of mission case studies. The main challenge in implementing the recommended modulation schemes for nano-satellites is that, most of the spectral efficient schemes have a non-constant envelope profile. This variation in the amplitude can cause a degradation in the performance of the PA in terms of either efficiency or linearity, thus an analysis needs to be done to determine if one of the conventional operating classes (Class – A, AB, B, C, D, E, F, F-1 or J) is sufficient or efficiency/linearity enhancement technique needs to be performed. In case if an enhancement

33


technique needs to be implemented, a trade-off between the different architectures needs to be performed and the most suitable architecture needs to be chosen. In the subsequent sections of this chapter, a few nano-satellite missions will be studied to establish the requirements for a nano-satellite transmitter.

2.2. Nano-satellite mission case study

The downlink data-rate requirement for a space mission mainly depends on: 1) Number of sub-systems in the spacecraft 2) The amount of house-keeping data generated by each of the sub-system 3) The amount of payload data generated, which depends on the type of payload 4) The contact time with the ground station. The amount of data generated by the payload depends on the service provided by the mission. The Figure 7 shows the relation between the downlink data-rate requirement and the type of the missions.

Figure 7. Pictorial representation of the relation between downlink data-rate, service provided and type of payload.

34


2.2.1.Spacecraft housekeeping data downlink requirement

In this case, a general 3U nano-satellite LEO mission is considered to calculate the total amount of house-keeping data that will be generated by the spacecraft bus. The subsystems that are assumed in a generic satellite bus are: On-board computer, antenna system, transceiver (primary, secondary), electronic power system (Battery system, solar cells, etc.), attitude control system (sensor and actuator) and payload telemetry. In order to get a more realistic number, the satellite bus of TRITON-1 mission was considered. The house-keeping data from each of the sub-systems is periodically sampled by the on-board computer and stored in the on-board memory. Based on the available memory and mission requirements, either all of the stored telemetry is downlinked or only the mission critical data is sent to the ground station. In the case of TRITON-1, only some of the mission critical telemetry was downlinked. This case considers a hypothetical situation where all the sampled telemetry needs to be downlinked to the ground station. The size of each of the telemetry packets is approximately 16 bytes and it is assumed that each of the subsystems is sampled for telemetry on an average every 15 seconds. The total amount of house-keeping data generated was calculated to be 7.7 Mbits per day. The Table 4 gives the calculation done in-order to find the required downlink data-rate.

Table 4. Required downlink data-rate to downlink all the generated house-keeping data.

TT&C data rate requirement. Units

Typical sampling time 1 sec Data per subsystem 16 bytes 128 bits Duty-cycle 100% Data generated per day per subsystem 7.732E+05 bits Number of subsystems (approx.) 10 Total data generated per day 7.732E+06 bits Avg. GS contact time per day 50 min Percentage of contact time to transmit house-keeping data. 10 % Effective TT&C downlink data rate 24576 bps Data-overhead 50% Data rate including over-head 36864 bps

Considering that a nano-satellite will not have a dedicated telemetry downlink, the same link has to be shared between telemetry and payload data. Thus, in the above calculation it is assumed that 10% (a rough estimate) of the total contact time is used to downlink the telemetry data. The calculated downlink data-rate requirement is approximately 37 kbps. 2.2.2.Payload downlink data-rate requirement for a EES service

The Remote Sensing payloads mainly fall under the category of EES services. Some of the commonly used payloads under this service include optical imaging payload, infra-red imaging payload and SAR (Synthetic Aperture Radar). Among nano-satellites, optical imaging payload

35


has been the most commonly used payload. As a case study, the mission requirements stated for MISC-1 [6] mission will be considered. The mission was proposed to capture 7.5 m GSD (Ground Spatial Distance) multispectral imagery from an altitude of 540 km. The payload for this mission comprises of a 16 MP Kodak KAI-16000-CXA-JD CCD image sensor. The mission overview are stated in Table 5 below:

Table 5. MISC-1 mission overview

MISC mission units

Payload Multi-spectral imaging Volume 3U Altitude 540 km Image sensor KAI-1600-CXA Number of Pixel 16 MP Active pixel rows 4872 Active pixel column 3248 Number of Ground stations 13 Ground swath/image 702 Sq.km Target area per day 137,500 Sq.km Total number of images required per day 196 Size of a single un-compressed image (10 bit pixel depth) 19.78 MB

In the paper [6], it could be clearly seen that the mission was constrained by the downlink radio capabilities (38.4 kbps downlink), as a result the image had to be compressed to 1/12 th its original size and the mission required 13 ground stations to fulfill the requirement of downlinking 196 images per day. The mission was finally not realized using the above formulated mission requirements Now, the required downlink data-rate will be calculated for different cases to see an optimal solution for the mission to be feasible. Table 6. Downlink data-rate requirement when un-compressed image is transmitted and 13 ground stations are

used.

MISC mission downlink data rate

requirement

Number of images needed to be transmitted/day 196 Size of an image (Pixel depth assumed to be 10bits) 1.58E+08 bits Total image data generated/day 3.101E+10 bits Avg. Ground station contact time/day 36 min Total Ground station contact time (13 ground stations) 468 min effective payload downlink data rate 1.10E+06 bps Over-head 50% Total required downlink data rate 1.65E+06 bps

36


From the above calculation it can be seen that the downlink data-rate required to transmit un-compressed image using 13 ground stations is 1.65 Mbps. Similarly, two other cases were considered because the idea of using 13 ground stations to downlink the payload data from one satellite means highly synchronized operation between the ground stations which is complex. The two other scenarios that were considered are: 1) Downlink data rate requirement when one satellite and one ground station is used, 2) Downlink data-rate requirement when 3 satellites and one ground station is used. In the first scenario, the calculation showed a required downlink data rate of 21.52 Mbps and the second scenario required a down link data-rate of 7

Mbps per satellite. 2.2.3.Downlink data-rate requirement for nano-satellite providing data

services

For this case, VDE-Sat is considered. VDE stands for VHF Data Exchange Satellite. It is a technological concept developed by IALA e-NAV8 committee and now widely discussed at ITU, IMO and other organizations. VDES was originally developed to address emerging indications of overload of VHF Data Link (VDL) of AIS and simultaneously enables a seamless data exchange for the maritime community. This is an on-going project at ISISpace BV, which is in a conceptual stage. The Figure 8 gives a pictorial representation of the mission concept.

Figure 8. Pictorial representation of VDES [7].

Some of the mission requirements are already formulated, the mission needs to have a high-speed downlink at VHF (161MHz), the available bandwidth is 50 kHz and a downlink symbol rate of 40 ksps. Based on these requirements, link budget was calculated for various

8 International Association of Marine Aids to Navigation and Lighthouse Authorities

http://www.iala-aism.org/committees/enav.html

37


possible modulation schemes and finally 16-APSK modulation was chosen. The link budget analysis for this case is explained in the next section.

38


2.3. Link budget analysis

For the cases discussed in the previous section, a link budget was calculated to check if the assumed system design meets the requirements. The calculations were done by creating a MS excel template with the relevant equations and the possibility to vary the various system parameters (transmit power, transmit antenna gain, receiver antenna gain, etc.). In the case of spacecraft house-keeping data, the required downlink data-rate was calculated to be ~37 kbps. The link budget was calculated with realistic numbers from a commercially available UHF transmitter and the antenna gain was considered to be 0 dBi. With this, the link margin was calculated for different elevation angles and shown in Figure 10.

Figure 9. Link budget calculation of spacecraft housekeeping data downlink.

Figure 10. Link margin for different elevation angles.

39


The modulation scheme chosen in this case is QPSK/OQPSK, and the occupied bandwidth of the transmitted signal is 23 kHz. In the case of an imaging payload discussed in the second case study, after various iterations, an X-band downlink was chosen to downlink at 7 Mbps. The calculations and plot of the link margin for various elevation angles are shown in Figure 11 and Figure 12.

Thus, it can be seen that it was possible to close the link for a downlink data-rate of 7 Mbps, a transmit power of 2 Watts and an EIRP of 35.5 dBm. In the case of VDE-Sat, again various options for modulation scheme and transmit power were tried and it was possible to close the link with 16-APSK modulation scheme and a transmit power of 6 Watts. The link calculations and plot are shown in Figure 13 and Figure 14.

Figure 11. Link budget for an optical payload data downlink.

Figure 12. Link margin vs elevation angles for optical payload downlink.

40


2.4. Summary of the case study and conclusion

A summary of the case study is shown in Table 7, it can be seen that there is a direct relation between the required downlink data-rate and the type of service the mission provides. It is not possible to implement these high data-rates in the amateur bands where the available bandwidth is limited. Thus, there is a need to move to the higher frequency bands and also use spectrally efficient modulation schemes. In general, missions with remote sensing payload needs the fastest downlink. In the above case a 7 Mbps downlink was required to downlink all the generated payload data. The

Figure 13. Link budget for VED-Sat downlink.

Figure 14. Link margin vs elevation for VDE-Sat.

41


frequency band for satellite house-keeping data downlink was chosen to be UHF and it can be seen that it is possible to implement this relatively high-speed down link of 36 kbps within the available bandwidth of 25 kHz.

Table 7. Summary of mission case study.

Spacecraft housekeeping

data

MISC-1 payload downlink

VDE-sat downlink.

Frequency Band UHF 8.4 GHz [X-band] 161.9 MHz [VHF]

Available

bandwidth

25 kHz 10 MHz 50kHz

Occupied

bandwidth

23.3 kHz 6.3 MHz 50 kHz

Downlink data rate 36 kbps 7 Mbps 40 ksps (160 kbps)

Modulation scheme QPSK/OQPSK 16-APSK 16-APSK

Transmit power 500 mW 3 W 6 W

PAPR [dB]9 3.4 4.5 4.5 The table also shows the PAPR of the spectral efficient modulation schemes used. The main disadvantage in using a linear PA in the transmitter to amplify such non-constant envelope signal is that, the efficiency drops drastically. Some of the transmitter components and parameters that depend upon the efficiency and power consumption of the PA are:

- Battery life, - Solar-cell and battery capacity, - Heat-sink size and weight, - Need for additional coolers.

Thus, the main challenge in implementing these modulation schemes for nano-satellite missions is to implement an efficient amplifying architecture and at the same time not compromise on the linearity. In order to achieve this, different efficiency and linearity enhancement techniques need to be investigated and a suitable architecture needs to be chosen for nano-satellite application.

9 Peak to Average Power Ratio calculated including a SRRC filter with a roll-off of 0.5

43


3 3. High efficiency transmitter

architectures ome of the main constraints that needs to be considered while designing a subsystem for nano-satellite mission are: power consumption, efficiency, low complexity, efficient thermal design and sustain harsh space environment such as temperature variations and radiation effects. Transmitter is one of the most power hungry

subsystems of a nano-satellite, based on the mission requirements the transmitter of a nano-satellite can consume up to 30% of the total generated power. Thus, it is very important to investigate a transmitter architecture that can efficiently transmit spectral efficient modulation schemes that were discussed in the previous section, without compromising on the overall performance. The main challenge involved in designing an efficient architecture is: the spectral efficient modulation schemes have a non-constant envelope profile as a result of modulating the message signal into both phase and amplitude of a carrier, the amplitude variation in the modulated carrier is characterized by the Peak to Average Power Ratio (PAPR) of the signal. The PAPR is the ratio of the peak power of the envelope to the average power of the signal envelope. This is one of the critical parameters that influence the trade-off between linearity and efficiency of a Power Amplifier in a transmitter. This chapter gives an overview of the challenges posed by spectral efficient modulation schemes on the performance of the transmitter, followed by different efficiency and linearity enhancement techniques from the literature, a trade-off between the different architectures to determine a suitable one for nano-satellites and a conclusion of this study.

3.1. Overview

The basic functionality of a transmitter is to collect the telemetry/payload data from the On-board computer, put the data into packets, modulate the information in the packet, amplify

S

Figure 15. Block-diagram of a transmitter with digital modulation.

44


the modulated signal to appropriate levels and pass it on to the antenna which radiates the modulated RF signal. A generic block-diagram representation of a transmitter is shown in Figure 15. The baseband board is responsible for interacting with the on-board computer and collect the data to be transmitted (telemetry/payload), implement the data link layer functions, perform signal processing on the baseband digital data (Square Root Raised Cosine filtering) and generates the stream of I (In-phase) and Q (Quadrature) data as an input to the modulator. The IQ modulator, modulates the baseband IQ signal onto the carrier frequency and the output power of the modulator is generally not enough to directly be transmitted and needs amplification. Some of the IQ modulators currently available in the market produce an output power between 0 dBm (1mW) to 10 dBm (10mW). Thus, the amplifier performs the operation of converting the DC power (supply) into a significant amount of RF/microwave output power that replicates the modulated signal that is inputted to the PA. The key design parameters of a PA are efficiency and linearity. With the conventional class of operation such as Class-A, B, AB, C, D. E and F, a trade-off has to be done between linearity and efficiency. There is no one single PA or transmitter technique that suits all applications. A relation between efficiency and linearity for different Classes of operation is shown in Figure 16.

When a PA is used to amplify constant envelope signals such as CW (Continuous Wave), FM, classical FSK or GMSK, linearity of the PA is not critical and can be compromised for efficiency. A Class C, D, E or F would be a logical choice for a constant envelope signal. On the other hand, linearity of a PA becomes critical when the signal contains both amplitude and phase modulation such as QAM or APSK. In this thesis the scope of modulation schemes that will be considered to characterize the performance of the PA are: 16 and 32-APSK, which will be compared with 16, 32 and 64-QAM modulation schemes. The PAPR of the QAM scheme is more than the APSK modulation scheme. The PAPR contributed due to variation in the amplitude of the symbols is shown in Table 8.

Figure 16. Linearity vs Efficiency trade-off for conventional PA classes.

45


Table 8. PAPR contribution due to the envelope profile of various modulation schemes .

In addition to the PAPR contributed by the modulation scheme, additional amplitude variation is caused due to passing the digital data through an SRRC filter. In order to minimize inter-symbol interference, the digital data needs to be passed through a SRRC filter. The PAPR depends on the roll-off factor of the filter. The following table (Table 9) shows the additional PAPR contributed due to the SRRC filter for different roll-off factors [8].

Table 9. PAPR contribution due to the SRRC filter

Roll-off 0.1 0.2 0.3 0.4 0.5

PAPR 7.5dB 5.8dB 4.6dB 3.7dB 3.4dB

Thus the PAPR seen by the PA is the sum of the above two contributors. For example, if 32-APSK modulation is implemented with a SRRC filter having a roll-off 0.5, then the combined PAPR seen by the amplifier is 4.8 dB. Peak efficiency of a PA is obtained when driven at saturation but this is also a non-linear region. Thus, alternate architecture to amplify efficiently without compromising on the linearity is studied in this chapter. Some of the techniques analyzed in this thesis are:

3.1.1. Characterization of linearity and efficiency

The non-linearity in a PA is caused mainly due to gain compression and saturation. The effects of non-linearity can be characterized using various methods: carrier-to-intermodulation (C/I)

Modulation M-PSK 16-APSK 16-QAM 32-APSK 32-QAM

PAPR 0dB 1.1dB 2.6 dB 1.4 dB 3.9 dB

Figure 17. Efficiency and linearity enhancement techniques.

46


ratio is used to compare the amplitude of the desired output carrier to the intermodulation distortion (IMD) products, adjacent channel power ratio (ACPR) which is used to characterize the quality of the spectrum, the power ratio in dBc is measured between the spectral components within the allotted bandwidth and adjacent channel components. Another method to characterize the linearity is by measuring EVM (Error Vector Magnitude), EVM gives the vector distance between the received symbol and expected symbol for a given modulation scheme in the constellation diagram. Efficiency is one other important design parameter, from the literature there are various definitions for efficiency of a PA. Drain efficiency: It is the ratio of the output power from the PA (in Watts) to the supplied DC power (in Watts). It is given by:

outd

DC

P

P (1)

Power added efficiency (PAE): The PAE is defined as the ratio of the RF power that is amplified (difference between the RF output power and RF input power) to the DC power consumed. It is given by:

( )out inPAE

DC

P P

P

(2)

The PAE is a more widely used metric as it takes into account the gain of the PA. When a non-constant envelope signal is amplified by a PA, a more useful metric to measure is the average efficiency [9], it is mathematically expressed as

oAVGAVG

dcAVG

P

P (3)

PoAVG and PdcAVG are a function of the Probability Density Function (PDF) of the input non-constant envelope signal. The PDF gives the relative amount of time an envelope spends at various amplitude levels. Figure 18 gives the PDF of a 16-APSK signal when passed through a SRRC filter that has a roll-off of 0.5.

Figure 18. PDF of 16-APSK modulation scheme, implemented using an SRRC filter with a roll-off of 0.5(4000 sps)

47


Although this matric is straight forward to measure for the conventional PA classes (Classes A-F), the above equation cannot be directly used in all the efficiency and linearity enhancement techniques that will be discussed later in this chapter. In the case of efficiency and linearity enhancement techniques such LINC, EER and ET the message signal that is fed to the PA is modified so that the PA is driven at its peak efficiency. In such cases PAE is a more preferred metric to gauge the performance of the architecture.

3.2. Doherty architecture

The Doherty architecture (shown in Figure 19 ) was first proposed by W.H. Doherty in 1936 [10]. This architecture makes use of active load-pull technique where the load seen by one PA can dynamically be varied by applying current from another PA which is coherent in phase.

3.2.1.Working Principle

Figure 19. Doherty PA architecture.

Figure 20. Schematic representation of Doherty.

48


This architecture comprises of two10 signal paths, amplified by two non-identical PAs called Carrier PA and Peaking PA (sometimes also called Auxiliary PA). At low input levels the peaking PA is in cut-off (consuming no current) and only the carrier PA is operational. Once the input level reaches a certain threshold, the peaking PA turns ON and the output power is contributed by both the PAs. The transfer curves of the two PAs is shown in Figure 21 [11]. The carrier PA operates linearly till the point output power reaches Pmax/4 as there is no influence from the peaking PA (not yet turned ON), at Pmax/4 (which is 6 dB below the peak envelope power), Peaking PA turns ON and current flows from both the PAs to a common load

leading to load modulation. From this point onwards the transfer curve of both the PAs are non-linear. The process of active load pulling can be better explained using the schematic representation of the architecture shown in Figure 20. Both the PAs are considered to be current sources. When the Peaking PA is OFF, I2 = 0 and only I1T flows through the load resistor RL. The voltage drop across the resistor is a linear function of the current flowing through the resistor RL. But when the Peaking PA turns ON, both I1L and I2 flows through the load resistor and the impedance seen by the individual PAs is given by:

10 It is possible to have N paths, but the architecture was introduced with two paths.

Figure 21. Input power vs Output power in linear scale [11]

49


21

1

2

11

2

12

1

12

2

1

1

1

T L

T

T

T

T

TL

TL

IZ R

I

ZZ

Z

ZZ

I ZR

V

IZ R

I

(4)

It can be seen from the above equation that the impedance seen by each of the PA is a function of the current flowing through the other PA. 3.2.2.Advantages and dis-advantages of Doherty architecture

Some of the advantages of Doherty architecture are: - It is possible to achieve a good efficiency enhancement in the power back-off region. - Suitable to amplify signals with a PAPR as large as 10dB. - The architecture is relatively easy to implement compared to EER and Envelope

Tracking architectures. - The performance in the power back-off region can be significantly improved by using

more stages (N-way Doherty) [12]. Dis-advantages of Doherty are:

- Gain degradation: Due to the low bias voltage, Peaking PA contributes lower output compared to Carrier PA.

- Peaking PA has to switch ON at exactly Pmax/4. - Both PAs have to follow the nonlinearity of Figure 21 and generally pre-distortion

circuit is necessary. - Higher intermodulation distortion: It is caused again due to low biasing of the Peaking

PA. - Narrow bandwidth: The quarter wave lines used for impedance transformation reduces

the bandwidth of operation.

3.3. Kahn/EER (Envelope Elimination and Restoration)

3.3.1.Working principle of Kahn/EER architecture

This technique of amplification was first proposed by Kahn in 1952. In a non-constant envelope signal, message is modulated onto both amplitude and phase of the carrier signal. In the case of Kahn/EER method, the amplitude of the message is first eliminated from the modulated signal, leaving behind a constant envelope phase modulated signal and the amplitude envelope profile. Now, the constant envelope signal can be amplified using a high-efficiency, non-linear PA such as class C, D and the envelope profile is separately amplified and fed in as the supply

50


voltage of the high efficiency PA. A block diagram representation of this architecture is shown in [figure]. Thus, it is possible to achieve high efficiency and linearity regardless of the PAPR of the input signal. From the literature, it can be seen that very high efficiencies can be achieved but the design is fairly complex to implement.

3.3.2.Advantages and dis-advantages of Kahn architecture

Some of the advantages of the architecture are: - Ideally can be operated at the peak efficiency of the final PA (depends on the

operating Class chosen for the final stage). - Provides excellent linearity as the performance is not dependent on the linearity of the

amplifying transistor. Some of the dis-advantages of this architecture are:

- Circuit complexity: Implementation of the envelope restoration circuitry is complex, as perfect synchronization has to be achieved between the two signals.

- Needs pre-distortion at higher frequency. - The switching frequency of the class-S modulator should be at least 6 times the RF

bandwidth.

3.4. Envelope Tracking (ET) architecture

3.4.1.Working principle of ET

ET and EER technique use the concept of supply modulation, where EER uses a combination of non-linear PA and an envelope re-modulation circuit but in the case of ET, a combination of linear PA and a supply modulation circuit which tracks the envelope profile of the input signal . The envelope profile is converted to discrete DC levels using the power conditioner and fed as the drain voltage to the final PA. The supply voltage is varied dynamically with sufficient

Figure 22. Block diagram representation of a modern implementation of the Kahn architecture.

51


headroom to allow the RF PA to operate in a linear mode. The block diagram representation of an ET architecture is shown in Figure 23

3.4.2.Advantages and disadvantages of ET architecture

The advantages of ET architecture are as follows:

- The implementation is slightly less complex when compared to EER. - The efficiency of linear amplifiers such as Class-A or AB can be enhanced using this

technique. - Constant efficiency can be maintained in the power back-off region.

The main disadvantages of this architecture are: - The peak efficiency achieved cannot exceed that of Class-AB. - Circuit overhead is high. - Synchronization between the carrier containing phase information and the time varying

supply is the main challenge in this technique. - The PAE is poor compared to other efficiency enhancement techniques due to the

additional circuitry.

3.5. Switched Capacitor Digital Power Amplifiers SCDPA

3.5.1.Working principle of SCDPAs

Instead of making use of the linear trans-conductance property of a CMOS transistor, SCDPA makes use of its switching property. The block diagram of SCDPA is shown in Figure 24. Deeply scaled CMOS transistors are poor trans-conductors but very good switches. SCDPAs exploit this property. SCDPAs incorporates the functionality of a Digital-to-analog converter and a PA into the same circuitry. Each of the CMOS transistors are represented by switches (C0, C1, C2…. C7) and they are either at toggle (between Vdd and ground) or only ground potential. This is decided by the control signal given to the gate from the micro-controller/DSP.

Figure 23. Block diagram of ET architecture.[11]

52


The output voltage depends on the ratio of the switches that are ON (toggle state) to the total number of switches. For example, C4, C5, C6 and C7 are at ground potential (representing OFF) C0, C1, C2 and C3 are in toggle mode (ON state) then Vout will be 0.5Vdd. Similarly, when all the switches are in toggle mode, maximum output power can be achieved. More details about this architecture can be found in [13].

3.5.2.Advantages and dis-advantages of SCDPAs

Some of the advantages of this architecture are:

- Low power consumption. - High signal bandwidth. - Good performance in the power back-off region. (flat efficiency uo-to 13dB PAPR

reported [14]) Some of the disadvantages of this architecture are:

- Not highly linear: AM-PM distortions are high. - Parasitic capacitance in the current technology still limit its performance. - Low PAE (45% reported[14]) - Low output power.

Figure 24. Block diagram representation of SCDPA.

53


3.6. LINC (Linear amplification using non-linear components) architecture

3.6.1. Working principle

LINC amplifiers, also known as out-phasing amplifiers were first developed by Henri Chireix in 1935 [15] followed by L. F. Gaudernack’s work in 1938 [16]. These systems were proposed to improve both efficiency and linearity of an AM- broadcast transmitter. The concept slowly gained popularity over the years and in 1974 D. C. Cox [17] introduced the term LINC. As the name suggests, linear amplification was obtained using non-linear components. The working principle of this architecture can be explained using the block diagram shown in Figure 25. An amplitude modulated (AM) and phase modulated (PM) signal (non-constant

envelope signal) such as 16-APSK is split using a signal processing unit into two PM (constant envelope) signals S1(t) and S2(t). The signals S1(t) and S2(t) are then amplified independently, denoted by G.S1(t) and G.S2(t). The amplified signals are then combined using a power combiner to replicate the amplified AM and PM signal (in this case 16-APSK). This method has gained popularity due to the recent developments in signal processing units. The main advantage in using this system is that, as the signals being amplified by the PA blocks (amplifier 1&2) are constant envelope signals, thus it is possible to use high efficiency PAs and not worry about the non-linearity caused by the PAs. The operation can be explained using the following mathematical expressions: The AM and PM modulated signal can be represented using the expression:

j( ( ))( ) r(t) t t

ins t e (5) Where r(t) is the AM and can be represented as r(t) = rmax cos(θ(t)).

( ) ( )maxrr(t)2

j t j te e (6)

Thus, the input non-constant envelope signal can be represented as a sum of two constant envelope signals:

Figure 25. LINC block diagram.

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( ( ) ( )) ( ( ) ( ))maxr( )2

( ) 1( ) 2( )

j t t t j t t t

in

in

s t e e

s t S t S t

(7)

Where,

( ( ) ( ))max

( ( ) ( ))max

r1( ) ,2

r2( )2

j t t t

j t t t

S t e

S t e

(8)

The above mathematical expressions can be described using 16-APSK constellation diagram

as shown in Figure 26. In this case, the symbol 1100 is a non-constant envelope signal with its amplitude 2.84 times lesser than the peak amplitude. This symbol is decomposed into constant envelope signals S1(t) and S2(t) with appropriate out phasing angle θ(t) and constant amplitude rmax/2.

max

( )( ) arccos r tt

r

(9)

In the above case, the ratio between instantaneous amplitude and peak amplitude is (1/2.84), thus using the above relation, θ(t) = 69.38 deg. In order to represent a constellation point on the outermost ring (1000, 1001, …) the out-phasing angle θ(t) = 0 deg. 3.6.2. Advantages and dis-advantages of LINC architecture

Some of the advantages of LINC architecture are as follows:

- AM/AM and AM/PM distortions caused by individual PA blocks does not affect the overall performance of the architecture.

Figure 26. 16-APSK decomposition into constant envelope signal.

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- Identical PAs are used in the two signal paths, leading to symmetrical designing of the two paths. Thus, the matching network and the biasing network designed for one of the paths can simply be replicated for the other.

- Most of the complexity is in splitting the signal component and synchronizing them, which is taken care by the signal processing unit, the RF design complexity of the PAs is much lesser compared to the other efficiency/linearity enhancement techniques.

Some of the disadvantages of the LINC architecture are as follows: - Although using isolating power combiners gives the best performance in terms of

linearity, the overall efficiency of the system drops considerably in an isolating power combiner such as WPC. Half of the power is dissipated in the isolation resistor, bringing down the overall efficiency of the system.

- In the case of non-isolating power combiners (Chireix combiner), the linearity is compromised for efficiency at large out-phasing angles as a result of load modulation.

3.7. Trade-off analysis

All of the efficiency and linearity enhancement techniques discussed so far have both advantages and dis-advantages, there is no one architecture that suits all applications. Thus, a trade-off needs to be performed to determine the most suitable architecture for nano-satellite application. The parameters and its weights that are considered for a trade-off can vary based on the application. The parameters considered for this trade-off are: Complexity: This parameter is given the maximum weight of 5. While designing a system for space application, it is very important to that the system is robust and less complex. Circuit over-head: In some cases the over-head in accurately designing the biasing circuitry or synchronization circuitry adds on to the complexity of the design. Thus, an architecture with minimal over-head is preferred over the rest. This parameter is given a weight of 4. Form-factor and performance in the power back-off: The next highest priority is given to form-factor and the performance in power back-off with a weight of 3. As mass and volume is a major constraint in nano-satellite design, it is important to choose an architecture that is less voluminous. The form-factor of all of the architectures discussed above is considerably large compared to the conventional operation classes such as Class-A, B, C, etc. Thus, this parameter is given a weight of 3. All of the architectures discussed are capable of performing equally well in the power back-off levels associated with 16 and 32 APSK modulation schemes, thus the weight is not high. Efficiency and Linearity: Although the discussion so far has been mainly towards choosing a highly efficient and linear architecture, based on the literature study it was seen that there was not much to differentiate between the architectures. Thus, when it comes to trade-off analysis, these parameters are not given a very high weight.

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Based on the weights assigned to each of the parameter, the highest an architecture can score is 95. This following table gives the trade-off:

It can be seen from the above table that LINC architecture proves to be a better choice for nano-satellites compared to the other efficiency and linearity enhancement architectures.

3.8. Conclusion

An analysis of different efficiency and linearity enhancement architectures were performed and it could be seen that there was no one architecture that suited all applications. All the architectures had a certain advantage over the other, thus a trade-off was performed on the different architectures to see which one would suit nano-satellite application the best. The parameters considered for the trade-off were: complexity/cost, design over-head, form-factor, performance, linearity and efficiency. Based on the trade-off, LINC architecture proved to be the most suitable architecture for nano-satellite application due to its reduced complexity in the RF design, lower over-head and smaller form factor. Thus, it was decided to investigate the LINC PA in more detail to understand some of the implementation challenges.

Table 10. Performance trade-off

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4 4. LINC Architecture

s explained in the previous section, the principle of operation of the LINC architecture is by splitting the non-constant envelope signal (AM and PM) into two constant envelope signals (PM signal) and amplify them individually. By doing this, it is possible to drive the PAs at its peak efficiency. The efficiency of the

complete system is dependent on: efficiency of the individual PA blocks and the combiner efficiency. On the other hand, as the PAs are driven by constant envelope signals, the linearity of the system is mainly dependent on the linearity of the combiner. There are various choices one can make while selecting an appropriate operating class for a PA and a power combining technique. The subsequent section will give an overview of how an appropriate operating class and power combining technique was decided.

4.1. Class of operation

The operation classes of PAs in class AB, B, C, D, E, F, F-1 and J are possible to implement on LINC architecture. Based on the literature, it was seen that LINC/Chireix amplifiers were implemented using Class-B [18], C [19], AB and F [20]. The analysis performed in [21] was taken as a reference to choose the best mode of operation. In the analysis, the operating modes considered were Class-AB, B, C, CMCD (current mode Class-D), VMCD (Voltage mode Class-D), E, F, F-1and J. Simulations were performed on these Classes using CREE GaN HEMT (CGH27015), it was designed for an output power of 40 dBm (10 Watts) at 900 MHz for 16-QAM modulation. Details about the simulation models can be found in [21]. The simulations were done for a symbol rate of 3.84 Msps and roll-off of the filter equal to 0.35. The drain efficiencies of the different classes are as follows:

Table 11. Drain Efficiency (DE) for different Classes [21]

Class AB B C CMCD VMCD E F F-1 J DE (%)

60 62 64 83 71 88 71 69 75

The PAE for these classes were as follows:

A

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Table 12. Simulation results from [21 ] for PAE of the different amplifier classes.

Class AB B C CMCD VMCD E F F-1 J PAE (%)

56 55 53 79 69 84 67 65 69

The publication gives a detailed analysis of efficiencies simulated for two different stub compensation angles: 10 deg and 30 deg. The simulated output spectrum which helps is shown in Figure 27.

Thus, based on the analysis and results from this publication, it was seen that Class-F, F-1 and CMCD give the best performance in terms of both efficiency and linearity, but CMCD requires two transistors for their implementation whereas Class-F or F-1 require a single transistor per PA. For this work, Class-F implementation was chosen and simulations were performed to verify the results from literature. 4.1.1. Class-F implementation

Class-F PAs use the technique of shaping the drain/collector waveform to achieve a better efficiency. Shaping of the waveform is done by appropriate harmonic termination. In the case of class-F mode of operation, the drain is presented with a short-circuit termination at the even harmonics of drain voltage/current, open-circuit termination at odd harmonics and the desired load based on the required output power at the fundamental frequency. This helps in flattening of the voltage waveform, allowing the majority of the drain current to flow when the drain

Figure 27. Output spectrum of 16-QAM for different classes of operation. [21]

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voltage is low, resulting in reduced power dissipation. By lowering the power dissipation, the efficiency is improved. The increase in efficiency is directly related to the number of harmonics handled [22]. Figure 28 shows a functional representation of Class-F mode of operation. By implementing such harmonic tuning at the output, it is possible to shape the drain voltage into a square wave, which helps in reducing the overlap between the current and voltage waveform. An ideal case drain voltage and current waveform is shown in Figure 29. The expression for drain voltage and current can be obtained as explained in [22], but in practical implementations it is not possible to handle all the harmonic components. From [23] it is seen that, by handling only the first four harmonics, it was possible to attain a PAE larger than 80%.

Based on the equations derived in [22], γv and γi were used to fine tune the load resistance corresponding to the desired output power.

Figure 28. Functional representation of Class-F PA

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The Figure 29 shows a schematic of the output harmonic matching circuit simulated on ADS using ideal components and [figure] shows the output impedance seen by the PA at different harmonic frequencies. The circuit is designed for a fundamental frequency at 900 MHz.

The implementation of the Class-F amplifier was done using a GaN HEMT from CREE CGH27015F.

Figure 29. Class-F harmonic matching circuit.

Figure 30. Load impedance by the drain of the transistor at different harmonic frequencies.

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The IV characteristics the used HEMT is as shown in [figure]. Using the large signal model of CGH27015F from CREE, the simulations were performed on ADS.

The harmonic impedance matching circuit shown at the output was implemented at the output of the HEMT and the following figure [figure] shows the drain current and voltage waveforms.

The final implementation of the class-F PA was done based on the [20]. More details about the final design integrated with the Chireix combiner is elaborated in subsequent sections.

Figure 31. IV characteristics of CGH27015F.

Figure 32. Drain voltage and current waveform for a Class-F PA implemented using CGH27015F GaN HEMT.

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4.2. Different power combining techniques

Once the out-of-phase signals are independently amplified using Class-F PAs, the signals are then combined to produce the amplified non-constant envelope signal. The combiner network acts as an adder, enabling the net output amplitude to be controlled via the relative phase of the two non-constant envelope signals. There are various methods of power combining from the literature. The various methods of power combining for LINC system are shown in Figure 33

The method of combining can broadly be classified into two types: using matched combiners and using un-matched combiners. A matched combiner would provide high isolation between the two input ports and provides a constant load impedance to each of the PA for all out-phasing angles. Examples of matched combines are Wilkinson Power Combiner (WPC), Hybrid couplers, rat-race, etc. Although the performance of such combiners are very good in-terms of linearity, the efficiency of combining is traded for the isolation. Almost half the power is dissipated in the isolation resistor. Thus, these techniques were not very popular for the final implementation of LINC, they were mainly used for calibration purpose. In recent time, use of such combiners have re-gained popularity as methods such as RF-DC conversion can be used to recycle the power dissipated in the isolation resistor [24]. Although this method shows promising performance, the circuit over head is quite large and best suited for more than four-way power combining. Due to these reason, this method was not incorporated as the power combining network but ADS simulations were performed to quantify the performance of different matched power combining techniques.

Figure 33. RF power combining techniques for LINC.

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Figure 34 shows the two out-of phase input signals V1(t) & V2(t), combined signal (vload) and efficiency as a function of out-phasing angle.

Although WPC can provide a good isolation between the two input ports, sometime it is not good enough to completely isolate the two ports. Some of the commercially available WPC s such as ZAPD-30-S+ (from Mini-circuits) provide an isolation between 12 and 15 dB. The can

Figure 34. WPC as a power combining network for LINC

Figure 35. Plot of the two out-of-phase signals, combiner voltage waveform and efficiency of WPC as a function of outphasing angle.

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lead to power getting coupled into the other port and seen at the output of the PA, this leads to degradation in the performance of the PA. This was noticed during the measurement done latter. In-order to over-come this, a quadrature coupler stage can be implemented before a WPC and simulation results show a significant improvement in the isolation. Figure 36 shows the schematic of this setup,

Figure 37 shows the coupling between the ports.

The simulation was done for a CPW (co-planar waveguide) on a FR-4 substrate. The substrate parameters can be seen in Figure 38. The efficiency and voltage waveforms can be seen in Figure 39

Figure 36. Improvement of isolation by introducing a quadrature coupler between the PA and WPC.

Figure 37. Coupling between the two ports.

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Apart from these methods where it is possible to attain high isolation at the cost of efficiency at larger out-phasing angles, there are methods of power combining using unmatched networks. Chireix combiners is one such example of a lossless combiners with low isolation. A Chireix combiner mainly comprises of two simple λ/4 lines with one end having a compensation element (capacitive/inductive) and the other end connected to the load. The concept of using Chireix combiner in LINC can be understood using the schematic representation of the combiner shown in Figure 39. [25].

The output of the PAs are represented by voltage sources V1 and V2. These voltage sources can be expressed in phasor form as follows:

( )1 .(cos sin )

j

o

o

V V e

V j

(10)

Figure 38. Efficiency and voltage waveforms when a quadrature coupler is introduced between the PA and combiner.

Figure 39. Schematic representation of Chireix combiner.

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( )2 .(cos sin )

j

o

o

V V e

V j

(11)

Here, θ is the out-phasing angle that can vary between 0 and 90 deg. Thus, the voltage across the resistor is:

| 1 2 | 2 . sin( )L oV V V V j (12) Based on the derivation from [25], the impedance seen by the two amplifiers can be expressed as:

1 1 cot2

LRZ j (13)

2 1 cot2

LRZ j (14)

The corresponding admittance are:

2

12.sin sin 2

L L

Y jR R

(15)

2

22.sin sin 2

L L

Y jR R

(16)

It can be seen that the susceptance seen by both the PAs are different and is a function of the out-phasing angle. The susceptance seen by the PA is depicted in Figure 40.

As these susceptances are a function of the out-phasing angle, compensation can be provided based on the PDF of the modulated signal. The relation between out-phasing angle and amplitude is described in equation (9). The value of the inductor and capacitors can be calculated using the imaginary part of the conjugate of admittance from equation (15) and (16).

Figure 40. Chireix combiner as seen at the inputs when fed out-of-phase signals.

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Based on the above equations, a chireix combiner was designed and simulated on ADS, the substrate used was FR4 and the transmission lines used were CPW lines. LineCalc tool in ADS was used to calculate the dimensions of the CPW corresponding to the electrical length and characteristic impedance of the line. Figure 41 shows the schematic of the Chireix combiner.

The above design is implemented for a compensation angle of 10 degrees. This can be seen from the S-parameter simulation results shown in Figure 42. The initial phase difference

without compensation stubs between the input port-1/2 to output port-3 was measured to be

Figure 41. Chireix combiner design on FR4 using CPW.

Figure 42. S-parameter simulation of the Chireix combiner with compensation stubs placed at 10 deg.

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118.2 deg. It can be seen that by adding 10 degree compensation stub, Phase(S(2,3)) has reduced by 10 degrees and the Phase(S(1,3)) has increased by 10 degrees. A comparison was done between WPC and Chireix combiner to quantify the efficiency of the two combiners as a function of out-phasing angle, this plot is shown in Figure 43.

It can be seen from the plot in figure that the efficiency is improved for larger out-phasing angles when a Chireix combiner is used. The main drawback with Chierix combiner while compared to WPC is that the linearity is poor at larger out-phasing angles due to the fact that Chireix combiner provides poor isolation between the two input ports. This can be seen in Figure 44 which shows the poor isolation exhibited by the Chierix combiner.

Figure 43. Efficiency vs out-phasing angle comparison between Chireix combiner and WPC.

Figure 44. Comparison of coupling between Chireix combiner and WPC.

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One other power combining technique that has surfaced in recent time is, spatial power combining. In this technique, the output of the PAs are fed directly to two separate antennas and the radiated power from the individual antennas combine in the air to form the non-constant envelope signal. Using the idea from [26], simulations were performed on a UHF antenna combiner on nano-satellites. A simulation on FEKO was done on crossed dipole for different out-phasing angles to see its effect on the radiation pattern. The radiation pattern when the dipoles are fed with signals having an out-phasing angle of 45 degrees is shown in Figure 45

The radiation pattern of the crossed dipole when fed with zero out-phasing angle is shown in Figure 46.

Figure 45. Radiation pattern of crossed dipole on a 3U cube-sat fed with an out-phasing angle of 45 degrees.

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It can be seen that the beam-width has increased and the peak gain has decreased from 3.4 dBi to 3 dBi.

Thus, it could be seen that the main effect on the radiation pattern due to the change in out-phasing angle is the variation in the beam-width. The gain variation in the bore side was not much. The crossed dipole arrangement provides fair isolation between the two antennas.

Figure 46. Radiation pattern of the crossed dipole when fed with signals with 0 degree outphasing angles.

Figure 47. Radiation pattern of crossed dipole fed with signals with 90 degrees out-phasing angle.

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Although, the simulation results show promising results, further analysis needs to be performed to use it in a LINC architecture, making this an interesting topic for future work. After analyzing various amplifier cells and different power combining techniques, it was decided implement the LINC architecture using Class-F PA and a Chireix combiner. In order to show a proof of concept, a LINC architecture designed by CATENA microelectronics BV was used [27] This board had implemented Class-F PA cells and a Chireix combiner, thus this board was used to validate the above simulation results and propose possible improvement to the design. The further sections will describe the measurements performed on the LINC PA for different modulation schemes.

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4.3. Experimental setup

The experimental setup used to validate the performance of the LINC/Chireix PA architecture plays a very crucial role in determining its linearity. It is very important that the measurement setup does not add additional non-linearity to the measurements. As the output of the final PA is obtained by combining two out-phased signals S1(t) and S2(t), it is critical to synchronize the phase between the two signals in order to minimize the AM-AM distortion that phase miss-match can cause. The measurement setup used by CATENA microelectronics BV made use of two separate signal generators, which were synchronized externally, and the vector spectrum analyzer also needed to be synchronized with the RF generators in order to perform EVM measurements. More details about the measurement setup used by CATENA can be found in [27]. Based on the conclusion from [27], it was found that there was quite some discrepancy between the simulation results and measurements. Thus, it was decided to take a different approach to perform the measurements. The measurement setup used in this work was initially developed at the Mixed-signal Microelectronics (MsM) group at the Technical University of Eindhoven as a part of a FP7 project. The subsequent section will discuss about the measurement setup used and modification done to the original measurement setup to improve its performance. 4.3.1. Single RF source measurement setup

A block diagram representation of the measurement setup used in this work is shown in [ figure: measurement setup], the setup comprises of a National Instruments PXIe-108211 with two FlexRIO FPGAs (Xilinx Virtex 5 SX50t) add-ons with baseband transceiver NI 578112 adapter modules. It comprises of two 100MS/s, 14-bit, differential, DC-coupled inputs (ADCs) and two 16-bit, differential, DC-coupled outputs (DACs) that are optimized for interfacing with baseband to RF up-converters and down-converters. Thus, it makes it easy to synchronize the two constant envelope signals.

11 http://sine.ni.com/nips/cds/view/p/lang/nl/nid/207346 12 http://sine.ni.com/nips/cds/view/p/lang/nl/nid/208378

Figure 48. Complete measurement setup.

http://sine.ni.com/nips/cds/view/p/lang/nl/nid/207346

http://sine.ni.com/nips/cds/view/p/lang/nl/nid/208378

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The PXIe-1082 runs a LabView GUI in the front-end which controls the flow of signal. It executes a MATLAB script in the backend which is used to generate the IQ baseband signal based on the defined modulation scheme then separates the baseband signal of the non-constant envelope signal into two constant envelope signals: S1 and S2 (SCS code included in Appendix). The generated IQ signal is then written onto the FPGA. Each of the FlexRIO (RIO0 and RIO1) has two inputs, two outputs (differential) which are used to convert the digital data into analog waveform and passed it to the IQ mixers. The IQ mixers (MAXIM 2021) modulate the baseband signal onto the carrier frequency fc. The MAXIM 2021 is a differential IQ up/down-converter, thus the same mixer can be used in both transmit path as well as the receive path. The received signal is then down converted to the baseband and the received symbols are compared with the expected symbols to determine the EVM. A flow of the signal is depicted in the Figure 49.

Initially, in-order to validate the setup, the output of the mixers are directly connected to a Wilkinson power combiner (WPC). A block diagram of this setup is shown in the Figure 50. The output of the two mixers S1(t) and S2(t) are combined using the WPC and the desired non-constant envelope signal S(t) is obtained. This signal is then demodulated using the third MAXIM 2021 and the demodulated IQ signal is given as the input to RIO0.

Figure 49. Signal flow diagram.

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A screenshot of the labview interface is shown in Figure 51. The EVM for this case was measured to be 0.045%, showing very negligible non-linearity contributed by the test setup and the mixers.

Next, the non-linearity introduced by the driver PA is measured and if required, calibration is done to compensate for the non-linearity. The driver PAs used for this measurement setup are

Figure 50. Initial setup for calibration.

Figure 51. Screenshot of calibration measurement.

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two identical Mini-circuits ZHL-2-8, the characterization performed on these PAs can be found in the Appendix. The measurement setup is shown in Figure 52.

After the calibration is performed, the LINC/Chireix PCB is connected to the two outputs of mini-circuits driver PAs. The LINC/Chireix PCB was developed by CATENA Microelectronics BV, it has two Class-F PAs implemented using CREE’s CGH27015, gallium nitride (GaN) High Electron Mobility Transistor (HEMT). The output of the Class-F PAs are given to a lossless combiner (Chireix combiner) to combine the amplified constant envelope signals. The measurement setup used to measure the EVM of the LINC/Chireix PCB is shown in Figure 48. The gate of the two transistors are biased with -2.7V, and the drain is supplied with 22V. Each of the transistors consume about 150mA current without any input signal. As the PCB is designed to produce a peak output power of 42 dBm, it needs to be operated with a cooling fan under the PCB. The output of the LINC PCB is given to a coupler with a coupling

Figure 52. Setup to measure the non-linearity caused by mini-circuits PAs

Figure 53. Picture of the measurement setup.

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of 40dB. The in-line power is terminated using a load termination. The coupled signal (S(t) in figure) is used to demodulate the desired non-constant envelope signal using a MAXIM 2021 in the receive path and the demodulated IQ signal is given to the input port of RIO0. A picture of the measurement setup is shown in Figure 53. 4.3.2. Modifications done to the original setup

As stated earlier, the setup was developed as a part of Master thesis, but the setup could not be used directly. The setup was not validated earlier with a LINC architecture, and when it was used with a LINC PA in the complete chain, some of the unforeseen issues had to be tackled. Certain modifications were done to the setup and measurement procedure in-order to make the measurements more reliable and repeatable. Here are some of the modifications:

MATLAB libraries were written for 16 and 32-APSK modulation and demodulation.

The radius ratio used for 16-APSK was R1 = 2.85 and for 32-APSK: R1 = 2.84 and R2 = 5.27. This was chosen based on the coding rate assumed (coding rate: 3/4).

MATLAB Script: The algorithm used in signal components separator (SCS) was implemented based on the [28], this algorithm is valid only if the IQ components of the non-constant envelope signal is normalized. But, it was noticed that for a certain symbols (symbols at the peak power) was exceeding unity due to the SRRC filter. This

condition is shown in Figure 54. The red points represent output from RIO0 (S1) and the blue point represent output from RIO1 (S2). Figure 55 shows the constant envelope signal from RIO0 and RIO1 without any anomalies in it the amplitude. The new code made use of directly manipulating on the phase by representing the constant envelope signal in polar form.

Figure 54. Variation in the constant envelope signal due to the previous SCS algorithm.

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MATLAB code and Labview flow: While using the previous algorithm, the final constellation did not change because, the calibration function would add the desired off-set to finally obtain the required constellation. One other observation from the calibration sequence was: The offset scaling was performed on both amplitude and phase of the constant envelope signal. Thus, the calibration sequence was removed from the code and executed without any calibration. An architectural modification had been proposed in Chapter-6 which can be used for calibration and real-time feedback for distortion correction.

Receiver demodulator: As a future work from the previous work, it was proposed to use a MAXIM 2021 in the receiver instead of using discrete mixers for down conversion. This was implemented during the measurement and observed that there was significant improvement.

Additional input step attenuators: It was observed that the HEMTs in the two signal paths had different threshold voltages (HEMT in S1 had -4V and the one in S2 had -4.5V). In order to compensate for this mismatch, step attenuators needed to be placed at the inputs of the PA. This helped in improving the EVM.

4.4. Catena LINC/Chireix PA architecture

The LINC PA used in the measurements was developed by CATENA Microelectronics BV. A picture of this PA is shown in Figure 56. It makes use of two CREE’s CGH27015, GaN HEMTs to amplify the constant envelope signals S1(t) and S2(t). The individual PA block is implemented using Class-F operation. Thus the drain of each of the PA looks at low impedance (ideally 0 Ohms) at even harmonics and high impedance (ideally ∞ Ohms) at odd harmonics. This is implemented using the harmonic load matching network. More information about Class-F matching can be found in [22].

Figure 55. Constant envelope signals after the modified SCS algorithm.

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The output of the PAs are given to a Chireix combiner which has a inductive short circuited compensation stub in the signal path S1(t) (upper path) and capacitive open circuited stub in signal path S2(t). The drains are connected to +22 Volts and the gates are biased by -2.7 Volts. The PCB was designed for an RF output power of 42 dBm. A schematic diagram of the LINC PCB is shown in Figure 57. In order to understand this design, ADS simulations were performed using this schematic as a reference. The simulation results can be found in the Appendix and recommendations to improve this design is discussed in the future works chapter.

Figure 56. LINC/Chireix PA from CATENA Microelectronics.

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Figure 57. Schematic representation of the LINC/Chireix PCB from CATENA

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5 5. Measurement Results

he setup described in the previous chapter was used for the measurements, in addition MATLAB libraries were written for 16 &32-APSK modulation and demodulation. The coding rate used for APSK modulation scheme in the libraries is (¾) which gives a spectral efficiency (bits/symbol) of 2.9667 for 16-APSK and 3.7033 for 32-APSK

(spectral efficiencies corresponding to different code rates is provided in the appendix.) . This chapter will provide the results of the measurements conducted.

5.1. Modulation schemes tested

The modulation schemes tested were 16-QAM, 16-APSK, 32-QAM, 32-APSK and 64-QAM. Constellation diagrams of these modulation schemes are shown in Figure 58, the PAPR for these modulation schemes are discussed in chapter-3. The roll-off factor used for RRC filter was 0.5 for all the modulation schemes.

T

Figure 58. Constellation diagrams of 16-QAM, 16-APSK, 32-QAM, 32-APSK and 64-QAM.

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The modulation and de-modulation libraries for QAM were already available in MATLAB, but in the case of APSK modulation schemes, individual libraries had to be written because 32-APSK was not directly scalable from 16-APSK. The stub-lengths corresponding to different compensation angles at 900MHz is shown in Table 13.

Table 13. Stub lengths corresponding to different out-phasing angles.

Compensation angle [deg] Inductive stub length (Short circuit stub) [mm]

Capacitive Stub length (Open circuit stub) [mm]

0 -- -- 10 40.2 11.47 20 35.3 19.34 30 32.02 23.68 40 30.42 25.56

The results presented here are for compensation angle of 0 degrees and 20 degrees.

5.2. Performance characterization: EVM

As discussed earlier, one of the methods used to measure the linearity of a PA is by measuring the EVM. In-order to measure the EVM, a portion of the amplified signal is coupled to the receiver chain, demodulated using MAXIM 2021 and the demodulated IQ data is compared with ideal constellation to determine the EVM. 5.2.1.16-QAM modulation scheme

The constellation diagram of 16-QAM when no compensation stubs are used in the LINC/Chireix architecture is shown in Figure 59:

Figure 59. Received constellation diagram of 16-APSK when no compensation stubs (0 degrees compensation).

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The blue points depict the received symbol points, the red points depict the expected symbol points. The measured EVM in this case is 0.42 %. The output spectrum of the amplified 16-QAM is shown in Figure 60.

It can be seen that for 16-QAM, the ACLP is approximately -24 dBc. The constellation diagram when compensation stubs of 20 degrees were added at the input of Chireix combiner is as shown in Figure 61.

Figure 60. Output spectrum of amplified 16-QAM.

Figure 61. 16-QAM constellation diagram with 20 degrees compensation stub.

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When a compensation stub of 20 degrees was added, it was observed that the EVM improved to 0.3%. The output spectrum of 16-QAM is shown in Figure 62.

From the output spectrum, it can be seen that there is a significant improvement in ACLP. The ACLR is now ~34 dBc. 5.2.2.16-APSK modulation scheme

Figure 62. Output spectrum of amplified 16-QAM with stubs.

Figure 63. Constellation diagram of 16-APSK modulation without compensation stubs.

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The constellation diagram of 16-APSK modulation when amplified by LINC/Chireix PA without any compensation stubs is shown in Figure 63. The EVM measured in this case was 0.36 %. It can be seen from the figure that the received symbol points in the outer ring suffers minimal EVM, this is because the out-phasing angles for these points is 0 degrees. The inner, four symbols suffer from maximum EVM due to the large out-phasing angle (~61 degrees). The output spectrum of 16-APSK is shown in Figure 64.

It can be seen that the ACLP of 16-APSK without compensation stubs is slightly better than that of 16-QAM, this can be attributed to the larger PAPR of QAM compared to APSK modulation.

Figure 64. Output spectrum of 16-APSK modulation without compensation stubs.

Figure 65. 16-APSK constellation diagram with a compensation stub of 20 degrees.

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The constellation diagram of 16-APSK modulation with a compensation stub of 20 degrees is shown in [figure]. In this case, the EVM was measured to be 0.273 %. From the constellation diagram, it can be seen that there is a slight EVM increase in the symbols on the outer ring but, the symbols in the inner ring show better EVM compared to the result without compensation stubs. The output power spectrum is shown in Figure 66.

5.2.3.32-QAM modulation scheme

The constellation diagram of the received 32-QAM modulation without compensation stubs is shown in Figure 67.

Figure 66. Output power spectrum of 16-APSK with a compensation stub of 20 degrees.

Figure 67. 32-QAM without compensation stubs.

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The measured EVM in this case is 0.44 %. The output power spectrum is shown in Figure 68.

The constellation diagram of 32-QAM after introducing a compensation stub of 20 degrees is shown in Figure 69.

The measured EVM in this case was 0.36 %. The output power spectrum is shown in Figure 70 The ACLP measured in the case of 32-QAM with compensation was ~ -34 dBc.

Figure 68. Output power spectrum of 32-QAM without compensation.

Figure 69. Constellation diagram of 32-QAM with compensation stub at 20 degrees.

88


5.2.4.32-APSK modulation scheme

In the case of 32-APSK modulation, the best performance was obtained for a compensation stub at 20 degrees. The constellation diagram of the received 32-APSK modulation is shown in Figure 71.

The EVM measured for 32-APSK with 20 degrees compensation angle is 0.32%. The output power spectrum when amplifying 32-APSK modulation is shown in Figure 72.

Figure 70. Output power spectrum of 32-QAM with a compensation at 20 degrees.

Figure 71. Constellation diagram of 32-APSK with 20 degrees compensation stubs.

89


5.2.5.64-QAM modulation scheme

For 64-QAM, again both the compensation angles were tried and the performance of the LINC/Chireix was the best at a compensation angle of 20 degrees. The constellation diagram of the received symbols are shown in Figure 73.

Figure 72. Output power spectrum of 32-APSK modulated signal with a compensation stub at 20 degrees.

Figure 73. Constellation diagram of 64-QAM with 20 degree compensation in the LINC/Chireix amplifier.

90


The EVM measured for this case was 0.36 %. Not all the symbols are visible because the symbol rate used by the transmitter was 4000 sps, this was not enough to sample all the 64 randomly generated symbols at the same instant. In order to validate the result, the measurement was carried out multiple times and the EVM did not vary by much. The output power spectrum of the amplified 64-QAM is shown in Figure 74.

5.2.6.Summary of EVM and ACLP measurement.

A summary of the above discussed measurements is listed in [table].

Table 14. Summary of EVM and ACLP for different modulation schemes.

Modulation scheme

Compensation angle

EVMRMS % ACLP (dBc) Att (S1)

dB Att(S2)

dB

16-QAM 0 deg. 0.42 - 24.42 7 8

20 deg. 0.3 - 34.17 4 8

16-APSK 0 deg. 0.3 - 30.2 1 2 20 deg. 0.273 - 34.8 4 8

32-QAM 0 deg. 0.34 - 27.52 8 8

20 deg. 0.36 - 34.82 3 7

32-APSK 0 deg. 0.58 - 24.42 1 2

20 deg. 0.318 - 29.3 4 8

64-QAM 0 deg. 0.48 - 27.05 7 8

20 deg. 0.38 - 31.25 2 7

Figure 74. Output power spectrum of 64-QAM amplified by LINC/Chireix PA with 20 degrees compensation stubs.

91


5.3. Performance characterization: LINC/ Chireix efficiency

The other key parameter that needs to be characterized is the efficiency of the PA architecture. In this case, PAE from equation (2) is considered to characterize the efficiency. The table provides the current consumed, output power and the PAE for each of the modulation scheme.

Table 15. Characterization of the efficiency on LINC/Chireix PA for different modulation scheme .

Modulation scheme

Compensation angle

PA1 current (mA)

PA2 current (mA)

Output power (dBm)

PAE

PAE

16-QAM 0 deg. 664 388 38.5 30

20 deg. 642 432 41.2 51.5

16-APSK 0 deg. 281 821 41.21 50.37

20 deg. 672 546 41.27 49.99

32-QAM 0 deg. 639 391 38.8 29.06

20 deg. 653 550 41.2 46.03

32-APSK 0 deg. 320 790 41 49.88 20 deg. 630 582 40.5 38.32

64-QAM 0 deg. 652 520 41.5 50.9

20 deg. 666 375 38 23

92


93


6 6. Conclusion and Recommendations

for future work Based on the market survey and future trends in nano-satellite missions, it is evident that the nano-satellite missions are getting more and more complex. Complex payloads have started to fly in nano-satellites missions, this has led to an increase in the downlink data-rate. This was seen in the missions flown in 2013, where the number of remote sensing missions, military and government missions had increased and the their respective downlink data-rate were much higher than the rest. A lot of nano-satellite missions have also started moving to the higher frequency bands due to the limited bandwidth availability in the amateur frequency bands. The analysis on the trends also show that transmitters that are compatible with various ground station standards around the world is popular among nano-satellite developers, as the data collected from the satellite can get distributed. Thus, a sudden increase in the number of missions also means than spectrally more efficient modulation schemes need to be implemented in nano-satellites. During the course of this thesis, it was possible to identify the need for such high-speed transmitters for nano-satellites, establish the design requirements using nano-satellite mission case studies and finally design and characterize the performance of a LINC transmitter for various spectrally efficient modulation schemes proposed in CCSDS standards. The recommendations based on the literature study, simulations and experiments are discussed in this chapter. Section 6.1 covers the recommendations on communication standards based on the literature study done on CCSDS recommendations, section 6.2 covers some of the recommendations for the LINC architecture based on simulations performed in ADS and measurements, section 6.3 provides recommendations for improvement in the measurement setup and finally the chapter concludes with future work.

6.1 Recommendations on communication standards. Thus, recommendations stated by CCSDS are investigated in this thesis. The transmitter design specs are discussed in chapter-2 and it is concluded that spectrally efficient modulation schemes proposed in CCSDS space link protocols over ETSI DVB.S2 such as QPSK, MSK, 16-APSK and 32-APSK are good choices. These standards already excites among the bigger satellites and standard ground stations around the world, thus they can be incorporated in nano-satellites without too much modification.

94


The challenges posed by such a spectrally efficient modulation scheme on the performance of the transmitter PA are investigated. As power is a major constraint in nano-satellites, it is recommended that efficiency and linearity enhancement techniques need to be incorporated. Based on a trade-off analysis done on different efficiency and linearity enhancement techniques, it is recommended that LINC architecture is the more suitable choice for nano-satellites.

6.2 Recommendations for a LINC architecture The recommendations to implement the LINC architecture are mainly focused on designing the PA cell and combiner architecture: 6.2.1.PA cell

Although based on some initial literature study it was decided that class-F or Class-F-1 are good choices for a LINC architecture without signal pre-distortion, it was later found that for frequencies below 3 GHz, Class-F-1 is a better choice compared to Class-F. The main reason Class-F-1 was not a popular choice earlier was, unlike in Class-F mode of operation, where the maximum voltage swing is less than double the normal supply voltage, in Class-F-1 mode of operation, the peak voltage can be larger than double the supply voltage at the expense of device stress [29] relative to the device breakdown voltage. This was the cause of concern while using laterally diffused metal-oxide semiconductor (LDMOS) and gallium-arsenide (GaAs) devices. But with the advances made in wide bandgap semiconductor technology (GaN HEMT), it is possible to handle the high stress levels. Thus it is recommended to implement Class-F-1 for frequencies below 3 Ghz so that better efficiency can be achieved. Based on simulations performed using ADS, it was seen that the harmonic matching circuit designed on the Catena’s LINC board can be improved. The Figure 75 shows the impedance seen by the drain of the HEMT, it can be seen that at even harmonics the drain does not see a short circuit (low impedance). Thus, by improving the harmonic matching circuit it would be possible to attain better efficiencies from the present Class-F design.

Figure 75. Harmonic matching circuit for the Class-F PA on CATENA LINC board.

95


6.2.2.Combiner architecture

It is found that a Chireix combiner can be used to combine the output of the two PAs. The combining efficiency of Chireix proves to be much better than that of WPC. Some of the other power combining methods discussed in this thesis are not mature enough to be recommended for space applications. More work needs to be done to prove the proof-of-concept.

6.3 Recommendations for the experimental setup A recommendation proposed to improve the temperature drift of the measurement setup is to get a real-time feedback and use it as the reference. A block-diagram of the proposed setup is shown in Figure 76. The second input port of RIO is not used at the moment, this can be used to decode the output from an intermediate point in the signal path and be used as a reference. By combining the signals from the outputs of the mini-circuits driver PA and using it as a reference, the non-linearity introduced by the driver can be completely negated. It was observed that when the measurement setup was kept ON for a long duration of time, the driver PA starts to heat-up and the initial calibration is no more valid. Thus, by coupling a small portion of the signal from the output of the driver PA and using it as a reference, it would improve the reliability in characterizing the LINC better even when the setup remains ON for a long time.

Figure 76. Proposed measurement setup.

96


6.4 Future work LINC architecture being a very recent development, there is scope for a lot of research in this area. Base on the observations made during the course of this thesis, the future work that is proposed are as follows:

Improvement in the PA cell: Class-F-1 can be implemented instead of Class-F for designs below 3 GHz.

As recommended in the previous section, the harmonic matching circuit of the Class-F PA can be implemented to verify better efficiency.

Alternate power combining architecture : This work gives an overview view of some of the alternate power combining techniques. Some of the promising techniques that can be further investigated are:

Antenna combiners: The main advantage in using antenna combiners is that the design complexity of the PA can be brought down drastically, the form factor reduces, matching between the PA and antenna becomes less complex and integration of the PAs in an IC becomes simpler.

Hybrid-Chireix combiner: During measurements, it was observed that when a hybrid coupler was placed between the output of a PA and input of Chireix combiner, it is possible to obtain better isolation. A picture of this setup is shown in Figure 77. The performance of this system needs to be studied further.

Implementation of circulator/isolator at the input of Chireix combiner: As circulators/isolators are capable of providing good isolation to the return path, implementation of circulators/isolators at the input of the Chireix combiner can improve the isolation between the two signal paths. More investigation needs to be performed to validate this design.

Implementation of circulator/isolator at the input of Chireix combiner: As circulators/isolators are capable of providing good isolation to the return path, implementation of circulators/isolators at the input of the Chireix combiner can improve the isolation between the two signal paths. More investigation needs to be performed to validate this design.

Figure 77. Hybrid-Chireix combiner.

97


The need for per-distortion circuits needs to be studied. Architecture of a low power signal separator needs to be investigated.

Implementation methodology (analog, digital or mixed signal) focusing towards compatibility with nano-satellite design needs to be looked into.

More precise target needs to be set on the EVM and PAE that needs to be achieved.

99


Nomenclature APSK Amplitude Phase Shift Keying

AM/AM Amplitude Modulation to Amplitude Modulation

AM/PM Amplitude Modulation to Phase Modulation

CCSDS Consultative Committee for Space Data Systems

ECSS European Cooperation for Space Standardization

GSD Ground Sample distance

DVB Digital Video Broadcast

EER Envelope Elimination and Restoration

ET Envelope Tracking

EES Earth Exploration Satellites

SR Space Research.

ITU International Telecommunication Union

EVM Error Vector Magnitude

SO Space Operations

LINC Linear amplification using Non-linear components

OBDH On-Board Data Handling

PAE Power Added Efficiency

QPSK Quadrature Phase Shift Keying

OQPSK Off-set Quadrature Phase Shift Keying

PAPR Peak to Average Power Ratio

ISIS Innovative Solutions in Space B.V

PA Power Amplifier

101


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ess

http

s://d

irect

ory.

eopo

rtal.o

rg/w

eb/e

opor

tal/s

atel

lite-

miss

ions

/s/sr

msa

t

NA

SA E

laN

a-3/

NP

Aub

iesa

t-1Ed

ucat

ion/

Tech

nolo

gy

dem

onstr

atio

nSo

lar c

ell

Mel

exis

TH72

011

UH

FU

HF

800

1200

AFS

KCW

AX

.25

UI

Hal

f Dup

lex

Succ

ess

http

://w

ww

.spac

e.au

burn

.edu

/#!a

ubie

sat-1

/czi

g

DIC

E (2

)Sc

ienc

ePl

asm

a den

sity

mea

sure

men

t and

ele

ctric

fie

ld m

easu

rem

ent

L3 C

adet

UH

FU

HF

1000

3.00

E+06

BPSK

N.A

Prop

rieta

ryH

alf D

uple

xSu

cces

sht

tp://

digi

talc

omm

ons.u

su.e

du/c

gi/v

iew

cont

ent.c

gi?a

rticl

e=10

83&

cont

ext=

smal

lsat

http://www.utias-sfl.net/nanosatellites/CanX1/CanX1System.html

http://dtusat1.dtusat.dtu.dk/group.php?c_gid=7&PHPSESSID=881561aae51736370dac1f18d02d764e

http://www.space.aau.dk/cubesat/

http://www.space.t.utokyo.ac.jp/CubeSat/index-e.html

http://www.space.t.utokyo.ac.jp/CubeSat/index-e.html

http://lss.mes.titech.ac.jp/ssp/cute1.7/cute1.7-1/index_e.html

http://www.mae.cornell.edu/research/groups/campbell/satellite.cfm

http://www.mae.cornell.edu/research/groups/campbell/satellite.cfm

http://www.raumfahrttechnik.tu-berlin.de/menue/publications/small_satellite_database/parameter/en/

http://www.nasa.gov/centers/ames/missions/2007/genesat1.html

http://polysat.calpoly.edu/launched-missions/cp4-cp2-relaunch/

http://www.boeing.com/news/frontiers/archive/2006/october/i_ids02.pdf

http://www.tethers.com/papers/MASTJPC2003Paper.pdf

http://polysat.calpoly.edu/launched-missions/cp3/

http://ulcape.org/

http://www.usergioarboleda.edu.co/proyecto_espacial/docs/english_libertad.pdf

http://www.utias-sfl.net/nanosatellites/CanX2/system.html

http://www.space.aau.dk/aausatii/homepage/?language=en&page=home

http://www.raumfahrt.fh-aachen.de/compass-1/overview.htm

http://www.delfispace.nl/delfi-c3

http://lss.mes.titech.ac.jp/ssp/cute1.7/projectoverview_e.html

http://cubesat.aero.cst.nihon-u.ac.jp/english/seeds_e.html

http://www.utias-sfl.net/nanosatellites/CanX6/

http://space.skyrocket.de/doc_sdat/rubin-8-ais.htm

https://directory.eoportal.org/web/eoportal/satellite-missions/a/aerocube-3

http://polysat.calpoly.edu/launched-missions/cp6/

http://www.nasa-usa.de/centers/wallops/pdf/326337main_Cubesat backgrounder.pdf

http://space.skyrocket.de/doc_sdat/pharmasat-1.htm

http://web.archive.org/web/20120316204856/http:/www.raumfahrttechnik.tu-berlin.de/beesat/v-menue2/beesat-1/

https://directory.eoportal.org/web/eoportal/satellite-missions/u/uwe-2

http://usl.itu.edu.tr/en/

http://swisscube.epfl.ch/

http://leo.sci.kagoshima-u.ac.jp/~n-lab/KSAT-HP/Ksat1_E.html

http://www.miyashita.mmech.waseda.ac.jp/Waseda-Sat2/sub2.html

http://space.skyrocket.de/doc_sdat/negaistar.htm

http://www.spacelab.dti.supsi.ch/tiSat1.html

http://www.teamstudsat.com/

http://www.dk3wn.info/sat/afu/sat_rax.shtml

http://www.astrobio.net/exclusive/4644/o-oreos-nanosatellite-success-in-orbit

http://www.nasa.gov/mission_pages/smallsats/11-148.html

http://www.lanl.gov/science/NSS/issue1_2011/story2c.shtml




http://digitalcommons.usu.edu/cgi/viewcontent.cgi?article=1090&context=smallsat


http://www.smdc.army.mil/FactSheets/SMDC-One.pdf


http://www.iitk.ac.in/me/jugnu/index.htm

https://directory.eoportal.org/web/eoportal/satellite-missions/s/srmsat

http://www.space.auburn.edu/#!aubiesat-1/czig�


user

Typewritten Text

Appendix

user

Typewritten Text

1. The database of nano-satellites launched between 2003 and 2013 is as follows:

HRB

E/Ex

plor

er-1

PR

IME-

2Sc

ienc

eM

onito

r CC

1000

UH

FU

HF

850

1200

BPSK

N.A

AX

.25

UI

Hal

f Dup

lex

Succ

ess

http

://sp

aceg

rant

.mon

tana

.edu

/Exp

lore

r1Pr

ime.

htm

lM

-Cub

eTe

chno

logy

dem

onstr

atio

nIm

agin

gA

DF7

020-

1V

HF

UH

F10

0012

00FS

KCW

AX

.25

UI

Full

Dup

lex

Succ

ess

http

://w

ww

-per

sona

l.um

ich.

edu/

~mjre

gan/

MCu

bed/

Page

s/Doc

umen

ts/M

-Cub

edA

IAA

Pape

r.pdf

RAX

-2(T

T&C)

Scie

nce/

Tech

nolo

gy

dem

onstr

atio

nSt

udy

Plas

ma f

ield

-alig

ned

irreg

ular

ities

in

iono

sphe

re.

Lith

ium

-1U

HF

UH

F10

0096

00.0

0G

MSK

N.A

AX

.25

UI

N.A

Succ

ess

http

s://d

irect

ory.

eopo

rtal.o

rg/w

eb/e

opor

tal/s

atel

lite-

miss

ions

/r/ra

x-2

RAX

-2(D

ata)

Scie

nce/

Tech

nolo

gy

dem

onstr

atio

nSt

udy

Plas

ma f

ield

-alig

ned

irreg

ular

ities

in

iono

sphe

re.

Mic

roha

rd M

HX

240

0S-

band

S-ba

nd10

001.

15E+

05FH

SSN

.APr

oprie

tary

N.A

Succ

ess

2012

19V

ega V

V01

Xat

cobe

oTe

chno

logy

dem

onstr

atio

nV

erifi

catio

n of

a ne

w sy

stem

for m

easu

ring

the

amou

nt o

f ion

izin

g ra

diat

ion

(RD

S)G

omSp

ace

U48

2CV

HF

UH

F50

012

00M

SKCW

AX

.25

UI

Full

Dup

lex

Succ

ess

http

://w

ww

.xat

cobe

o.co

m/c

ms/i

ndex

.php

?opt

ion=

com

_con

tent

&vi

ew=a

rticl

e&id

=41&

Item

id=1

6

ROBU

STA

Educ

atio

n/Te

chno

logy

de

mon

strat

ion

Mon

itor t

he d

eter

iora

tion

of in

-flig

ht

elec

troni

cs.

MC1

2181

VH

FU

HF

800

1200

AFS

KCW

AX

.25

UI

Full

Dup

lex

Parti

al-S

ucce

ssht

tps:/

/dire

ctor

y.eo

porta

l.org

/web

/eop

orta

l/sat

ellit

e-m

issio

ns/r/

robu

stae-

st@r

Educ

atio

nA

DCS

, Sat

ellit

e bu

sBH

X2-

437-

5U

HF

UH

F50

012

00A

FSK

CWA

X.2

5 U

IH

alf D

uple

xPa

rtial

-Suc

cess

http

://w

eb.ar

chiv

e.or

g/w

eb/2

0120

8291

1183

8/ht

tp://

aree

web

.pol

ito.it

/rice

rca/

E-ST

AR/

#

Gol

iat

Tech

nolo

gy d

emon

strat

ion

Imag

ing,

mea

sure

s rad

iatio

n le

vels

Alin

co D

J-C7

UH

FU

HF

300

1200

AFS

KCW

AX

.25

UI

Hal

f Dup

lex

Parti

al-S

ucce

ss

http

s://w

ww

.goo

gle.

com

/url?

sa=t

&rc

t=j&

q=&

esrc

=s&

sour

ce=w

eb&

cd=3

&ve

d=0C

DoQ

FjA

C&ur

l=ht

tp%

3A

%2F

%2F

ww

w.ro

sa.ro

%2F

inde

x2.p

hp%

3Fop

tion%

3Dco

m_r

esou

rce%

26ta

sk%

3Ddo

wnl

oad2

%26

no_h

tml%

3D1%

26fil

e%3D

NV

9JQ

UM

tMD

gtQ

jQtN

i1BM

TMxL

nBkZ

g%3D

%3D

%26

dow

nloa

dNam

e%3D

SUFD

LTA

4LU

I0LT

YtQ

TEzM

S5w

ZGY

%3D

%26

id%

3D31

&ei

=bqX

KU

u3EL

LCN

0wX

4mY

BY&

usg=

AFQ

jCN

EuLg

17fX

60jz

YD

4oU

KM

zPs_

gwsc

A&

sig2=

M47

PN-1

glM

Fwgz

iaab

-gI

g&bv

m=b

v.58

1871

78,d

.d2k

&ca

d=rja

Gol

iat

Tech

nolo

gy d

emon

strat

ion

Imag

ing,

mea

sure

s rad

iatio

n le

vels

Mic

roha

rd M

HX

-242

0N

.AS-

band

1000

1.15

E+05

GFS

KN

.APr

oprie

tary

PW-S

atEd

ucat

ion/

Tech

nolo

gy

dem

onstr

atio

nde

-orb

iting

sail

ISIS

TRX

UV

UH

FV

HF

200

1200

BPSK

CWA

X.2

5 U

IFu

ll D

uple

xPa

rtial

-Suc

cess

http

://w

ww

.pw

-sat

.pl/

Mas

at-1

Educ

atio

n/Te

chno

logy

de

mon

strat

ion

Imag

ing,

Sat

ellit

e bu

sSi

4432

UH

FU

HF

400

1200

GFS

KCW

AX

.25

UI

Hal

f Dup

lex

Succ

ess

http

://cu

besa

t.bm

e.hu

/en/

proj

ekte

k/m

asat

-1/

Uni

Cube

Sat-G

GEd

ucat

ion/

Scie

nce

Stud

y gr

avity

gra

dien

t usin

g a d

eplo

yabl

e bo

om.

Astr

oDev

N.A

UH

F50

096

00G

FSK

CWA

X.2

5 U

IN

.APa

rtial

-Suc

cess

http

s://d

irect

ory.

eopo

rtal.o

rg/w

eb/e

opor

tal/s

atel

lite-

miss

ions

/u/u

nicu

besa

t-gg

Hor

yu-2

Tech

nolo

gy d

emon

strat

ion

VH

FU

HF

1000

1200

FSK

AX

.25

UI

Full

Dup

lex

Succ

ess

http

://w

ww

.raum

fahr

ttech

nik.

tu-b

erlin

.de/

men

ue/p

ublic

atio

ns/sm

all

sate

llite

data

base

/par

amet

er/e

n/N

ASA

ELa

Na-

6/N

ROL-

36SM

DC-

ON

E 2.

1M

ilita

ryCo

mm

unic

atio

nPe

ricle

UH

FV

HF

N.A

N.A

N.A

N.A

N.A

N.A

Succ

ess

http

://w

ww

.duc

omm

un.co

m/p

df/S

MD

C-O

NEM

edia

Dec

k.pd

fSM

DC-

ON

E 2.

2M

ilita

ryCo

mm

unic

atio

nPe

ricle

UH

FV

HF

N.A

N.A

N.A

N.A

N.A

N.A

Succ

ess

http

://w

ww

.duc

omm

un.co

m/p

df/S

MD

C-O

NEM

edia

Dec

k.pd

fA

eroC

ube-

4AM

ilita

ryCo

mm

unic

atio

nFr

eew

are

MM

2U

HF

UH

F20

003.

84E+

04FS

KN

.APr

oprie

tary

N.A

Succ

ess

http

s://d

irect

ory.

eopo

rtal.o

rg/w

eb/e

opor

tal/s

atel

lite-

miss

ions

/a/a

eroc

ube-

4A

eroC

ube-

4BM

ilita

ryCo

mm

unic

atio

nFr

eew

are

MM

2U

HF

UH

F20

003.

84E+

04FS

KN

.APr

oprie

tary

N.A

Succ

ess

http

s://d

irect

ory.

eopo

rtal.o

rg/w

eb/e

opor

tal/s

atel

lite-

miss

ions

/a/a

eroc

ube-

4A

eroC

ube-

4CM

ilita

ryLa

unch

veh

icle

env

ironm

ent d

ata l

ogge

rCC

1101

N.A

UH

F13

005.

00E+

05FS

KN

.APr

oprie

tary

N.A

Succ

ess

http

s://d

irect

ory.

eopo

rtal.o

rg/w

eb/e

opor

tal/s

atel

lite-

miss

ions

/a/a

eroc

ube-

4A

enea

sTe

chno

logy

dem

onstr

atio

nPa

rabo

lic d

ish,

GS

track

ing

Mic

roha

rd M

HX

-425

UH

FU

HF

1000

1920

0A

FSK

AX

.25

Prop

rieta

ryFu

ll D

uple

xSu

cces

sht

tp://

ww

w.is

i.edu

/pro

ject

s/ser

c/ae

neas

CSSW

EEd

ucat

ion/

Scie

nce

Stud

y th

e re

latio

n be

twee

n so

lar f

lare

s and

en

erge

tic p

artic

les.

SX12

31U

HF

UH

F10

0096

00G

MSK

AX

.25

AX

.25

UI

Hal

f Dup

lex

Succ

ess

http

s://d

irect

ory.

eopo

rtal.o

rg/w

eb/e

opor

tal/s

atel

lite-

miss

ions

/c-m

issio

ns/c

ssw

eCP

5Te

chno

logy

dem

onstr

atio

nD

e-or

bitin

g ex

perim

ent

CC10

00U

HF

UH

F10

0012

00A

FSK

N.A

AX

.25

UI

Hal

f Dup

lex

N.A

http

://po

lysa

t.cal

poly

.edu

/CP5

.php

CXBN

Eudc

atio

n/Te

chno

logy

de

mon

strat

ion

Spec

trom

eter

Lith

ium

-1U

HF

UH

F75

03.

84E+

04G

FSK

N.A

AX

.25

UI

N.A

Succ

ess

http

s://d

irect

ory.

eopo

rtal.o

rg/w

eb/e

opor

tal/s

atel

lite-

miss

ions

/c-m

issio

ns/c

xbn

CIN

EMA

-1Sc

ienc

ePa

rticl

e de

tect

or an

d m

agne

tom

eter

sEm

hise

rU

HF

S-ba

nd10

001.

00E+

06FS

KN

.APr

oprie

tary

Full

Dup

lex

Succ

ess

http

://sp

ace.

skyr

ocke

t.de/

doc

sdat

/cin

ema.h

tm

STA

RE (R

e)Te

chno

logy

dem

onstr

atio

nTe

lesc

ope

(look

ing

for n

ear-e

arth

spac

e de

bris)

Hel

ium

-100

UH

FU

HF

1000

5.76

E+04

FSK

N.A

AX

.25

UI

Full

Dup

lex

Faile

dht

tp://

wee

bau.

com

/sate

llite

/S/st

are1

.htm

2013

83PS

LV C

20ST

RaN

D-1

Tech

nolo

gy d

emon

strat

ion

And

roid

mob

ile p

hone

, im

agin

gN

.AV

HF

UH

FN

.A9.

60E+

03G

MSK

CWA

X.2

5 U

IFu

ll D

uple

xSu

cces

sht

tp://

ww

w.ss

tl.co

.uk/

Div

ision

s/Ear

th-O

bser

vatio

n---S

cien

ce/S

cien

ce---

Expl

orat

ion/

STRa

ND

-nan

osat

ellit

e/IA

C11-

Ken

yon-

STRa

ND

-1-B

4-6B

-8

AA

U-C

ube-

3Ed

ucat

ion/

Tech

nolo

gy

dem

onstr

atio

nA

IS rx

, Sat

ellit

e bu

sN

.AU

HF

UH

FN

.A2.

40E+

0324

00CW

AX

.25

UI

Hal

f Dup

lex

Succ

ess

http

://w

ww

.spac

e.aa

u.dk

/aau

sat3

/inde

x.ph

p?n=

Tech

.AA

USA

T3In

Det

ails

Uni

BRIT

E (C

anX

-3A

)Sc

ienc

ePh

otom

etric

obs

erva

tion

of b

right

star

sse

lf de

velo

ped

UH

FS-

band

500

2.56

E+05

N.A

CWPr

oprie

tary

Full

Dup

lex

http

s://d

irect

ory.

eopo

rtal.o

rg/w

eb/e

opor

tal/s

atel

lite-

miss

ions

/b/b

rite-

austr

ia

Soyu

z-2-

1aD

ove-

2Te

chno

logy

de

mon

strat

ion/

Indu

stry

Imag

ing

payl

oas

Mic

roha

rd M

HX

-240

0U

HF

S-ba

nd10

003.

00E+

04FH

SSCW

Prop

rieta

ryH

alf D

uple

xSu

cces

s h

ttp://

digi

talc

omm

ons.u

su.e

du/c

gi/v

iew

cont

ent.c

gi?f

ilena

me=

0&ar

ticle

=287

2&co

ntex

t=sm

allsa

t&ty

pe=a

dditi

onal

BeeS

at-2

Educ

atio

n/Te

chno

logy

de

mon

strat

ion

3-A

xis a

ttitu

de co

ntro

l in

orbi

tCM

X90

9BU

HF

UH

F50

04.

80E+

03G

MSK

CWM

obite

xH

alf D

uple

xSu

cces

sht

tps:/

/dire

ctor

y.eo

porta

l.org

/web

/eop

orta

l/sat

ellit

e-m

issio

ns/b

/bee

sat-2

-3

BeeS

at-3

Educ

atio

n/Te

chno

logy

de

mon

strat

ion

S-ba

nd tr

ansm

itter

HIS

PICO

HIS

PICO

UH

FS-

band

500

2.00

E+06

DQ

PSK

CWM

obite

xFu

ll D

uple

xSu

cces

sht

tps:/

/dire

ctor

y.eo

porta

l.org

/web

/eop

orta

l/sat

ellit

e-m

issio

ns/b

/bee

sat-2

-3

SOM

PEd

ucat

ion/

Scie

nce

Mea

sure

atom

ic o

xyge

n in

upp

er

atm

osph

ere.

SDR&

Si44

20U

HF

UH

FN

.A1.

80E+

05Q

PSK

CWPr

oprie

tary

Hal

f Dup

lex

Parti

al-S

ucce

ssht

tp://

phpw

eb.tu

-dre

sden

.de/

stard

/SO

MP/

?lan

g=en

OSS

I1Te

chno

logy

dem

onstr

atio

nLE

D o

ptic

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http://www-personal.umich.edu/~mjregan/MCubed/Pages/Documents/M-CubedAIAAPaper.pdf

https://directory.eoportal.org/web/eoportal/satellite-missions/r/rax-2

http://www.xatcobeo.com/cms/index.php?option=com_content&view=article&id=41&Itemid=16

https://directory.eoportal.org/web/eoportal/satellite-missions/r/robusta

mailto:e-st@r

http://web.archive.org/web/20120829111838/http:/areeweb.polito.it/ricerca/E-STAR/

https://www.google.com/url?sa=t&rct=j&q=&esrc=s&source=web&cd=3&ved=0CDoQFjAC&url=http%3A%2F%2Fwww.rosa.ro%2Findex2.php%3Foption%3Dcom_resource%26task%3Ddownload2%26no_html%3D1%26file%3DNV9JQUMtMDgtQjQtNi1BMTMxLnBkZg%3D%3D%26downloadName%3DSUFDLTA4LUI0LTYtQTEzMS5wZGY%3D%26id%3D31&ei=bqXKUu3ELLCN0wX4mYBY&usg=AFQjCNEuLg17fX60jzYD4oUKMzPs_gwscA&sig2=M47PN-1glMFwgziaab-gIg&bvm=bv.58187178,d.d2k&cad=rja






http://www.pw-sat.pl/

http://cubesat.bme.hu/en/projektek/masat-1/

https://directory.eoportal.org/web/eoportal/satellite-missions/u/unicubesat-gg

http://www.ducommun.com/pdf/SMDC-ONEMediaDeck.pdf

http://www.ducommun.com/pdf/SMDC-ONEMediaDeck.pdf




http://www.isi.edu/projects/serc/aeneas

https://directory.eoportal.org/web/eoportal/satellite-missions/c-missions/csswe

http://polysat.calpoly.edu/CP5.php

https://directory.eoportal.org/web/eoportal/satellite-missions/c-missions/cxbn

http://space.skyrocket.de/doc_sdat/cinema.htm

http://weebau.com/satellite/S/stare1.htm

http://www.sstl.co.uk/Divisions/Earth-Observation---Science/Science---Exploration/STRaND-nanosatellite/IAC11-Kenyon-STRaND-1-B4-6B-8

http://www.space.aau.dk/aausat3/index.php?n=Tech.AAUSAT3InDetails

https://directory.eoportal.org/web/eoportal/satellite-missions/b/brite-austria

http://digitalcommons.usu.edu/cgi/viewcontent.cgi?filename=0&article=2872&context=smallsat&type=additional

https://directory.eoportal.org/web/eoportal/satellite-missions/b/beesat-2-3

https://directory.eoportal.org/web/eoportal/satellite-missions/b/beesat-2-3

http://phpweb.tu-dresden.de/stard/SOMP/?lang=en

http://opensat.cc/wiki/ossi1beacon

http://digitalcommons.usu.edu/cgi/viewcontent.cgi?filename=0&article=2872&context=smallsat&type=additional

http://www.phonesat.org/index.php



https://www.google.com/url?sa=t&rct=j&q=&esrc=s&source=web&cd=3&ved=0CD4QFjAC&url=http%3A%2F%2Fwww.dtic.mil%2Fcgi-bin%2FGetTRDoc%3FAD%3DADA531480&ei=DFvNUpmmEKem0QXRzIEg&usg=AFQjCNGUtehry5B4PimCXdOXYjhGjKSEGg&sig2=UFJvpCfdTbhl0bWdoc1B_Q&bvm=bv.58187178,d.d2k&cad=rja




http://mstl.atl.calpoly.edu/~bklofas/Presentations/DevelopersWorkshop2012/La_Tour_SENSE.pdf

http://mstl.atl.calpoly.edu/~bklofas/Presentations/DevelopersWorkshop2012/La_Tour_SENSE.pdf

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6

2. Spectral efficiency of the modulation schemes proposed in DVB.S2 for different coding rates.

7

3. CATENA’s final LINC PA configuration:

8

Drain voltage and current waveform:

Final_29_09_v3 (1)

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Transcript of Final_29_09_v3 (1)