Development of Dielectric Resonator Antenna
Transcript of Development of Dielectric Resonator Antenna
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Development of Dielectric
Resonator Antenna (DRA)K. W. Leung
State Key Laboratory of Millimeter Waves &
Department of Electronic Engineering,City University of Hong Kong
Presented to the IEEE LI Section Antenna and
Propagation Society on Monday October 8th 2012
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I. Introduction
II. Circularly Polarized DRA Using a Parasitic Strip
III. Frequency Tuning Technique
IV. Omnidirectional Circularly Polarized DRAs
V. Dualband & Wideband DRAs
VI. Dualfunction DRAs
Outline
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3
The DRA is an antenna that makes use of a radiating modeof a dielectric resonator (DR).
It is a 3-dimensional device of any shape,
e.g., hemispherical, cylindrical, rectangular,
triangular, etc.
Resonance frequency determined by the its dimensions and
dielectric constant r.
What is Dielectric Resonator Antenna (DRA) ?
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Some DRAs :
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Advantages of the DRA
Low cost
Low loss (no conductor loss) Small size and light weight
Reasonable bandwidth (~10% for r ~10)
Easy of excitation High radiation efficiency ( generally > 95%)
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Excitation schemes
(i) Microstrip line feed
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Ground plane
Microstrip
feed line
Feed
Substrat
DRA
Slot
(ii) Aperture-couple feed
Excitation schemes
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(iii) Coaxial feed
Excitation schemes
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Bottom viewTop view
Coaxial feed
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Bottom view Top view
Aperture-coupled feed
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Slot-fed DRA array using corporate
microstrip feed network
Corporate feedline for DRA array
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Conformal-Strip Method
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Rectangular DielectricResonator Antennas
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Proposed Antenna Geometry
W1
l1
a
d
b
x
y
zGround
Plane
Rectangular
DRA
Coaxial
Aperture
Conducting
Strip
r
a
(mm)
b
(mm)
d
(mm)
l1(mm)
W1(mm)
r
14.3 25.4 26.1 10 1 9.8
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Resonant frequency of TEmnl(y) mode
Analytical Solution
Dielectric Waveguide Model (DWM)
222
0
2zyx
r
kkkcf
d
l
kb
n
ka
m
k zyx 2,,
2
0
222 kkkk rzyx
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Numerical Solution
Advantages
- Very simple- High modeling capability for general EM structures
- No spurious modes nor large matrix manipulation
- Provide a very wideband frequency response
Finite-Difference Time-Domain (FDTD) method
Disadvantages
- Time consuming, powerful computer required
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Baseband Gaussian pulse
])3(exp[22 TTntEz T : pulse width
0
011
)(
)(
ZZ
ZZS
tIFFT
tVFFTZ
in
in
in
Conformal Strip
Ground Plane
Ez (V)Hy Hy
HxHx
(I)
Source occupies only one grid
Source model and extraction of S parameters
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Uniform Cartesian grids
T = 0.083ns, t0
= 3T
10-cell-thick PML with polynomial spatial scaling
(m = 4 and max = 1)
total grid size : 80x 110y 112z
total time steps : 10000
x= 0.715 mm, y= 0.508 mm, z= 0.5 mm
Parameters
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3 3.5 4 4.5 5 5.5 6
-100
-50
0
50
100
150
200
Frequency (GHz)
2 2.5 3 3.5 4 4.5 5 5.5 6-30
-25
-20
-15
-10
-5
0
ExperimentTheory
Frequency (GHz)
|S11|(dB)
InputImpedance(ohm)
ExperimentTheory
Resistance
Reactance
Input Impedance/S11
Reasonable agreement.
Wide Bandwidth of ~ 43%.
Dual resonantTE111y and TE113y modes are excited.
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y
111TE
y
113TE
ResonantModes
Measuredresonant
frequencies
Calculated resonantfrequencies (FDTD)
Predicted resonantfrequencies (DWM)
fmea
(GHz)fFDTD
(GHz)
error
(%)
fDWM
(GHz)
error
(%)
3.81 3.90 2.3 3.95 3.6
N/A N/A N/A 4.26 N/A
4.57 4.60 0.7 4.7 1.7
Comparison between Theory and Measurement
y
112TE
Reasonable agreement.
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Field Distribution --- TE111y
Electric field
Magnetic field
z
2dx
b
x-z
bx
y
a
x-y
Imaged DRA (gound plane removed) With gound plane
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Field Distribution --- TE112y
x-z
bx
y
a
x-yElectric field
Magnetic field
z
2d x
b
Imaged DRA (gound plane removed)
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Field Distribution --- TE113y
x-zElectric field
Magnetic field
bx
y
a
x-y
z
2dx
b
With gound planeImaged DRA (gound plane removed)
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E (xz) - plane H (yz) - plane(+x)(-x) (-y) (+y)-40
0o
-30 -20-10 0
30o
30o
60o
60o
90o
90o
-40 -30 -20-10 0
0o
30o
30o
60o
60o
90o
90o
E (xz) - plane H (yz) - plane(+x) (-y)(-x) (+y)
-40 -30-20-10 0
0o
30o
30o
60o
60o
90o 90o
-40 -30-20-10 0
0o
30o
30o
60o
60o
90o 90o
f= 3.5 GHz
f= 4.3 GHz
Radiation Patterns
Broadside radiation patterns are observed.
Measured E-plane crosspolarized fields mainly caused by finite
ground plane diffraction.
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III. Circularly Polarized Design
using a Parasitic Strip
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Proposed Antenna Geometry
a
(mm)
b
(mm)
d
(mm)
l1(mm)
W1(mm)
l2(mm)
W2(mm)
0(degree)
r
24 23.5 12.34 10 1 12 1 225.6 9.5
W1
l1
ad
b
x
y
z
Ground
Plane
Rectangular
DRA
Coaxial
ApertureConducting
Strip
r
C
W2l2 Parasitic
Patch
(Center of
Parasitic Patch)
0
Top
ViewParasitic
Patch
Conducting
Strip
x
y
r
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Input Impedance/S11
Reasonable agreement.
Bandwidth ~ 14%.
Two nearly-degenerate TE111(y) modes are excited.
CP operation
2.5 3 3.5 4 4.5-25
-20
-15
-10
-5
0
|S11|(
dB)
Frequency (GHz)
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Axial Ratio in the boresight direction
3-dB AR bandwidth is ~ 2.7%, which is a typical value for
a singly-fed CP DRA.
3 3.1 3.2 3.3 3.4 3.5 3.6 3.7 3.8
0
5
10
15
20
AxialRat
io(dB)
Frequency (GHz)
ExperimentTheory
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The H field of the DRA without and with parasitic
strip (Top view)
Without parasitic strip - LP field With parasitic strip - CP field
3.4 GHz 3.4 GHz
Feeding strip Feeding strip
Parasitic
strip
29
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Radiation Patterns (f= 3.4GHz, )
LHCP
RHCP
xz plane yz plane(+x)(-x) (-y) (+y)
-40 -30 -20-10 0
0o
30o
30o
60o
60o
90o
90o
-40 -30 -20-10 0
0o
30o
30o
60o
60o
90o
90o
A broadside radiation mode is observed.
For each radiation plane, the LHCP field is more than 20dB
stronger than the RHCP field.
The maximum gain is 5.7 dBic (not shown here).
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Effects of feeding strip length l1
Input impedance changes substantially with l1.
AR is almost unchanged for different l1.
l1 can be adjusted to match the impedance without changing AR.
2.4 2.6 2.8 3 3.2 3.4 3.6 3.8 4 4.2-20
-15
-10
-5
0
|S11|
(dB)
Frequency (GHz)
l1 = 8 mm
l1 = 10 mm
l1 = 12 mm
3.1 3.2 3.3 3.4 3.5 3.6 3.70
5
10
15
20
AxialRa
tio(dB)
Frequency (GHz)
l1 = 8 mm
l1 = 10 mm
l1 = 12 mm
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II. Frequency Tuning Technique
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The DRA for a paticular frequency may not be
available from the comericial market.
Fabrication tolerances cause errors betweenmeasured and calculated resonant frequencies.
Frequency tuning methods:(i) loading-disk; and
(ii) parasitic slot.
Backgruond
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Frequency Tuning Technique- using a loading disk
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Side view Top view
The slot-coupled DRA with a conducting loading
cap
d
z
Slot
Hemispherical DRA
Dielectric
substrate rs
microstrip
line
Lt
Ground
planeara
xConducting
Loading Cap
Lt
x
y
Ls
Ws
Hemispherical
DRA
microstrip
line
Slot
Wf
Conducting
Loading Cap
Hemispherical DRA: radius a= 12.5 mm, dielectric constant r
= 9.5.
Coupling slot : length L s , width WsOpen-circuit stub: length L
t
Grounded dielectric slab: rs
= 2.33, height d= 1.57 mm
Microstrip feedline: width Wf= 4.7 mm
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Calculated and measured return losses
(L s = 12 mm and Ws = 1 mm)
3 3.2 3.4 3.6 3.8-70
-60
-50
-40
-30
-20
-10
0
Frequency (GHz)
|S11|(dB) = 26.38
Lt = 4.42 mm
o
= 52.8Lt= 13.6 mm
o
Theory
Experiments
Without cap (= 0 )Lt= 10.63mm
o
Resonance frequency:
3.52 GHz without any conducting cap ( = 00), withLt = 4.42 mm
3.25 GHz ( = 26.38o andLt = 4.42 mm)
3.68 GHz ( = 52.8o andLt = 13.6 mm)
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Calculated and measured radiation patterns
-40 -30 -20 -10 0
0
30
60
0
30
60
90o o
o o
o o
o
(a) (b)
-40 -30 -20 -10 0
0
30
60
90
30
60
90o o
o o
o o
oCo-pol
X-pol
-40 -30 -20 -10 0
0
30
60
0
30
60
90o o
o o
o o
o
(c)
-40 -30 -20 -10 0
0
30
60
90
30
60
90o o
o o
o o
o
(d)
Co-pol
X-pol
3.58 GHz ( = 52.8o
andLt = 13.6 mm)
Reasonable agreement
between theory andexperiment.
The effect of loading
cap on field pattern is not
significant.
3.25 GHz ( = 26.38o andLt = 4.42 mm)
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Calculated and L t for having a good return loss
(minimum |S11| < -20dB)
3.2 3.3 3.4 3.5 3.6 3.70
10
20
30
40
50
60
2
4
6
8
10
12
14
16
Tuning frequencyfr(GHz)
Angle(d
egree)
StubLengthLt(mm)
The resonant frequency can be tuned by varying and L t decreases from 26.38o to 0o(3.25
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Impedance bandwidth
The bandwidth decreases after a loading cap is added.
3.2 3.3 3.4 3.5 3.6 3.7
2
4
6
8
10
Frequency (GHz)
ImpedanceBandw
idth(%)
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Frequency Tuning Technique
- using a parasitic slot
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(a) Side view
(b) Top view
The annular-slot-excited cavity-backed DRA
z
x
Hemispherical DRA
Coaxial cable Parasitic slot
a
r
metallic cavity b
Ground plane
WB WA
Feeding slotrB
rA
Coaxial cable
Hemispherical DRA Feeding slot
a
r
metallic cavity
b
Parasitic slot
A
rA rBWB
WA
y
x
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IV. Omnidirectional CircularlyPolarized DRA
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CP DRAs concentrated on broadside-mode designs only.
Provide larger coverage.
Advantages of omnidirectional CP antenna
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Slotted omnidirectional CP DRA
Design I:
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Antenna configurations
h
wd
a
b
Dielectric
block
x
h
g
z
Slot
SMA connector
w
Probel
12r
Perspective view Front view
Dielectric cube with oblique slots (polarizer) fabricated onits four sidewalls.
Centrally fed by a coaxial probe extended from a SMAconnector, whose flange used as the small ground plane.
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Wave polarizer
h
wd
a
b
Dielectric
block
LP omnidirectional DRA
Dielectric block
Pol
ariz
er
Dielectric block with the wave
polarizer
Proposed compact omnidirectional CP DRA
x
y
z
Dielectricslabs
D
E
+
Antenna principle
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Prototype for 2.4 GHz WLAN design
Top face and sidewalls Bottom face
Design parameters
r= 15, a = b = 39.4 mm, h = 33.4 mm, w = 9.4 mm,
d =14.4 mm, r1= 0.63 mm, l= 12.4 mm,g= 12.7 mm
Photographs of the prototype
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Simulated and measured results
Reflection coefficient
Impedance bandwidth: AR bandwidth:Simulated: 20.3% (2.34-2.87 GHz) Simulated: 8.2% (2.34-2.54 GHz)Measured: 24.4% (2.30-2.94 GHz) Measured: 7.3% (2.39-2.57 GHz)
Axial ratio
2 2.2 2.4 2.6 2.8 3
-30
-20
-10
0
Frequency (GHz)
|S11| (dB)
HFSS Simulation
Experiment
Axial Ratio (dB)
HFSS Simulation
Experiment
2 2.2 2.4 2.6 2.8 30
3
6
9
Frequency (GHz)
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Very good omnidirectional characteristic
In the horizontal plane,LHCP fields > RHCP fields by~20 dB .
Simulated and measured radiation patterns
LHCP
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o o
o o
o
dB
xz-plane
0o
RHCP
(+x)(-x)
-40 -30 -20 -10 0
30
210
60
240
90270
120
300
150
330
180
o
o o
o
o o
o o
o o
o
dB
xy-plane
0o
RHCP
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Simulated and measured antenna gain
2 2.2 2.4 2.6 2.8 3-3
-2
-1
0
1
2
3
Frequency (GHz)
Gain (dBic)
HFSS Simulation
Experiment
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Wideband omnidirectional CP antennawith parasitic metallic strips
Design II:
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Perspective view Front view
x
h
g
z
ls
ws
Slot
Strip
SMA connector
w
Probe
Hollow circular
cylinder
l
12r
Four parasitic metallic strips are embedded in thelateral slots to enhance the AR bandwidth.
The hollow circular cylinder is introduced to enhancethe impedance bandwidth.
Antenna configurations
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Photographs of the prototype
Top face and sidewalls Bottom face
Design parametersr= 15, a= b= 30 mm, h= 25 mm, r= 3 mm, w =7 mm, d =10.5 mm
ls
= 30.5 mm, ws
= 1 mm, x0 = 6.4 mm, r1= 0.63 mm, l= 19 mm.
Prototype for 3.4 GHz WiMAX design
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Overlapping bandwidth: 22.0%; bandwidth widened by ~3 times.
Simulated and measured reflection
coefficient and axial ratio
Axial Ratio (dB)
2.8 3.2 3.6 4-30
-20
-10
0
Frequency (GHz)
|S11| (dB)
HFSS Simulation
Experiment
2.8 3.2 3.6 40
6
12
18
Frequency (GHz)
Impedance bandwidth:Simulated: 22.3% (3.11-3.89 GHz)Measured: 24.5% (3.08-3.94 GHz)
AR bandwidth:Simulated: 24.8% (3.11-3.99 GHz)Measured: 25.4% (3.16-4.08 GHz)
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Measured gain: wider bandwidth.
Measured antenna efficiency: 84-98% (3.1-3.9 GHz).
Antenna gain
Simulated and measured results
Radiation efficiency
Gain (dBic)
HFSS Simulation
Experiment
2.8 3.2 3.6 4-4
-3
-2
-1
0
1
2
3
Frequency (GHz)2.8 3.2 3.6 40
0.2
0.4
0.6
0.8
1
Frequency (GHz)
Efficiency
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3.4 GHz 3.8GHz
LHCP fields > RHCP fields by more than 15 dB in horizontal plane. Stableradiation patterns across the entire passband (3.13.9 GHz).
LHCP
-40 - 30 - 20 -10 0
30
210
60
240
90270
120
300
150
330
180
o
o o
o
o o
o o
o o
o
dB
xy-plane
0o
RHCP
-40 - 30 - 20 -10 0
30
150
60
120
9090
120
60
150
30
180
o o
o o
o o
o o
o o
o
dB
xz-plane
0o
RHCP
(+x)(-x)
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o o
o o
o
dB
xz-plane
0oLHCP
RHCP
(+x)(-x)
-40 -30 -20 -10 0
30
210
60
240
90270
120
300
150
330
180
o
o o
o
o o
o o
o o
o
dB
xy-plane
0o
RHCP
Simulated and measured radiation patterns
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V. Dualband & Wideband DRAs
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(i) Rectangular DRA
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Dualband and wideband antennas are extensively used
(e.g., WLAN)
Multi-element DRA [1]
- requiring more DR elements and space
Hybrid slot-DRA [2]
- coupling slot used as the feed and antenna
- inflexible in matching the impedance
[1] Petosa, N. Simons, R. Siushansian, A. Ittipiboon and C. Michel, Design and analysis of
multisegmentdielectric resonator antennas, IEEE Trans. AP, vol.48, pp.738-742, 2000.
[2] Buerkle, K. Sarabandi, and H. Mosallaei, Compact slot and dielectric resonator antenna with
dual-resonance, broadband characteristics,IEEE Trans. AP, vol. 53, pp.1020-1027, 1983.
Background
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Wideband DRA [1]
Dualband DRA [2]
Trial-and-error approach is normally used
Systematic design approach is desirable[1] B. Li and K. W. Leung, Strip-fed rectangular dielectric resonator antennas with/without a
parasitic patch,IEEE Trans. Antennas Propagat., vol.53, pp.2200-2207, Jul.2005.
[2] T. H. Chang and J. F. Kiang, Dual-band split dielectric resonator antenna, IEEE Trans.
Antennas Propagat., vol.55, no.11, pp.3155-3162, Nov.2007.
Use of higher-order DRA
mode
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Design Formulas for Dual-Mode rectangular DRA
yTE111yTE112
yTE113
The E-field should vanish on the PEC and the TE112 mode
cannot be excited properly.
The TE111 mode and TE113 mode are used in the dual-
mode design.
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x
y
z
b
a
d
Ground plane
Rectangular DRAr
Formula Derivation
The wavenumbers kx1,x2 and
kz1,z2 can be written as follows:
akk xx
21
dkz
21
dkz
2
32
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From the DWM model, the frequenciesf1,f2 are given by:
2
2,1
2
2,1
2
2,12,1 2 zzyyxxr
kkkc
f
2
2,1
2
2,1
2
2,12,1 zzxxyy kkkk
in which are wavenumbers in the
dielectric, with c being the speed of light in vacuum.
where
cfk r /2 2,12,1
(*)
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dkkd 2122
2
)96.3(
22
21
1
2
32.10
9
32.10 ff
kka
21 35.065.0 bbb
Engineering Formulas for the DRA dimensions
2
1
23
1
24
1
2 4422.113209.21393.0f
f
f
f
f
fd
3
1
2 104437.184984.23
f
f(m)
11
1tan2
2
2,1
2,11
2,1
2,1
yyryy k
k
kb
where
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65
)96.3(
22
21
1
2
32.10
9
32.10 ff
kka
Limit of frequency ratio f2/f1
From
3k1 > k2 or 3f1 >f2We have
giving
f2/f1 < 3
which is the theoretical limit that is not known before.
09 2221 dkk
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66
1 1.2 1.4 1.6 1.8 2 2.2 2.4 2.6 2.80
1
2
3
4
5r
10r
30r
70r
Error of (%)1f
f f2 1/
Compared with DWM results, errors off1, f2 are both
less than 2.5% for 1
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67
A. Example for Dual-band Rectangular DRA Design
a = 20.8 mm, b = 10.5 mm, and d= 18.5 mm.
Given:f1 = 3.47 GHz (WiMAX)
f2 = 5.2 GHz (WLAN), r=10
Using dual-mode
formulas
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68
Configuration of the dualband DRA
b
ad
Ground plane
Microstripline
Coupling
slot
LSW
L
Wf
h
Substraters
Rectangular
DRAr
z
y
x
W= 2.6 mm,L =10.6 mm,Ls=7.2 mm, Wf=1.94 mm,
h=0.762mm, rs= 2.93
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69
Measured and simulated reflection coefficients
Measured bandwidths:
Lower band: 15% (3.25-3.78 GHz) covering WiMAX (3.4-3.7 GHz).
Upper band: 8.3% (5.03-5.47 GHz) covering WLAN (5.15-5.35 GHz).
3.2 3.6 4 4.4 4.8 5.2 5.6-40
-30
-20
-10
0
HFSS Simulation
Measurement
Frequency (GHz)
Reflection coefficient |S | (dB)11
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70
COMPARISON OF DESIGN, SIMULATED, AND MEASURED
RESONANCE FREQUENCIES OF TE111y AND TE113
y MODES
Resonant
Mode
Measured
frequency
(GHz)
Design
frequency
Simulated HFSS
frequency
f1,2(GHz)
Error(%)
fHFSS(GHz)
Error(%)
TE111y 3.40 3.47 2.05 3.47 2.05
TE113y
5.18 5.30 2.32 5.24 1.15
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71
TE111y mode: measured (3.40 GHz), simulated (3.47 GHz).
Broadside radiation patterns are observed for both planes.
Co-polarized fields > cross-polarized fields by more than 20 dB in
the boresight direction.
Measured and simulated radiation patterns
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
0oo
o o
o
o o
o o
o o
o
dB
9090
150
180
o
o
o
o
o
dB
-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
0 oo o
o o
o o
o o
o o
o
dB
(a)
Simulation
H-plane (y-z plane)
Measurement
E-plane (x-z plane)
-x
120
Co-pol
Cross-pol
Co-pol
Cross-pol
+x -y +y
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72
Measured and simulated radiation patterns
TE113y mode: measured (5.18 GHz), simulated (5.24 GHz).
Broadside radiation patterns are observed for both planes.
Co-polarized fields > cross-polarized fields by more than 20 dB in
the boresight direction.
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
0oo
o o
o
o o
o o
o o
o
dB
9090
150
180
o
o
o
o
o
dB
-40 -30 -20 -10 0
30
150
60
90
120
60
150
300 oo o
o o
o o
o o
o o
o
dB
(b)
120
180o180o
Simulation
H-plane (y-z plane)
Measurement
E-plane (x-z plane)
-x +x -y +y
Co-polCo-polCross-pol Cross-pol
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73
TE111y mode: Maximum gain of 4.02 dBi at 3.48 GHz.
TE113y mode: Maximum gain of 7.52 dBi at 5.13 GHz.
Electrically larger antenna has a higher antenna gain.
Measured antenna gain
3 3.5 4 4.5 5 5.5 6-5
0
5
10Gain (dBi)
Frequency (GHz)
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74
B. Example for Wideband DRA Design
a = 30.7 mm, b = 24.7 mm, and d= 47.7 mm.
Given:f1 = 1.98 GHz (PCS)
f2 = 2.48 GHz (WLAN), r=10
Using formulas fordual-mode
rectangular DRA
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75
Configuration of the wideband DRA
b
Conducting
feeding strip
Coaxial
aperture
a
dW
l
Ground plane
Rectangular
DRArx
y
z
l= 17 mm, W= 1 mm
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76
Measured and simulated reflection coefficients
Measured bandwidths : 30.9% (1.83-2.50 GHz)
PCS (1.85-1.99 GHz), UMTS (1.99-2.20 GHz)
& WLAN (2.4-2.48 GHz)
1.5 2 2.5 3-40
-30
-20
-10
0
HFSS Simulation
Measurement
Reflection coefficient |S | (dB)11
Frequency (GHz)
Measured and simulated radiation patterns
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77
Measured and simulated radiation patterns
Measured (2.16 GHz), simulated (2.11 GHz).
Broadside radiation patterns are observed.
Co-polarized fields > cross-polarized fields
by more than 20 dB in the boresight direction.
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
0o
o
o o
o
o o
o o
o o
o
dB
9090
150
180
o
o
o
o
o
dB
-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
0o
o o
o o
o o
o o
o o
o
dB
120
Simulation
H-plane (y-z plane)
Measurement
E-plane (x-z plane)
-x +x -y +y
Co-pol Co-pol
Cross-pol Cross-pol
(a)
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78
Measured and simulated radiation patterns
Measured (2.41 GHz), simulated (2.46 GHz).
Broadside radiation patterns are observed.
Co-polarized fields > cross-polarized fields by more
than 20 dB in the boresight direction.
-40 -30 -20 -10 0
30
-150
60
-120
90-90
120
-60
150
-30
180
0o
o
o o
o
o o
o o
o o
o
dB
90-90
150
180
o
o
o
o
o
dB
-40 -30 -20 -10 0
30
-150
60
90
120
-60
150
-30
180
0o
o o
o o
o o
o o
o o
o
dB
H-plane (y-z plane)
Measurement
E-plane (x-z plane)
Simulation
Co-pol Co-polCross-pol Cross-pol
(b)
-x +x -y +y
M d t i
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79
Measured antenna gain
The maximum gain of 6.98 dBi at 2.47GHz.
TE113y -mode gain > TE111
y -mode gain.
1.6 1.8 2 2.2 2.4 2.6 2.80
2
4
6
8
Frequency (GHz)
Gain (dBi)
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(ii) Cylindrical DRA
R f f th HEM d f th li d i l DRA
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81
2
0
22
irzii kkk Ground plane
z
a
Cylindrical
DRA
h
r
ki & kzi :dielectric wavenumbers along the &zdirections
k0i = 2fi/c : wavenumber in air
(1)
Resonance frequency of the HEMmnrmode of the cylindrical DRA
i = 1, 2 forf1,f2
f1 : HEM111 mode frequencyf2 : HEM113 mode frequency
R f f h HEM d f h li d i l DRA
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82
z
x
y
r
a
Infinitedielectric rod
Resonance frequency of the HEMmnrmode of the cylindrical DRA
Fork:
220)1(' iiri kkk
where
is the radial wavenumber outside the
dielectric rod
Jm(x) : Bessel function of the first kind
Km(x): modified Bessel function of the second kind.
(2)
(3)
D. Kajfez and P. Guillon, Dielectric resonators, Norwood, MA, Artech House, Inc., 1986.
24
22222
)'(
)')('(
'
''
'
1
)(
'
'
''
'
1
)(
'1
akk
kkkkm
akK
akK
kakJ
akJ
kakK
akK
kakJ
akJ
k
ii
iriii
im
im
iim
im
i
r
im
im
iim
im
i
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83
h
z
r
Infinite dielectric
slab
zi
ziirr
i
zi
k
kk
p
hk22
01 )1(tan
Forkz: approximated by the
TM01-mode wavenumber
Resonance frequency of cylindrical DRA
(i = 1, 2 forf1,f2)
wherep1 = 1 andp2 = 3
correspond to the HEM111 andHEM113 modes, respectively.
(4)
R. K. Mongia and P. Bhartia, Dielectric resonator antennas- a review and general design relations for
resonant frequency bandwidth,International Journal of Microwave and Millimeter-Wave Computer-
Aided Engineering, vol. 4, no. 3, pp 230-247, 1994.
Design formula of ratio h/a for given f f and
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84
4
1
44
1
2
1
i
i
i
f
fBi
i
rr
S D
Ce
AE
a
hi
0
0
0
4444
3333
2222
1111
DCBA
DCBA
DCBA
EDCBA s
=
0996.1982.3162.123.19
02.160713.4511.115.36
07.3682402.42.6253.680
11650034800937.0234.07.489
(1)
f1 : HEM111 mode frequency (lower band)
f2 : HEM113 mode frequency (upper band)Ground plane
z
a
Cylindrical
DRA
h
r
Design formula of ratio h/afor given f1, f2, and r
Using the covariance matrix adaptation
evolutionary strategy again,
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85
i
i
a
hBi
ii
rr
S D
Ce
AE
fa
i
4
1
44
1r
1
2
c
0
0
0
4444
3333
2222
1111
DCBA
DCBA
DCBA
EDCBA s
0057.0114.6659.5429.4
0814.179764.00368.0152.0
01.3049973.0005.00571.0
109328.310700152.0751.1109.1
=
(2)
Design formula of radius a
Radius a can be found by inserting h/a into (2) below:
Aftera isfound, h can be determined from h/a.
Maximum error ofa: 2.1% for 1 h/a 3.5, 9 r 27
Maximum error ofh: 3.0% for 1.28 h/a 1.85, 9 r 27
A f i i A i
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86
A. Example for dualband cylindrical DRA design
a = 17.9 mm & h = 42.5 mm
Given:f1 = 1.71 GHz (DCS:1.71- 1.88 GHz)
f2 = 2.4 GHz (WLAN:2.4 - 2.48 GHz),
r=9.4
Using formulas
(1) & (2)
C fi i f h d lb d LP DRA
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87
Matching slotExcitation
stripVia
x
y
Cylindrical
DRAra
Aperture
Feedline
Ground plane
Wf
WsDsLs
a
h
x
z Cylindrical
DRAr
Matching slotVia
Feedline
d
Ground planeAperture
for via
Excitation strip
w
l
Configuration of the dualband LP DRA
Top view Side view
a = 18.7 mm, h = 42.5 mm, r= 9.4, l= 12.5 mm, w = 1 mm,
Ls = 20 mm, Ws = 1.5 mm, andDs = 12.75 mm.
Radius a has been slightly increased to reduce the merging effect
M d d Si l t d R fl ti ffi i t
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88
1.6 1.8 2 2.2 2.4 2.6-30
-20
-10
0
Frequency (GHz)
HFSS Simulation
Measurement
Reflection Coefficient |S11| (dB)
Measured and Simulated Reflection coefficients
Reasonable agreement
Lower band impedance bandwidth: 15.5% (1.70-2.00 GHz)
Upper band impedance bandwidth: 3.7% (2.39-2.48 GHz)
M d d i l t d di ti tt
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89
oo
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o o
o o
o
dB
9090
150
180
o
o
o
o
dB
-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
o o
o o
o
o o
o o
o
dB
120
Col-pol
Cross-polCross-pol
Simulation
H-plane (y-z plane)
Measurement
E-plane (x-z plane)
-x +x -y +y
0o 0o
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o o
o o
o
dB
9090
150
180
o
o
o
o
o
dB
-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
o o
o o
o o
o o
o o
o
dB
120
Col-pol
Cross-pol
Cross-pol
Simulation
H-plane (y-z plane)
Measurement
E-plane (x-z plane)
-x +x -y +y
0o 0o
Measured and simulated radiation patterns
HEM111 mode: measured (1.8 GHz), simulated (1.8 GHz)
HEM113 mode: measured (2.42 GHz), simulated (2.45 GHz)
(a) (b)
Broadside radiation patterns are observed.
Co-polarized fields > cross-polarized fields by more than 20 dB in the
boresight direction.
M d d i l t d i
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90
1.6 1.65 1.7 1.75 1.8 1.85 1.90
2
4
6
Frequency (GHz)
Lower band gain (dBi)
HFSS Simulation
Measurement
Frequency (GHz)
Upper band gain (dBi)
2.3 2.35 2.4 2.45 2.5 2.550
2
4
6
8
1010
8
6
42
02.3 2.35 2.52.4 2.45 2.55
Measured and simulated gain
HEM111 mode: Maximum measured gain of ~6 dBi (1.75 GHz)
HEM113 mode: Maximum measured gain of ~ 8 dBi (2.43 GHz)
D lb d CP DRA
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91
L2
L3
L1W2
W0
WS
W3 W1
LS
DS
Input port
Via
Cylindrical
DRAr
x
ya
quadrature
coupler
Dualband
grounding
via
To
planeGround
portIsolation
slotMatching
stripExcitation
viaTo grounding
Dualband CP DRA
a = 18.7 mm, h = 42.5 mm, r= 9.4, l= 12.5 mm, w = 1 mm,Ls = 21 mm, Ws = 1.5 mm,Ds =
12.75 mm, L1 = 26.9 mm,L2 =26.5 mm,L3 = 56.65 mm, W0 = 4.66 mm, W1 = 7.3 mm, W2 =
4.44 mm, and W3 = 0.46 mm.
a
h
x
z Cylindrical
DRAr
Matching slotVia
Feedline
d
Ground planeAperturefor via
Excitation strip
w
l
Top view Side view
M d d i l t d fl ti ffi i t
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1.6 1.8 2 2.2 2.4 2.6-40
-30
-20
-10
0
Reflection Coefficient |S11| (dB)
Frequency (GHz)
HFSS Simulation
Measurement
Reasonable agreement
Lower band bandwidth:18.9% (1.58-1.91 GHz).
Upper band bandwidth:7.8% (2.33-2.52 GHz).
Measured and simulated reflection coefficients
M d d i l t d i l ti (AR )
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93
1.6 1.65 1.7 1.75 1.8 1.85 1.90
2
4
6
8
10 Lower band AR (dB)
Frequency (GHz)
HFSS Simulation
Measurement
Upper band AR (dB)
Frequency (GHz)
2.3 2.4 2.50
2
4
6
88
6
4
2
02.3 2.4 2.5
Measured and simulated axial ratios (ARs)
Reasonable agreement
Lower band AR bandwidth: 12.4% (1.67-1.89 GHz)
Upper band AR bandwidth: 7.4% (2.34-2.52GHz)
M d d i l t d di ti tt
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94
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o
o
o o
o
dB
9090
150
180
o
o
o
o
o
dB
-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
o o
o o
o o
o o
o o
o
dB
120
LHCP
RHCP RHCP
Simulation
H-plane (y-z plane)
Measurement
E-plane (x-z plane)
-x +x -y +y
0o 0o
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o
o
o o
o
dB
9090
150
180
o
o
o
o
o
dB
-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
o o
o o
o o
oo
o o
o
dB
120
RHCP
LHCP
RHCP
Simulation
H-plane (y-z plane)
Measurement
E-plane (x-z plane)
-x +x -y +y
0o 0o
Measured and simulated radiation patterns
(a) (b)
HEM111 mode: measured (1.8 GHz), simulated (1.8 GHz)
HEM113 mode: measured (2.42 GHz), simulated (2.45 GHz)
Broadside radiation patterns are observed.
LHCP fields > RHCP fields by ~20 dB in the boresight direction.
B Example for wideband cylidnrical DRA design
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95
B. Example for wideband cylidnrical DRA design
a = 10.3 mm & h = 34.3 mm
Given:f1 = 2.90 GHz,f2 = 3.72 GHz, r= 9.4
Using formula(5) & (6)
Wideband LP cylindrical DRA
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96
CylindricalDRAr
Conducting
feeding stripCoaxial
aperture
l
w h
a
Ground plane
z
x
y
2.8 3 3.2 3.4 3.6 3.8 4 4.2-40
-30
-20
-10
0Reflection Coefficient |S11| (dB)
Frequency (GHz)
HFSS Simulation
Measurement
Configuration Reflection coefficient
a = 10.3 mm, h = 34.3 mm, r= 9.4,
l= 12 mm, and w = 1 mm.
Good agreement
Measured impedance bandwidth:
23.5% (3-3.8 GHz)
Wideband LP cylindrical DRA
Measured and simulated gain
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97
3 3.2 3.4 3.6 3.8 40
2
4
6
8
10
12
Frequency (GHz)
Antenna Gain (dBi)
HFSS Simulation
Measurement
Measured and simulated gain
HEM111 mode: Maximum measured gain of ~7 dBi (3.29 GHz)
HEM113 mode: Maximum measured gain of ~10 dBi (3.83 GHz)
Wideband CP cylindrical DRA
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98
L1
L1W0
W0
W1
Input port
Excitation
strip
Isolation
port
Ground
plane
via
Excitation strip
Cylindrical
DRAr
Widebandquadraturecoupler
via
x
ya
a
h
x
z Cylindrical
DRAr
Via
d
Ground planeAperturefor via
Excitation strip
w
l
Wideband quadrature coupler
a = 10.3 mm, h = 34.3 mm, r= 9.4, l= 11.5 mm, w = 1 mm,
L1 = 14.67 mm, W0 = 1.94 mm, and W1 = 3.21 mm.
Wideband CP cylindrical DRA
Top view Side view
Wideband CP DRA
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99
3 3.2 3.4 3.6 3.8 4
-30
-20
-10
0
Frequency (GHz)
Reflection Coefficient |S11| (dB)
HFSS Simulation
Measurement
3 3.2 3.4 3.6 3.8 40
2
4
6
8
Fre uenc (GHz
Axial ratio (dB)
HFSS Simulation
Measurement
Measured 3-dB AR bandwidth :
24.7% (3.05-3.91 GHz).
Measured impedance bandwidth:
25.5% (3.04-3.93 GHz).
Wideband CP DRA
Reflection coefficient Axial ratio
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VI. Dualfunction DRAs
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101
AdvantageSystem size and cost can be reduced by
using dualfunction DRAs.
Additional functions
- Packaging cover- Oscillator
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102
Packaging Cover
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103
Conventional
Aperture Metallic supports
for grounding
Dielectric resonatorantenna/oscillatorload and packagingcover
Metal ground
Power supply
z
y
Front view
hH
Microstrip
feedline
d
Transistor (other
RF components
not shown)
Proposal
Antenna Configuration
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104
Antenna Configuration
Aperture
Feedline and the
RF/MIC circuits
Metal ground
Coaxial line
Dielectric resonator antenna
and packaging coverMetallic supportsfor grounding
H
hy
z
Side view
Coaxial
x
yMicrostrip
feedline
Aperture
L
W
a
b
Top view
Resonant frequencyf0 = 2.4GHz
Parameters:
Hollow DRA:
L=30mm, W=29mm,
H=15mm, & r=12
Metallic Cavity:
a = 15mm, b = 21.6mm, h = 5mmTop face : Duroid r=2.94
thickness 0.762mm
Aperture: 0.2063 e
Design Procedure (Simulation):
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105
Design Procedure (Simulation):
Step 2Remove the lower center
portion concentrically to
form a notched DRA.
As a result, the resonant
frequency >2.4GHz
Step 3Cover the two sides with thesame material. Move the
frequency back to 2.4GHz
by increasing the thickness.
(thickness f0 )
Step 1Use the DWM to design
a solid rectangular DRA
at 2.4-GHz fundamental
TE111 Mode.
x
z
Experimental Verification:
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106
Experimental Verification:
- Hard-clad foam (r1) is used to form the
container.
- ECCOSTOCK HiK Powder ofr=12 is used as
the dielectric material.
Return Loss and Input Impedance
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2 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3
Frequency (GHz)
-60
-50
-40
-30
-20
-10
0
Return
Loss(dB)
Theory
Experiment
2 2.1 2.2 2 .3 2.4 2 .5 2.6 2 .7 2.8 2 .9 3
Frequency (GHz)
-60
-40
-20
0
20
40
60
80
InputImpedance(ohm)
(Passive hollow RDRA with a metallic cavity)
Good agreement.
Bandwidth ~ 5.6%.
Measured resonance frequency: 2.42GHz (error < 0.83%)
Radiation Patterns
(P i h ll DRA ith t lli it )
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-40 -30 -20 -10 0
30
-150
60
-120
90-90
120
-60
150
-30
180
0
co-pol
cross-pol
-40 -30 -20 -10 0
30
-150
60
-120
90-90
120
-60
150
-30
180
0
cross-pol
co-pol
Theory
Experiment
(a) (b)
x-z plane y-z plane
(Passive hollow DRA with a metallic cavity)
Broadside TE111y
mode is observed.
Co-polarized fields generally stronger than the cross-
polarized fields by 20dBin the boresight direction.
Return Loss of the Active Integrated Antenna
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2 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3
Frequency (GHz)
-40
-30
-20
-10
0
ReturnLoss(dB)
A A
DRA DRA
ReturnLoss
ReturnLoss
Receiver Transmitter
DRA (passive)
DRA (active, receiver)
DRA (active, transmitter)
Return Loss of the Active Integrated Antenna
Integrated with Agilent AG302-86 low noise amplifier (LNA)(gain of 13.6dB at 2.4GHz)
LNA prematched to 50 at the input.
A small hole is drilled on the ground plane to supply the DC bias to the LNA.
Amplified Radiation Pattern
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-180 -120 -60 0 60 120 180
Observation angle (degree)
-80
-70
-60
-50
-40
-30
-20
-10
0
Amp
litude(dB)
co-pol (passive)
co-pol (active)
cross-pol (passive)
cross-pol (active)
Amplified Radiation Pattern
Compared to the passive DRA,the active DRA has a gain of
7 - 12dB across the observation angle from -90o to 90o.
The gain is less than the specification due to unavoidable
impedance variations and imperfections in the measurement.
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Dielectric Resonator Antenna
Oscillator (DRAO)
Methodology
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The DRA is used as the oscillator load,
named as DRAO.
Methodology
The reflection amplifier method is used to
design the antenna oscillator.
DRAO Schematic Diagram
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DRAjXZin=Rin+X in
Lm
Transistor
C
ZL=RL+XL
DRAO Schematic Diagram
- Oscillate condition:XL+Xin=0 &RL
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L
w
Top view
Microstrip
feedlineAperture
y
x
Lm Ls
La
Wa
Transistor (other
RF components
not shown)
fo = 1.85GHz at TE111y
Parameters:DRA
L=52.2mm,
W=42.4mm,
H=26.1mm,
r= 6.
Aperture
La = 0.3561e, Wa = 2mm
Ls = 9.5 mm, Lm = 40 mm.
Duroid substrate
rs=2.94, d=0.762mm
antenna and
oscillating load
Microstrip
feedline Aperture
Ground
L
H
z
x
Substrate
Side view
d
Transistor (other
RF components
not shown)
r
Return Loss and Input Impedance
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Return Loss and Input Impedance
Good agreement.
Bandwidth ~ 22.14%.
Resonance frequency: Measured 1.86GHz
Simulated 1.83GHz (1.5% error).
1.6 1.7 1.8 1.9 2 2.1 2.2-60
-50
-40
-30
-20
-10
0
Frequency (GHz)
Return Loss (dB)
HFSS Simulation
Experiment1.6 1.7 1.8 1.9 2 2.1
-80
-60
-40
-20
0
20
40
60
80
Frequency (GHz)
Input Impedance (ohms)
Spectrum of the Free-running DRAO
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p g
Transmitting powerPt= 16.4dBm
DC-RF efficiency: ~ 13% (2-25% in the literature).
Phase noise: 103dBc/Hz at 5MHz offset
Second harmonic < fundamental by 22dB
Radiation Pattern
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Broadside TE111y
is observed.
Co-polarized fields are generally 20dB stronger
than the cross-polarized fields in the boresight direction.
Solid DRAO (measur.)
Passive Solid DRA (measur.)
HFSS Simulation
-40 -30 -20 -10 0
30
60
9090
120
150
-30
180
0
-60
-120
-150
co-pol
cross-pol
x-z plane
(a)
-40 -30 - 20 -10 0
30
60
90-90
120
150
-30
180
0
-60
-120
-150
y-z plane
(b)
co-pol
cross-pol
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DRA can be of any shape. Can it be made like a swan?
Yes!
DRA is simple made of dielectric. Can glass be used forthe dielectric?
It leads to probably the most beautiful antenna in
the world .
Yes!
Glass-Swan DRA
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Glass Swan DRA
Distinguished Lecture
Transparent antennas: From 2D to 3D119
Conclusion
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The DRA can be easily excited with various excitation schemes.
Frequency tuning of the DRA can be achieved by using
a loading-disk or parasitic slot.
The dualband and wideband DRAs can be easily designed using
higher-order modes.
Compact omnidirectional CP DRAs have been presented
Dualfuncton DRAs for packaging and oscillator designs have
been demonstrated.
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Q & A