Design of a High-Frequency Planar Power

8/4/2019 Design of a High-Frequency Planar Power

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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 38, N O. 2 , APRIL 1991 135

Design of a High-Frequency Planar PowerTransformer in Multilayer TechnologyDirk van der Linde, Corlex A . M . Boon, and J . Ben Klaassens

Abstract-A high-frequency power transformer in multilayer printedcircuit board (ML-PCB) technology is presented for applications inswitched-mode power supplies operating at frequencies up to severalmegahertz.The mechanical configuration of laboratory prototypes is discussed,as well a s the electrical, parasitic, and thermal behavior. The presenta-tion is focused on the leakage inductance since the analysis of otheraspects is relatively simple. Test results show high efficiency, low leakageinductance, good thermal behavior, and line insulation properties of thetransformer. Further, the topology enables the designer to make atradeoff between leakage inductance and interwinding capacitance. Dueto the well-defined geometry, parasitic interwinding capacitance andleakage inductance are reproducable and can be computed relativelyeasily.

I. INTRODUCTIONN THE NEW generation of power converters, miniaturizationI as become an important rule of design [11, [2], [5], [6], [111.

Modem semiconductors allow fast switching and can be used toincrease switching frequencies up to the megahertz region.Consequently, capacitive and magnetic components can be re-duced in weight and size. This miniaturization, however, givesrise to some specific problems:0 The parasitic components set limitations to the hf cut-off

The parasitic behavior of conventional transformers is notSufficient cooling of compact power devices is often a

frequency.reproducable.problem.

Parasitic components play an important role in circuit behav-ior due to the high switching frequencies. For good hf proper-ties, the leakage inductance and the interwinding capacitancehave to be small; they both limit the hf cut-off frequency [4].Energy stored in the parasitic leakage inductance may result inhigh-voltage peaks during switching of the vulnerable switchingdevices. These peaks cause dynamic power loss and excessivestress on components. A small leakage inductance can beachieved by a good inductive coupling between the primary andsecondary. In practice, the windings are interleaved with smalldistances between the windings. This leads, however, to a highinterwinding capacity. The requirements for a low leakage in-ductance and a low interwinding capacitance are contradictory.Therefore, the LC product of a transformer can be used as afigure of trouble.These parasitic effects and the related hf properties in conven-tionally wound transformers appear to be uncontrollable [7]. The

Manuscript received February 20, 1990; revised December 1990.D. van der Linde and C. A . M. Boon are with Hollandse SignaalapparatenJ . B . Klaassens is with Delft University of Technology, Delft, theIEEE Log Number 9142807.

B.V., the Hague, the Netherlands.Netherlands.

variables in the manufacturing process cause considerable toler-ances in the winding geometry. A high degree of reproducibilityis fully related to a strictly defined winding geometry. Only ifreproduction can be guaranteed, an attempt is useful to calculateparasitic components.A compact converter module has a relatively small surfacearea for the conveyance of internally dissipated heat. To keephot-spot temperature rise under control, a high converter effi-ciency is required. Additionally, a flat package provides thelargest surface area and, therefore, the best heat transfer to theenvironment. For this reason, there is a tendency towards con-verter modules in flat-pack housing. The shape of a conventionaltransformer, however, is not very suitable for use in flat-packmodules.In order to approach the problems mentioned above, the ideawas to design a transformer with its windings integrated into amultilayer printed circuit board (ML-PCB). Ferrite core-halveson either side of the multilayer winding package would completethe magnetic circuit (without air gap). This configuration wouldprovide the following advantages. The winding geometry and itsrelated parasitic behavior are defined within the (small) toler-ances of PCB manufacturing and are therefore reproducable.Further, the entire transformer can be flat (planar transformer)since the windings consist of thin copper layers. These thincopper layers reduce skin effect losses. The flat configurationprovides a relatively large surface area for the transfer ofdissipated heat to the environment. The manufacturing processcan be fully automated, although multilayer manufacturing re-quires dedicated facilities and skills. The ML-PCB transformercan be an integrated part of a circuit board with other compo-nents.In the following, the configuration of the ML-PCB trans-former will be discussed. Then, aspects such as parasitics, lineinsulation, and thermal behavior will be discussed on the basis ofprototypes.

11. MULTILAYERRANSFORMERA . Windings

A multilayer is composed of several double-sided printedcircuit boards (bilayers) pressed and cemented together withepoxy resin. Each bilayer consists of a standard epoxy base-layerof 76 pm covered on both sides with a 60-pm copper layer. Theepoxy resin layer between the bilayers measures 200 pm. In anML-PCB, several distinct copper layers are available for creat-ing the transformer windings. To connect individual turns inseries or in parallel, interconnections between different layershave to be made by via holes . The copper layers intersectedby the same via hole are electrically tied together.Fig. 1 shows the interconnection of individual turns in seriesusing via holes.As shown, each copper layer contains one single turn with0278-0046/91/0400-0135$01.00 0 991 IEEE


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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 38 , NO . 2, APRIL 199136

Fig. 1 . Winding arrangement primary. Fig. 3. Winding arrangement secondary.

Fig. 2. Alternative winding arrangement primary.

two connection flaps. Flaps to be interconnected are placedsuch that they can be intersected by the same via hole. Theamount of turns that can be placed in series is limited by theavailable space for the connection flaps and is determined by thecore type.A reduction in the number of flaps needed is achieved using aslightly different approach, as is depicted in Fig. 2. In thisconfiguration, the two turns on each side of a bilayer are placedin series by a local via hole or buried hole. This requiresthe special treatment of making via holes in each individualbilayer before multilayer assembly, whereas in the previousconfiguration, all via holes are made in one run after multilayerassembly. Although more expensive, the limited amount ofspace for the connection flaps may leave no alternative.On the low side of the transformer, turns are connected inparallel. This can be realized very easily with the standard viaholes, as is indicated in Fig. 3.The epoxy layers occupy a considerable part of the trans-formers winding space; the ML-PCB prototypes have an effec-tive window utilization of 0.3. On the other hand, one shouldrealize that in conventional transformers, for hf power applica-tions, litz wire is mandatory to eliminate the skin effect, result-ing in a comparable window utilization.To study the transformers parasitic effects, three types of theML-PCB transformer were made. Each version differs from theothers in winding configuration; thus, each has a different leak-age inductance and intenvinding capacitance. These versions areexplained with reference to Fig. 4. In type 1, the primary andsecondary windings are split up in two separate groups. In type2, the primary winding is sandwiched between two groups of

I r i m a r yo S e c o n d a r y

t y p e 1 t y p e 2 t y p e 3Fig. 4. Arrangement of windings in prototype.

secondary windings. In type 3-, the primary and secondarywindings are interleaved.B. Ferrite Core

The selection of a suitable core is another point of discussion.The ML-PCB transformers configuration is flat, whereas thewindow of available standard cores is relatively high. A standardcore can be adapted to the ML-PCB winding package by grind-ing off part of each core half, hence reducing the height of thetransformer core. The core should provide openings large enoughto enter and leave the winding space with the connection flaps.For the prototypes, the RM-14 core (Philips) is selected. Theunmodified RM-14 core parameters are as follows:

Effective cross-sectional area A , = 198 mm2effective magnetic path length I , = 70 mmeffective volume of the core V, = 13 900 mm3mass of the core set me 74 grrelative permeability /L = 2000.The RM-14 encloses a considerable part of the winding spacewhile sufficient space for the interconnection flaps remains. TheRM-14 is also available in Philips new 3F3 femte especiallydesigned for hf power applications.( B , , = 330mT, T, > 200 C ;PloSs 150W/dm3

at 400 kHz,50 mT).The prototype of the ML-PCB transformer was tested inexisting lab model 5-V-25-A converters operating at 1 MHz.This application required a 1:8 transformer turns ratio and acenter tap on the secondary side (see Fig. 3) .The primary winding consists of eight turns in series, using


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VAN DER LI NDE et a l . : DES I G N O F A P LANAR P O W ER TRANS F O RM ER I37

the buried-hole method as described with reference to Fig. 2.For each secondary tap, eight turns are connected in parallel inorder to divide copper losses equally between the primary andsecondary. Twelve bilayers appear to be necessary to obtain the24 distinct copper layers, resulting in a 5-mm-thick ML-PCBwinding package. The original RM-14 core has a window heightof ca. 20 mm, whereas the ML-transformers winding packageis only ca. 5 mm thick. To adapt the core to the ML-PCBtransformer package, the cores window height is reduced bygrinding off part of each core half.The modified RM-14 core has a window that is just highenough to fit the winding package (ca. 5 mm). The cores weightis reduced by 40 gr and the effective window perimeter by 30mm. The result is a flat transformer configuration, and with arelatively flat but wide window, providing sufficient space forthe amount of copper required in power applications. Next, theparasitic behavior of the prototypes will be discussed.

111. PARASITICSND HF-BEHAVIORA . Leakage Inductance

The leakage inductance is a lumped element, representing theparasitic effect of a nonperfect inductive coupling between theprimary and secondary windings. A coupling factor smaller thanone results in a magnetic field within the winding space of thetransformer. The magnetic energy E, store in this field isdirectly related to the leakage inductance L , byE , = iL,12 ( 1 )

where I is the current through the primary winding, and L , isthe total leakage inductance transformed at the primary side.To obtain the value of the leakage inductance, one needs tostudy the magnetic field in the transformers winding space.Once the B components are known, the energy stored in themagnetic field can be obtained simply by volume integration:E, = f p p O j v 2 d V . (2 )

In [8] and [lo], methods describing the magnetic field insidethe winding space of transformers have been presented and haveresulted in simple formulae for the leakage inductance, e.g.,Kapps formula. These methods, however, are known to beimprecise. Several simplifications are the cause of errors up to100% .First, the configuration is described as being rotation symmet-ric. In this case, the originally three-dimensional field analysiscan be reduced to a two-dimensional model, describing a radialintersection of the winding space.Second, a far-too-simple interpretation of Amperes law leadsto the incorrect conclusion that the leakage flux is restricted tothe area between the primary and secondary windings and hasthe same direction and value in every point of that area. In thecase of the ML-PCB transformers, the results, obtained withthese simplifications, showed considerable differences comparedwith measured values. Clearly, this approach is insufficient for asatisfactory description of the ML-transformers behavior.In a second attempt to compute the leakage inductance, theconfiguration was still described as rotation symmetric, but here,precise field calculations were made. In general, a magnetic fieldcan be described by means of a magnetic potential vector A . Inthe case of a (quasi-) two-dimensional configuration

(3 )

b

a; 4Cross section of the winding area by Roth.ig. 5.

where J is the current density at point ( x ,y ) . Within thewinding space, J = 0 (outside the copper) or J # 0 (inside theThe components B , and B y of the magnetic field can becopper).derived from the magnetic potential A :

aA aAB = - and B = - - .a y Y ax (4)A diversity of software packages is available solving theseequations by finite element methods. Examined packages eitherrequired extensive hardware facilities (mainframes) or gaveunsatisfactory results. Therefore, we tried to employ calculationmethods known from literature.Roth [121 presents analytical solutions to the magnetic fieldcomponents in quasi two-dimensional structures. Roth describesa radial intersection of the winding space enclosed by ferrite,containing rectangular conductors as the windings. Since themultilayer transformers windings consist of rectangular copperlayers, Roths method for describing the magnetic field is verysuitable in this case. Fig. 5shows a radial intersection of anarbitrary transformer configuration.Roth describes the magnetic potential A according to thegeometrical parameters as given in Fig. 5 :

m mA = i = l k = l ;, cos ( m i x ) os ( n , y )

wheremi = ( i - l ) a / an, = ( k - l ) a / b

16TlOabA ; , =-. 5 Ij[sin(mia>) - sin(miaj ) ] [s in(nkbj - sin(nkbj)]

j = ( a ) - a j ) b; - b j )m i n k (mf n i )and where q is the number of individual conductors in thewinding space, and I j is the current in conductor j .Now that the potential vector has been determined, the com-ponents of the magnetic field can be obtained as indicated informula (4). From the magnetic field components, the leakageinductance can be computed (formulae ( 1 ) and (2)). Resultsobtained by Roths method are verified by a two-dimensionalfinite element analysis using suitable software. Compared withthe outcomes of Kapps formula, Roths results match the resultsof measurements on prototypes that are considerably better. Theremaining errors are attributed to the inaccuracy of describingthe configuration as rotation symmetric. Improvement could be

1

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138 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 38, NO. 2, APRIL 1991

S e c o n d a r y

Fig. 6. Cross-section winding area.

(C)Fig. 7. Results of computer simulation of the magnetic field lines in thewinding area: (a) Expression following Roth; (b) finite element calculation;(c) first-order approximation.made by using an accurate three-dimensional (finite-element)analysis, which is, of course, more complicated.To illustrate the basic magnetic field inside the multilayertransformer, a configuration of one primary and one secondaryturn is discussed. The winding space is surrounded by high-per-meability ferrite. The radial intersection of the winding space isgiven in Fig. 6.The formulae according to Roths theory are implemented in astraightforward Pascal program for a personal computer (withcoprocessor). The intersection is divided into an n x m grid. Atevery single grid point, the magnetic-field components are com-puted. The results are displayed as line elements in the windingspace in Fig. 7(a). A line element shows the direction of themagnetic field at that point of the winding space, just like acompass needle. The line elements do not provide direct infor-mation about the absolute value of the magnetic field.As mentioned, results were verified with finite elementsoftware. Fig. 7(b) shows the magnetic lines of force accordingto the MAGGIE software package (Philips). These lines alsoprovide information related to the direction of the magnetic fieldat each point of the line but do not give direct information aboutthe absolute value of the magnetic field. In Fig. 7(c), the lineelements are shown according to the approximation leading toKapps formula. As shown, the approximations main errors

occur around the windings edges and in the space between thewindings and the ferrite.B . Magnetizing Inductanceters and by the number of primary turns:The magnetizing inductance is determined by the core parame-

wherepo permeability of airp ,A , effective core areale effective window perimeter.

relative permeability of the ferrite core

The modified RM-14 core, obtained by grinding off 15 mm ofits window height, has an effective window perimeter approxi-mately 2 x 15 = 30 mm smaller than the unmodified core.Hence, in this casep r = 2000 (3F3) I, = 40 mm

N = 8, = 198 mm2L e = 633pH (measured: 640 p H )

C. Interwinding CapacitanceThe parasitic capacitance Cp sbetween the primary and sec-ondary windings strongly affects the transformers hf properties.It is desirable to keep Cpssmall. In the case of the ML-PCBtransformer, Cps can be computed easily since the windingsconsist of parallel, flat conductors. The capacitance between twowindings can be found simply by using the formula for thecapacitance between two parallel conductive plates. Table Igives the results with respect to the ML-PCB transformer proto-types. Clearly, the capacitance between the windings is notparticularly small. In type 3, the Cps of 589 pF will beunacceptable in most cases. With the configuration of flat con-ductors at close distances in combination with the epoxys

E, = 7, one could hardly expect better results; it is the price onehas to pay for the very low leakage inductance. As mentionedbefore, the requirements for a low leakage inductance and a lowinterwinding capacity are contradictory. Nevertheless, an ex-change between the leakage inductance and interwinding capac-ity can be achieved simply by changing the winding configura-tion. The overall hf-properties are determined by the product ofL , and Cps.Table I1 shows the parasitics and LC product of a prototype ofthe planar transformer. This transformer with the winding con-figuration according to type 2 has the lowest figure of troubleand is the best approximation of the ideal transformer. Tocomplete the picture, in Table 11, this ML-PCB transformer iscompared with a conventional 1:8 transformer on an RM-10core designed for similar applications and with an extrapolationof the transformer designed by Estrov [ 2 ] .According to Table 11, the ML-PCB transformer has a verylow leakage inductance and a relatively high magnetizing induc-tance. In this case, the leakage inductance drops under 0.1 % ofthe magnetizing inductance. On the other hand, the ML-PCBtransformer has a relatively high interwinding capacitance. Thiscan be attributed to the use of flat windings at close interwindingdistances in combination with the epoxys E, = 7. Nevertheless,


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140 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS. VOL. 38 , NO. 2, APRIL 1991

.... .

0 P' J0 c.in

copper

L1 E1 Fe P/ 2______..-U 0( f=+=-jq qq----==(p %I(b )

Fig. 8. Model heat trans fer: (a) Smgle-slded cooling (print-boardmounted); (b ) double-sided coolingVII. CONCLUSIONS

A transformer in multilayer printed circuit-board technologyis presented, and properties are discussed on the basis of proto-types. The configuration has a low leakage inductance and a highmagnetizing inductance; the leakage inductance is less than0.1% of the magnetizing inductance. Although the interwindingcapacitance is relatively high, the LC product remains low.Further, the designer can trade leakage inductance against inter-winding capacitance in a predictable way by changing the ar-rangement of the windings. The use of a multilayer windingpackage provides enough copper for power applications up to200 W for the modified RM-14 core. Fig. 9 shows an experi-mental transformer that has been tested in a boost converteroperating at 1 MHz.

Fig. 9. Experimental transformer.The prototypes of this compact transformer showed good lineinsulation properties and thermal behavior. Only the tolerance inthe epoxy resin layers causes tolerances of 10% in the parasiticleakage inductance and interwinding capacity, setting some limi-tations to the reproducibility. It may be worthwhile to investigatepossibilities to reduce these tolerances in the production process.


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TABLE I

139

L S C P Scalc. meas.S Le CP SType meas. calc. meas. calc. meas.1 1050 nH 975 nH 640 pH 633 pH 117 pF 119 pF 1.23 S2 450 nH 365 nH 164 pF 159 pF 0.74 s23 285 nH 130 nH 589 pF 517 pF 1.68 s2

TABLE I1transformer Estrov design PlanarRM-10 (1986) transformerR p r i m ,DC 10.4 m fl 120 mO 41 mORS , , , DC 0.9 mO 1.25 m il 0.7 mOLS 1800 nH 2200 nH 450 nH50 pF 75 pF 164 pFLe 300 pH 445 WH 640 pHCP SI S C D S 90 s2 165 fs2 74 fs2

the LC product remains low due to the low leakage inductance.The ML-PCB configuration seems very suitable for hf powerapplications where a low leakage inductance has priority over alow interwinding capacity.IV. HIGH-VOLTAGENSULATION

If an off-line power supply needs to provide line insulation,the transformer will have to meet certain insulation require-ments. In the ML-PCB transformer, all copper layers are indi-vidually surrounded by epoxy (base-layer) and epoxy resin(between the bilayers). For this, requirements are within thecategory of distance through insulation, and the requirementsfor creepage and clearance are simply not applicable.In the prototypes, primary and secondary windings are sepa-rated by a 200-pm epoxy resin layer. According to manufactur-ers specifications, the resin can handle 30 V/pm, hence provid-ing insulation up to 6 kV. After prototypes were exposed to a90% humidity at 40C during a five-day period, the breakdownvoltage between primary and secondary still exceeded 4 kV,whereas breakdown between windings and core occurred at ca.2.5 kV. Although these results are very encouraging for acompact transformer under these circumstances, the insulationbetween windings and core is less than expected. The multi-layers edges probably suffer from very slight damage caused bythe cluttering process. Another explanation could be the pres-ence of small air bubbles in the epoxy resin.

V . THERMAL ANAGEMENTThe main mechanism for the disposal of the dissipated heat isthe thermal conduction in the axial direction, which is perpen-dicular to the layers. Therefore, a worst-case thermal analysiscan be made by assuming that this is the only mechanism forheat transfer. Two situations will be discussed: first, conductionof all dissipated heat to one side of the multilayer and second,conduction to both sides of the multilayer. Both situations can berepresented by an equivalent electrical network of current sourcesrepresenting the heat dissipated in the copper layers and ofresistors representing the thermal resistance of the individuallayers, as is indicated in Fig. 8. The computed node voltages

represent the temperatures in each layer.To compute the temperature distribution in the multilayer, oneneeds the amount of heat dissipated in each layer as well as thethermal resistance of each layer. For the thermal resistance of

the different layers yields60-pm copper layer 0.00034 C/W76-pm epoxy layer 0.67 C/W200-pm resin layer 1.8 C/WFerrite (PCB to heatsink) 2.9 OC/W

Assuming an effective secondary current of 25 A, the secondarywinding resistance of 0.7 mQ per tap and the primary resistanceof 41 mQ result in total copper losses of approximately 0.8 W ata frequency of 1 MHz. With an equal amount of core loss, themaximum temperature rise in the configurations according toFig. 4 are calculated.In the case of heat transfer to one side of the multilayer, theinsulated side of the multilayer will reach 15C above heatsinktemperature. In the case of heat transfer to both sides of themultilayer, the maximum temperature rise occurs in the middleof the multilayer at 5C above heatsink temperature. As shown,temperature rise remains moderate, especially when heat trans-fer can be realized in both directions. In practice, however, itmight be difficult to establish a thermal contact between theferrite and the multilayer on both sides since the thickness of themultilayer is subject to tolerances due to variations in thethickness of the epoxy resin layers.Due to the low dc resistance of the windings, in combinationwith the good heat transfer to the environment, the ML-PCBtransformer can handle high secondary currents. In the labora-tory models of a dc-dc converter the ML-PCB transformerdistributes 5 V-35 A without running hot. For this situation, thetransformers power dissipation is close to 1% of the total outputpower.VI. REPRODUCTION

One of the objectives while choosing the ML-PCB configura-tion was the reproduction of the parasitic effects. Since thelayout of all layers has to remain within small tolerances of PCBmanufacturing, a high degree of reproduction was expected.Only the thickness of the epoxy resin layers is subject to sometolerance. During multilayer assembly, the distinct bilayers arecemented together under pressure. During this process, controlover the thickness of the resin layers is limited. A tolerance of15% can be expected. This tolerance applies to the resin layersonly. The tolerance on the epoxy base layers and the copperlayers can be neglected.This tolerance affects the reproduction of the transformersparasitic behavior since the distances between the windings andtherefore the capacitive and inductive coupling are not preciselydetermined. In the case of the prototypes, the effect on theleakage inductance and interwinding capacity lies within a toler-ance of ca. 10%.The tolerance in the epoxy resin layers also results in atolerance in the thickness of the entire multilayer winding pack-age. For this reason, it will be difficult to establish a thermalcontact between the multilayer and the ferrite on both sidesunless each core is grinded to fit each winding package individu-ally.


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ACKNOWLEDGMENTThe authors wish to thank Hollandse Signaalapparaten B.V.for their permission to publish this material, D. Moezel forindicating the potentials of multilayer technology, C. van Akenfor the layout work, F . A . Ros from the multilayer workshop formanufacturing the prototypes, and Philips-Elcoma for grindingthe ferrite cores.

REFERENCES[ l ] W . C . Bowman et al . , A resonant dc-to-dc converter operatingat 22 MHz, in Third Ann. ZEEE Power Electron. Conf.R e c . , 1988, pp . 3 - 1 1 .121 A. Estrov, Power t ransformer design for 1 MHz resonantconverter, in High Frequency Power Conv. Conf . Rec. ,1986, pp . 36-54.[31 -, Planar magnetics for power converters, ZEEE Trans.Power Electron., vol. 31 , pp . 46-53, 1989.141 W. M. Flanagan, Handb ook of Transformer Applica tions.New York: McG raw-Hil l , 1986.[5 ] A . F. Goldberg, J. G . Kassakian, and M. F. Schlecht , Fini te

element analysis of copper loss in 1-10 MHz transformers, inZEEE Power Electron. Specialists Conf. Rec. (Japan), 1988,-, Issues related to 1-10-MHz transformer design,ZEE ETrans. P ower Electron., vol. 4, pp .113-123, 1989.P. M . Gradzk i and F. C . Lee, Design of high-frequency hybridpower transformer, in Third A n n . ZEEE Power Electron.C o n f . R e c ., 1988, 319-326.B. Hague, Electromagnetic Problems in Electrical Engineer-ing.A. L. Morris, Influence of various factors upon leakage reac-tance of t ransformers,J . ZEE, no . 86 , pp. 485-495, 1940.[ lo] F. E . Terman , Radio Engineers Handbook. New York: Mc-Graw-Hil l , 1943.[ l l ] S . Ohzora and T. Koyashiki, M iniaturization of low-powerconstant-curren t converter by applying a height reduced trans-former, in IEEE Power Electron. Specialist Conf. Rec.E. Roth,Analytical study of the leakage field of transformersand of the mechanical forces exerted on the windings,RevueGenPrale de IElectricitb ,no . 2 3 , pp . 113, 1928.

pp . 1105-1111.161[7 ]

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Design of a High-Frequency Planar Power

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