similar UWB characteristics can always be obtained by usingappropriate values of wa and wb. The VSWR curves shown inFigure 4 are for the case of d � 0.4 mm. When the tuning stub iswidened towards only one side (i.e., wa � 7 mm and wb � 0), theVSWR � 2 impedance band extends over a band of about 3–8GHz. With the other side of the tuning stub subsequently widenedup to wb � 1 mm, the band of 8–16 GHz can also be brought tohave an impedance match with VSWR � 2.

3.2. Radiation PatternsFigure 5 shows at 4, 7, 10, and 12 GHz the xz- and yz-planefar-field radiation patterns of the optimized antenna design for theproposed PSSA, where G � 25, L � 22, t � 11, d � 0.7, wa � 0.7,and wb � 1.5 (all units are in mm). The measured radiationpatterns agree reasonably well with the simulated ones. At thelower frequency (i.e., 4 GHz), the antenna has a nearly omni-directional radiation pattern, as is expected. However, at the fre-quency equal to or greater than 7 GHz, the asymmetry in theprotruded tuning stub has caused unequal copolarized radiations inthe �x and �x directions. This asymmetry in the xz-plane radia-tion pattern might be the reason that the cross-polarization levels inthe xz-plane are larger than the co-polarization ones in some spatialdirections. The maximum antenna gains in the band of 3–12 GHzwere measured to be between 2 and 5 dBi, and were simulated torange from 1.9 to 4.7 dBi, as shown in Figure 6.

4. CONCLUSION

This article has presented a new CPW-fed UWB PSSA, in whichthe protruded tuning stub is asymmetrically widened to its twosides. The antenna geometry is simple; the design procedure issimple; yet a wide VSWR � 2 impedance band of about 2.8–16GHz has been obtained for the fabricated antenna, whose occupiedcircuit area is as small as 25 � 25 mm2. The antenna gains in thestandard UWB were measured to be about 2–5 dBi.

ACKNOWLEDGMENT

This work was supported by the National Science Council ofTaiwan, the Republic China, under Grant NSC 95–2221-E-606–036.

REFERENCES

1. H.D. Chen, J.S. Chen, and J.N. Li, Ultra-wideband square-slot antenna,Microwave Opt Technol Lett 48 (2006), 500–502.

2. E.S. Angelopoulos, A.Z. Anastopoulos, D.I. Kaklamani, A.A. Alexan-dridis, F. Lazarakis, and K. Dangakis, Circular and elliptical CPW-fedslot and microstrip-fed antennas for Ultrawideband applications, IEEEAntennas Wireless Propagat Lett 5 (2006), 294–297.

3. R. Chair, A.A. Kishk, and K.F. Lee, Ultrawide-band coplanarwaveguide-fed rectangular slot antenna, IEEE Antennas WirelessPropagat Lett 3 (2004), 227–229.

4. G. Sorbello, F. Consoli, and S. Barbarino, Numerial and experimentalanalysis of a rectangular slot antenna for UWB communications, Mi-crowave Opt Technol Lett 46 (2005), 315–319.

5. J.Y. Sze and K.L. Wong, Bandwidth enhancement of a microstrip-line-fed printed wide-slot antenna, IEEE Trans Antennas Propagat 49(2001), 1020–1024.

6. J.Y. Chiou, J.Y. Sze, and K.L. Wong, A broad-band CPW-fed strip-loaded square slot antenna, IEEE Trans Antennas Propagat 51 (2003),719–721.

7. H.D. Chen, Broadband CPW-fed square slot antennas with a widenedtuning stub, IEEE Trans Antennas Propagat 51 (2003), 1982–1986.

© 2008 Wiley Periodicals, Inc.

ANALYSIS OF COMPACT H-SHAPEDMICROSTRIP ANTENNA

J. A. Ansari,1 Satya Kesh Dubey,1 Prabhakar Singh,1

R. U. Khan,2 and Babau R. Vishvakarma2

1 Department of Electronics and Communication University ofAllahabad, Allahabad, India2 Department of Electronics Engineering I. T. BHU, Varanasi 221005,India; Corresponding author: brvish@bhu.ac.in

Received 23 November 2007

ABSTRACT: Analysis of H-shaped patch is carried out using equivalentcircuit model. It is found that the antenna exhibits dual resonance behaviorwhich is very sensitive to the dimensions of the notch. The theoretical re-sults of proposed method are compared with experimental and theoreticalresult reported earlier using FDTD method. Comparison with other re-ported results justify the veracity of the proposed method. © 2008 WileyPeriodicals, Inc. Microwave Opt Technol Lett 50: 1779–1784, 2008;Published online in Wiley InterScience (www.interscience.wiley.com).DOI 10.1002/mop.23543

Key words: H-shaped patch; notch patch and dual band microstrip an-tenna

1. INTRODUCTION

With the advent of communication engineering the necessity ofsize reduction and wide bandwidth with the patch radiator areessential to meet the requirement for practical applications. Con-sequently the design of compact microstrip antennas with broad-band/dualband characteristics [1–3] is significant especially insatellite links, wireless local networks, cellular telephones, syn-thetic aperture radars, and radio frequency identification systems.A number of approaches have been reported to obtain compactdual band microstrip antenna such as loading of rectangular [4],circular [5], and triangular patches by shorting pins [6], loading bycrossed slot [7] and the use of a rectangular ring [8]. These dualband operations can be at fixed frequencies or tunable at both orone of the frequency.

In the present article an H-shaped microstrip patch antenna [9]has been analyzed using equivalent circuit model. An H-shaped

Figure 6 Simulated and measured maximum antenna gains of the de-signed PSSA from 3 to 12 GHz

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 50, No. 7, July 2008 1779

microstrip antenna (MSA) is obtained by symmetrically cuttingnotches (d � w) along nonradiating edges of the rectangularmicrostrip antenna as shown in Figure 1. The different antennaparameters such as input impedance, return loss, and resonancefrequency have been studied as a function of frequency for differ-ent value of notch width and depth, the details of which are givenin the following sections.

2. THEORETICAL CONSIDERATIONS

Rectangular microstrip antenna (L � W) is considered as a parallelcombination of resistance (R1), inductance (L1), and capacitance(C1) [Fig. 2(a)], the values of R1, L1, and C1 can be given as [10]

C1 ��0�eLW

2hcos�2��y0

L � (1)

L1 �1

�2C1(2)

R1 �Qr

�C1(3)

in which L � length of the rectangular patch, W � width of therectangular patch, y0 � feed point location along length of thepatch, h � thickness of the substrate material.

Qr �c��e

4fh

where c � velocity of light, f � the design frequency, �e iseffective permittivity of the medium which is given by [10].

�e ��r � 1

2�

�r � 1

2 �1 �10h

W ��1/ 2

where �r � relative permittivity of the substrate material.Because of the notches cut there is change in the surface current

of microstrip patch that changes the resonance behavior of thepatch. This is due to the fact that when the notch is cut along the

nonradiating edges of the rectangular patch the electric and mag-netic field distribution changes due to the lengthening of surfacecurrent around both the notches. Because of the notch an additional

d

w

Center strip length

Central strip width

Side strip length (w1)

Figure 1 H-shaped microstrip antenna

L1 C1R1

∆C

∆C∆L

C1L1

∆L

R1

L2 C2R2

(a)

(b)

(c)

Figure 2 (a) Equivalent circuit of the RMSA, (b) equivalent circuit ofH-shaped RMSA, (c) modified equivalent circuit of H-shaped RMSA

T

Cm

Lm

ZpZH

Cm

Lm

Figure 3 Coupled resonant circuit

1780 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 50, No. 7, July 2008 DOI 10.1002/mop

series inductance (�L) and series capacitance (�C) appear thatmodify the equivalent circuit of the H-shaped patch to Figure 2(c).

The input impedance of the patch can be calculated from Figure2(a) as

ZP �1

1

R1�

1

j�L1� j�C1

(4)

The second resonant circuit is shown in Figure 2(b) in which seriesinductance (�L) and series capacitance (�C) can be calculated as[11, 12]

�L �Z1

16 f cos�2��y0

L � tan��fd

c � (5)

where Z1 is the characteristic impedance of the microstrip line withwidth w1 and is given as

Z1 �120�

w1

h� 1.393 � 0.667 log�w1

h� 1.44�

and

�C � � d

w�Cs (6)

in which Cs � gap capacitance between two side strips [13]The input impedance of the circuit shown in the Figure 2(c) can

be given as

ZH �1

1

R1�

1

j�L2� j�C2

(7)

where

1.5 1.6 1.7 1.8 1.9 2

x 109

-30

-20

-10

0

10

20

30

40

50

Frequency (Hz)

Inpu

t Im

peda

nce

(ohm

)

R,d=5mmX,d=5mmR,d=7mmX,d=7mmR,d=9mmX,d=9mmR,d=12mmX,d=12mm

R

X

(i)

0.8 0.85 0.9 0.95 1 1.05 1.1 1.15 1.2 1.25

x 1010

-30

-20

-10

0

10

20

30

40

50

Frequency (Hz)

Inpu

t Im

peda

nce

(ohm

)

R,d=5 mmX,d=5 mmR,d=7 mmX,d=7 mmR,d=9 mmX,d=9 mmR,d=12 mmX,d=12 mm

R

X

(ii)

1.5 1.55 1.6 1.65 1.7 1.75 1.8 1.85 1.9 1.95 2

x 109

-30

-20

-10

0

10

20

30

40

50

Frequency (Hz)

Inpu

t Im

peda

nce

(ohm

)

R,w =5 mmX,w =5 mmR,w =7 mmX,w =7 mmR,w =9 mmX,w =9 mmR,w =12 mmX,w =12 mm

R

X

(i)

0.8 0.85 0.9 0.95 1 1.05 1.1 1.15 1.2 1.25

x 1010

-30

-20

-10

0

10

20

30

40

50

Frequency (Hz)

Inpu

t Im

peda

nce

(ohm

)

R,w=5 mmX,w=5 mmR,w=7 mmX,w=7 mmR,w=9 mmX,w=9 mmR,w=12 mmX,w=12 mm

R

X

(ii)

(a) (b)

Figure 4 Variation of input impedance with frequency for different notch (a) depth and (b) width for a given notch width (5 mm): (i) Lower resonance,(ii) Upper resonance. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 50, No. 7, July 2008 1781

L2 � L1 � 2�L

C2 �C1�C

2C1 � �C

It may be noted that the two resonant circuits are coupled throughmutual inductance (Lm) and mutual capacitance (Cm) expressed as

Lm �Cp

2�L1 � L2� � �Cp2�L1 � L2�

2 � 4Cp2�1 � CP

2�L1L2

2�1 � Cp2�

(8)

Cm � ��C1 � C2� � ��C1 � C2�

2 � 4C1C2�1 � Cp�2�

2(9)

where Cp �1

�Q1Q2

and Q1 and Q2 are quality factors of the two

resonant circuits.Now the equivalent circuit of the H-shaped patch can be given

as shown in Figure 3. The input impedance of the H-shapedmicrostrip patch can be calculated from Figure 3 as

ZT � ZH �ZPZm

ZP � Zm(10)

where

Zm �1

2� j�Lm �1

j�Cm�

2.1. Calculation of Various Antenna ParametersThe reflection coefficient of the patch can be calculated as

� �Z0 � ZT

Z0 � ZT(11)

where Z0 � characteristic impedance of the coaxial feed (50 )VSWR and return loss can be calculated as

VSWR �1 � ���1 � ��� (12)

1.5 1.6 1.7 1.8 1.9 2

x 109

-50

-45

-40

-35

-30

-25

-20

-15

-10

-5

0

Frequency (Hz)

Retu

rn lo

ss(d

B)

d=5 mmd=7 mmd=9 mmd=12 mm

(i)

0.8 0.85 0.9 0.95 1 1.05 1.1 1.15 1.2 1.25

x 1010

-50

-45

-40

-35

-30

-25

-20

-15

-10

-5

0

Frequency (Hz)

Retu

rn lo

ss(d

B)

d=5 mmd=7 mmd=9 mmd=12mm

(ii)

1.5 1.6 1.7 1.8 1.9 2

x 109

-45

-40

-35

-30

-25

-20

-15

-10

-5

0

Frequency (Hz)

Retu

rn lo

ss (d

B)

w=5 mmw=7 mmw=9 mmw=12 mm

(i)

0.8 0.85 0.9 0.95 1 1.05 1.1 1.15 1.2 1.25

x 1010

-35

-30

-25

-20

-15

-10

-5

0

Frequency (Hz)

Ret

urn

loss

(dB

)

w=5 mmw=7 mmw=9 mmw=12 mm

(ii)

(a) (b)

Figure 5 Variation of return loss with frequency for different notch (a) depth and (b) width for a given notch width (5 mm): (i) Lower resonance, (ii) Upperresonance. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com]

1782 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 50, No. 7, July 2008 DOI 10.1002/mop

Return loss � � 20 log��� (13)

3. DESIGN SPECIFICATIONS

Design specifications of the H-shaped patch is given asDielectric constant of the substrate material (�r) 2.5

Thickness of the substrate (h) 1.59 mmWidth of the patch (W) 3.30 cmLength of the patch (L) 5.45 cmDesign frequency (f) 1.952 GHz

4. DISCUSSION OF RESULTS

Variation of input impedance with frequency for H-shaped patch isshown in Figure 4(a) for the different notch depth (d) and for agiven notch width (w � 5 mm). It is observed that the antennashows dualband and separation between two band is maximum forlowest value of notch depth (d � 5 mm). The separation betweentwo resonances decreases with increasing notch depth. It may benoted that the first resonance occurs at lower frequency whichdecreases minutely with increasing notch depth (d) while upperresonance occurs on the higher frequency which shows consider-able shift in frequency with increasing notch depth (d), for a givennotch width (w � 5 mm). Ratio of upper and lower resonancefrequency is variable between 4.63 and 6.16. It is further observedthat the real part of input impedance is around 50 while it ishaving lower value at higher resonance and it decreases withdecreasing notch depth. This indicates that there is a good match-ing at lower resonance frequency where as matching decreases athigher resonance frequency with decreasing notch depth for givennotch width (w � 5 mm). Similar behavior is also observed fromFigure 4(b) in which impedance is plotted as a function of notchwidth (w) for a given notch depth (d � 5 mm). Behavior of thepatch is further corroborated from the return loss data shown inFigures 5(a) and 5(b), respectively.

Variation of real and imaginary parts of input impedance isshown in Figure 6 along with results reported by Palanisamy andGarg [9] and Gao et Al. [14] for lower resonance only. It isobserved that the results of proposed method are in close agree-ment with the experimental results by Palanisamy and Garg andthat by Gao et al. using FDTD method.

Variation of lower resonance frequency as a function of centralstrip width is shown in Figure 7(a) along with reported results [14].

It is found that the resonance frequency increases almost linearlywith central strip width and the result of proposed method isalmost in good agreement with that of Gao et al. [14] using FDTDmethod.

1.24 1.25 1.26 1.27 1.28 1.29 1.3

x 109

-40

-20

0

20

40

60

80

100

Frequency (Hz)

Inpu

t Im

peda

nce

(ohm

)

R, proposedX, proposedR, Experimental [9]X, Experimental [9]R, FDTD [14]X, FDTD [14]

R

X

Figure 6 Variation of input impedance with frequency (lower resonance,d � 9.15 mm, w � 26 mm). [Color figure can be viewed in the online issue,which is available at www.interscience.wiley.com]

0 5 10 15 20 25 30 35

1

1.5

2

2.5

3

Central strip width (mm)

Res

onan

ce fr

eque

ncy

(GHz

)

proposedFDTD [14]

5 10 15 20 25 30

1.4

1.6

1.8

2

2.2

2.4

2.6

2.8

3

w (mm)

Reso

nanc

e fre

quen

cy (G

Hz)

proposedFDTD [14]

0 5 10 15 20 25

1

1.5

2

2.5

3

Side strip length (mm)

Reso

nanc

e fre

quen

cy (G

Hz)

praposedFDTD [14]

(a)

(b)

(c)

Figure 7 Variation of lower resonance frequency with (a) centre stripwidth, (b) centre strip length (w), (c) side strip length (w1). [Color figurecan be viewed in the online issue, which is available at www.interscience.wiley.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 50, No. 7, July 2008 1783

Variation of lower resonance frequency with center strip lengthis shown in Figure 7(b). It is found that resonance frequencydecreases with increasing width of notch (i.e. center strip length).Results of proposed method are similar to that of FDTD method.Little variation in the two results is due to approximation involvedin the cavity theory that is used for equivalent circuit model.

The variation of lower resonance frequency as a function ofside strip length is shown in Figure 7(c) along with the reportedresults. It is observed that the resonance frequency decreases withside strip length and the results from proposed method are in goodagreement with the FDTD method.

It may be mentioned that all the data obtained from proposedmethod are almost in good agreement with reported results. Thisvalidates the veracity of proposed method of analysis.

5. CONCLUSION

It may therefore be inferred from this analysis that H-shaped patchexhibits dual resonance behavior and the resonance frequency arevery sensitive to the dimensions of the notch. Such antenna can besuccessfully used at two different frequencies as per requirement.

REFERENCES

1. K.L. Wong and K.P. Yang, Compact dual frequency microstrip an-tenna with a pair of bend slot, Electron Lett 34 (1988), 225–226.

2. H. Iwasaki, A circularly polarized small size microstrip antenna witha cross slot, IEEE Trans Antenna Propagat 44 (1996), 1399–1401.

3. S. Maci and G.B. Gentillin, Dual frequency patch antenna, IEEE APMag 39 (1996), 13–17.

4. K.L. Wong and W.S. Chen, Compact microstrip antenna with dualfrequency operation, Electron Lett 33 (1977), 646–647.

5. C.L. Tang, H.T. Chen, and K.N. Wang, Small circular microstripantenna with dual frequency operation, Electron Lett 33 (1997), 1112–1113.

6. S.C. Pan and K.L. Wong, Design a dual frequency microstrip antennausing a shorting pin loading, IEEE AP-5, Symp Dig (1998), 312–315.

7. K.P. Yang and K.L. Wong, Inclined slot coupled compact dual fre-quency microstrip antenna with cross slot, Electron Lett 34 (1998),321–322.

8. W.S. Chen, Single feed dual frequency rectangular microstrip antennawith square slot, Electron Lett 34 (1998), 231–232.

9. V. Palanisamy and R. Garg, Rectangular ring and H-shaped microstripantennas—Alternative to rectangular patch antennas, Electron Lett 21(1985), 874–876.

10. I.J. Bahal and P. Bhartia, Microstrip antennas, Artech House, Massa-chuaetts, 1980.

11. X.X. Zhang and F. Yang, Study of slit cut microstrip antenna and itsapplication, Microwave Opt Technol Lett 18 (1998), 297–300.

12. I.J. Bahal, Lumped elements for RF and microwave circuits, ArtechHouse, Boston, 2003.

13. V.K. Pandey and B.R. Vishwakarma, Theoretical analysis of lineararray antenna of stacked patches Indian J Radio Space Phys 34 (2005),125–127.

14. S.C. Gao, et al. Analysis of an H-shaped patch antenna by using theFDTD method, Prog Electromagn Res PIER 34 (2001), 165–187.

© 2008 Wiley Periodicals, Inc.

WIDEBAND CIRCULARLY POLARIZEDMICROSTRIP ANTENNA ARRAY USINGA NEW SINGLE FEED NETWORK

Nasimuddin,1 Zhi Ning Chen,1 and K. P. Esselle2

1 Institute for Infocomm Research, 20 Science Park Road, Singapore;Corresponding author: nasimuddin@i2r.a-star.edu.sg2 ICS Division, Macquarie University, Sydney, Australia

Received 5 November 2007

ABSTRACT: A new single feed system is presented for broadband andhigh gain circularly polarized stacked microstrip (CPSM) array applica-tions. It consists of a combination of a microstrip line and a via. Thegap between elements and location of the feed are optimized to achievea good circular polarization (CP) performance with high gain. A 2 � 2sequentially rotated array was fabricated and tested. The measured 3dB axial-ratio (AR) bandwidth is 27% (3.5–4.6 GHz) and at boresightgain is 9.0 dBic over the 3 dB AR bandwidth. The measured 10 dB re-turn loss impedance bandwidth is around 51% (3.28–5.54 GHz). Theproposed new feeding network system is very useful to design the highgain and wideband CP array in a space-limited environment. © 2008Wiley Periodicals, Inc. Microwave Opt Technol Lett 50: 1784–1789,2008; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.23481

Key words: stacked microstrip antennas; circular polarization; axial-ratio; wideband; single feed; array

1. INTRODUCTION

The research on achieving high scale integration of array an-tennas for microwave subsystem in applications such as wire-less communications and radar is challenging when these sys-tems operate over an increased frequency band. In particular, aconsiderable difficulty is faced when an array of CP with broadfrequency band has to be achieved. This challenge is due todifficulty of obtaining wideband and well balanced feedingnetwork, which could meet the constraint of space-limitedenvironment packed antenna array. Circularly polarized micros-trip (CPM) antennas are particularly useful in communicationsystems where the orientation of antennas is variable or isunknown. CP is usually achieved by combining two orthogonallinearly polarized waves radiating in phase quadrature. Thereare currently two techniques commonly used in CPM antennas.In the single feed technique, asymmetry is introduced into thegeometry of the microstrip radiator so that, when the antennaexcited at a carefully selected feed point, the antenna radiatestwo degenerated orthogonal modes with a 90° phase difference.In the dual feed technique, two separate and spatially orthog-onal feeds are excited with a relative phase shift of 90°. Thedual feed approach requires the use of a 90° hybrid or powersplitter with unequal lengths of transmission line to provide thenecessary phase shift. This method generally gives a larger ARbandwidth if both the microstrip radiator and feeding networkare broadband. This technique, however, suffers from poorpolarization purity due to the cross-polarized components gen-erated by the asymmetrical feed structure. Furthermore, dual-feed technique results in a complicated antenna structure andhigher cost, in particular when stacked patches are used forachieving broadband CP operation. In addition they requirelarger feed structures, which leave less space for other circuitcomponents in the feed layer. On the other hand, single-feedCPSM antennas can be arrayed and easily fed like any linearlypolarized patch antenna. Single feed leads to a reduction incomplexity, weight, and RF loss of the feed when these ele-

1784 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 50, No. 7, July 2008 DOI 10.1002/mop