An Optical ASK and FSK Phase Diversity Transmission System

by H. van den Boom and W. van Etten, W.H.C. de Krom, P. van Bennekom, F. Huijskens, L. Niessen, F. de Leijer.

EUT Report 92-E-268 ISBN 90-6144-268-0

Qr' or a .,

Eindhoven University of Technology Research Reports

Eindhoven University of Technology

ISSN 0167-9708

Faculty of Electrical Engineering

Eindhoven, The Netherlands

Coden: TEUEDE

AN OPTICAL ASK AND FSK PHASE DIVERSITY

TRANSMISSION SYSTEM

by

H. van den Boom and W. van Etten, W.H.C. de Krom,

P. van Bennekom, F. Huijskens, L. Niessen, F. de Leijer.

EUT Report 92-E-268

ISBN 90-6144-268-0

Eindhoven

December 1992

CIP-GEGEVENS KONINKLIJKE BIBLIOTHEEK, DEN HAAG

Optical

An optical ASK and FSK phase diversity transmission system I by H. van den Boom ... let al.]. - Eindhoven: Eindhoven University of Technology, Faculty of Electrical Engineering. - Fig., tab. - (EUT report, lSSN 0167-9708 ; 92-E-268) Met lit. opg. ISBN 90-6144-268-0 NUGI 832 Trefw.: optische datatransmissie-systemen.

ABSTRACT

This report describes the results of the contribution of the Telecommunications Division

of the Faculty of Electrical Engineering, Eindhoven University of Technology to the lOP

Electro Optics, cluster I project: • Phase Diversity System'.

Specifications of subsystems have been derived, together with tolerances and consequen­

ces of these tolerances for the final system performance. For the optical network of the

phase diversity receiver a manufacturing set-up for {3x3} fused biconical taper fiber

couplers has been developed. In order to characterize planar optical networks, a set-up

has been constructed to measure the phase relations at 1523 nm. The optical frequency of

the local oscillator laser has to be locked on to the frequency of the received optical

signal. A detailed description of this locking circuit is given. A complete optical {3x3}

phase diversity transmission system has been developed that can be used as a testbed for

subsystems. The sensitivity of the receiver at a BER of 10-' is -47.2 dBm, which is 4.2

dB better than the value of the specificatons.

Boom, H. van den and W. van Etten, W.H.C. de Krom, P. van Bennekom,

F. Huijskens, L. Niessen, F. de Leijer.

AN OPTICAL ASK AND FSK PHASE DIVERSITY TRANSMISSION SYSTEM.

Faculty of Electrical Engineering, Eindhoven University of Technology,

The Netherlands, 1992. EUT Report 92-E-268.

Address of the authors:

Telecommunications Division

Faculty of Electrical Engineering

Eindhoven University of Technology

P.O. Box 513, 5600 MB Eindhoven, The Netherlands

- IV -

CONTENTS

I INTRODUCTION I

2 ANAL YSIS OF THE PHASE DIVERSITY SYSTEM 3

2.1 Coherent system design 3

2.2 Phase diversity reception 4

2.3 Description of the transmission system 6

2.4 Sensitivity penalty due to non-ideal components 8

2.4.1 Impact of the laser phase noise and IF/LP bandwidth 8

2.4.2 Impact of the laser intensity noise 1l

2.4.3 Impact of phase mismatch and gain imbalance 13

2.4.4 Impact of the extinction ratio of the external modulator 15

3 THE TRANSMITTER AND LOCAL OSCILLATOR LASERS 20

3.1 The DFB lasers 20

3.2 Laser fiber coupling 22

3.3 Optical isolation 24

3.4 The temperature controllers 26

3.5 The current control circuit 28

4 THE TRANSMITTER 31

4.1 The optical amplitude modulator 31

4.2 The FSK modulator 33

5 THE RECEIVER 35

5.1 The optical network 35

5.1.1 The 3x3 fiber coupler 35

5.1.2 Phase diversity network measurement set-up 38

5.2 The PINFET OlE converter module 44

5.3 The IF section 46

5.4 The frequency controller 52

5.4.1 Principles of the frequency locked loop 53

5.4.2 The realized frequency locked loop 54

5.4.3 Conclusions 55

5.5 The clock and data recovery circuit 55

6 EXPERIMENTAL RESULTS

6.1 IF spectra

6.2 BER measurements

6.3 The video codec

REFERENCES

List of abbreviations

APPENDIX I Specifications

- v -

APPENDIX 2 Data sheets of the Philips DFB lasers

APPENDIX 3 Data of the GEC optical amplitude modulator

APPENDIX 4 Circuit diagrams

APPENDIX 5 Source code listings of the frequency locked loop

59

59

63

64

67

71

72

73

79

80

92

1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1

I

1

Chapter 1

Introduction

Optical coherent detection offers the benefits of a larger receiver sensitivity (10 a 20 dB)

and of a more dense multiplexing of channels in the frequency domain. Those advantages

are to be exploited for subscriber networks in the Integrated Broadband Communication

Network (lBCN).

A phase diversity receiver is extremely tolerant for laser linewidths compared with an

optical homodyne receiver and requires a smaller bandwidth of the IF electronics

compared with a heterodyne receiver.

The history of the project is as follows.

In September 1987 the steering committee SGOG (Stuurgroep Gemeenschappelijk Onder­

zoek Glasveze1communicatie in Nederland) initiated a research project entitled "Integrated

Optics voor het IBCN" and a proposal was submitted to the Ministry of Economic

Affairs, asking for financial support. This ministry, however, placed the proposal in the

broader research field Electro-optics and asked the IOP/IC-technology committee for

advice. This committee in its turn set up a sub-committee lOP Electro-Optics advisory

board. Based on letters of intend of the SGOG partners, the board issued a workplan on

electro-optics and invited, among others, the partners to submit project proposals in

accordance with the subjects defined in the workplan.

A combination of research groups from PTT Research Neher Laboratory, Eindhoven

University of Technology, Delft University of Technology, University of Twente and

Physics and Electronics Laboratory TNO submitted a proposal which was deduced from

the former SGOG proposal and which was entitled "Opto-electronic IC's for the Integrat­

ed Broadband Communication Network (lBCN)". This proposal, although slightly

adapted, has been approved by the committee and was named "Phase Diversity System" .

2

The contracts have been undersigned in October 1989, which meant the starting point of

the execution of the project. Both written and oral progress reports have been delivered

on February 15 1990, September 6 1990, February 14 and September 5 1991.

3

Chapter 2

Analysis of the phase diversity system

In this chapter the realized ASK (Amplitude-shift Keying) and FSK (Frequency-shift

Keying) phase diversity transmission system is described and the performance conse­

quences of non-ideal components are discussed. Specifications of subsystems and system

components are given. First the design of the system is described.

2.1 Coherent system design

To gain full advantage of the huge transmission capacity of optical fibers the use of

coherent optical reception is the best solution. Optical coherent detection offers two

important advantages with respect to conventional intensity modulation/direct detection:

1) improved receiver sensitivity;

2) greatly enhanced frequency selectivity.

In optical coherent detection the received signal is mixed with the signal of a local

oscillator. In this wayan IF signal (Intermediate Frequency) is obtained. The spectrum of

the IF signal is equal to the convolution of the spectrum of the received and local

oscillator signal. Just as in radio communication reception, if the local oscillator is phase

locked with the received optical signal this is called homodyne detection, if not this is

called heterodyne detection. With the present day trend towards higher bit rates, a

conventional heterodyne receiver is not practical because of the large bandwidth required

for the photodiodes, preamplifiers, filters and demodulators of the receiver (3-5 times the

bit rate) [1]. For homodyne receivers the baseband bandwidth is sufficient, and it has the

additional advantages of lower noise and a smaller frequency band in optical coherent

multicarrier systems. However a homodyne receiver requires phase-locking between the

transmitter and local oscillator waves, which is hard to realize. With a heterodyne phase

diversity receiver it is possible to choose a lower IF compared with a conventional

4

heterodyne receiver so the bandwidth required of the IF electronics is lower. It is even

possible to choose an IF of zero, which is often called a pseudo-homodyne receiver. This

receiver does not use phase-locking and the laser linewidth requirements are relaxed as

compared to those of a synchronous receiver [2] [3].

The developed transmission system uses a heterodyne phase diversity detection scheme.

The modulation scheme is either 140 Mb/s ASK or 140 Mb/s FSK. At the receiving end,

for FSK the single filter demodulation method is used, so the same receiver can be used

for the ASK as well as for the FSK modulation scheme. For ASK modulation an

expensive external optical amplitude modulator has to be used. FSK modulation can be

realized by direct modulation of the transmitter laser bias current which is easier,

however the transmission bandwidth for FSK modulation is much greater than for ASK

modulation.

2.2 Phase diversity reception

With direct detection systems the received optical signal is directly applied to a

photodiode which generates a current proportional to the optical power received. The

coherent receiver adds a locally generated optical wave to the optical received wave. This

combined signal is applied to the photodiode. Let us suppose, that both the signal wave

and the local oscillator wave are polarized linearly in the same direction and let us denote

these two waves by respectively,

(2.1)

(2.2)

The photodiode current is proportional to the power of the sum of the waves which is

(2.3)

5

In this expression the third term is the most interesting one. This term contains all the

information of the received signal, namely the amplitude Po, the frequency Wo and the

phase ,I>o(r). The magnitude of this third term can be increased by boosting the local

oscillator power PI' The first term Po is the power of the signal received which provides

the photocurrent for direct detection and is weak compared to the third term. The second

term PI from the power of the local oscillator is a DC term.

In case of ASK modulation the received optical power Po is modulated and in case of

FSK modulation the frequency of the received optical signal Wo is modulated. If for

instance the optical frequencies of both lasers are the same so WO-WI =0, the third term

will become zero if cPo(t)-,p'(r) = 7r12, rendering the system useless. This problem can be

solved by using a phase diversity detection scheme. In Figure 2.1 a general diagram of an

optical coherent phase diversity receiver with low pass filters (LP) and square law or

envelope detectors (D) is shown. In this diagram a {3x3} optical network, for instance a

fused fiber coupler, is used. Alternatively, a planer optical network can be used, and then

a {4x4} optical network seems to be more suitable.

~

P, Optical

P, hybrid

"

Fig. 2.1: General diagram of an optical coherent phase diversity receiver with low pass filters

(LP) and square law or envelope detectors (D).

When each output signal is detected by a photodiode, de signal components of the N

currents (N = 3,4) have 27rIN phase difference between them, provided that each output

has the same output amplitude. When the currents are squared, a baseband term arises, as

well as a double frequency term. Adding the squared current terms cancels out the double

6

frequency terms, whereas the baseband terms increase the signal in a single branch N ti­

mes. In this way fading due to the phase noise of the lasers is avoided.

2.3 Description of the transmission system

In Pig. 2.2. a block diagram of the system is shown. The upper part of the figure is the

transmitting-end and the lower part is the receiving-end of the system. The transmitter

and the receiver are connected by a standard single mode fiber (SMP).

Because the optical frequency of the laser is very susceptible to variations in temperature

a temperature controller is used for stabilization. The laser diode is mounted on a peltier

element and the temperature is detected by an NTC-resistor. Because the laser diodes are

very sensitive to optical reflections a 90 dB optical isolator is inserted right behind the

taper. In case of FSK modulation the output of the isolator is directly connected to the

transmission fiber (SMF). Por ASK modulation the isolator is connected to the transmis­

sion fiber via the optical modulator.

For FSK modulation the digital input signal is connected to the current control circuit

which modulates the current of the transmitter laser diode. This current control circuit

also provides a stable DC current for the laser diode. By modulating the current, the

output power of the laser is modulated as well, but this amplitude modulation is negligi­

ble. For ASK modulation an external optical amplitude modulator is used. In this case it

is not possible to modulate the current, because by modulating the current the optical

frequency of the laser is modulated as well.

At the receiving-end the optical signal received, is mixed with the optical output signal of

the local oscillator laser diode by the {3x3} optical network. Delft University of Technol­

ogy, Department of Electrical Engineering, Laboratory of Telecommunication and

Remote Sensing Technology (TUD/TTT) has developed a planar {4x4} optical network.

A phase measurement set-up for these planar networks has been developed. The local

oscillator laser diode is optically isolated as well and both the temperature and the current

are stabilized in the same way as the transmitter laser diode. The polarization of the

output of the local oscillator laser is controlled by a hand polarization controller. The

optical signals are detected by three PIN-PET modules. These PIN-FET modules consist

7

of a photodiode and a front-end amplifier. In order to obtain a stable difference frequency

or intermediate frequency (IF), the optical frequency of the local oscillator laser is

controlled by the frequency controller, which controls the laser current. The output

signals of the PIN-FET modules are filtered by the IF lowpass filters (LPF) and demodu­

lated by square law detectors. The signals of the three branches are added and filtered by

a post detection filter (LPF). The data clock and data are recovered by a phase-locked

loop and the sampler, respectively.

current control circuit

FSK-IN j

temperature controller

transmitter laser d·ode I

.~ peltier element

isolator optical ::;;;. modulator

ASK-IN J TRANSMITTER

PHASE DIVERSITY RECEIVER

Fig. 2.2: Block diagram of the phase diversity system.

8

2.4 Sensitivity penalty due to non· ideal components

2.4.1 bnpact of laser phase nolore and the IF/LP bandwidth

The spontaneous emission causes phase fluctuations in the laser output. This phase noise

decreases the coherence time of the laser light. Limited coherence time manifests itself as

a broadening of the laser spectrum. The laser spectrum, which under ideal conditions

consists of one sharp line, can be shown to have a Lorentzian line shape due to the

presence of phase noise.

LASER SPECfRUM

-5

~

~ -10 '0

.5 ~

M

~ -15 a

"-

-20

-25_5':---_':-4 --_73 ---"::"2---":-1--~0- 1 2 3 4 5

Frequency (in Hz) X108

Fig. 2.3: Example of an equivalent baseband laser spectrum due to laser phase noise.

The laser spectrum is given by the following equation [4-5]

21r[(.1v/2)' + (f-vJ'] (2.4)

where Vo is the resonant frequency of the laser cavity and .1v is the laser linewidth, which

9

is defined as the 3.0 dB linewidth, also called the Full-Width-Half-Maximum (FWHM)

laser linewidth. It is obvious that laser phase noise is a nonnegligible impairment in phase

modulated communication systems. However, even coherent systems employing

amplitude modulation suffer from sensitivity degradation due to the laser phase noise.

This has to do with the unavoidable filtering of the IF signal received. Filtering of phase

noise can lead to a conversion of laser phase noise into amplitude noise, which for

amplitude modulation is an important impairment [6].

Laser phase noise also causes a broadening of the IF spectrum, and therefore, the

bandwidth of the IF filter should be increased in order to accommodate the necessary

signal power. The latter implies that the noise bandwidth is extended and the impact of

the shot and thermal noise is increased at the expense of receiver sensitivity [7].

If the intersymbol interference may be neglected, the power spectrum of the signal at the

input of the IF filter is given by the following convolution

(2.5)

where SM(f) is the modulation spectrum. In case of ASK modulation SM(f) is given by [8]

sin2( 7rTof/-/'» TTl/-/)'

+ sin2( 7rTl/+/'» 1 TToif+/'Y J'

(2.6)

where A is a constant, and!, represents the carrier frequency. Equation (2.6) implies that

when Av increases, the power spectrum 0(1) of the signal at the input of the IF filter

broadens. Consequently the bandwidth of the IF filter should be increased in order to

accommodate the same amount of signal power as in case without laser phase noise.

However, this also increases the (shot) noise bandwidth, and therefore, results in a

sensitivity penalty. The loss in receiver sensitivity (SNR) due to the increase of the noise

bandwidth, is depicted in Fig. 2.4 versus the normalized laser linewidth AvTo. This for a

in the frequency domain rectangular IF filter, which passes 95 % of the incoming signal

power.

10

7 ---_ ...... / . ./--

6

5 ill , , TI ./

/ c 4 /

co /

>. / ~

Iii 3 c / IlJ

CL

2

OL-~~--~~--~~--~~--~~~~~~--~~

0.00 0.30 0.60 0.90 1.20 1.50

Normalized laser linewidth

Fig. 2.4: Loss of SNR versus the normalized laser line width AvTo.

When large IF filter bandwidths are used in ASK receivers to overcome the spectrum

broadening due to laser phase noise, a post-detection filter, with a bandwidth much

smaller than the IF filter bandwidth, is usually applied to improve the receiver sensitivity

[2]. By matching of the IF filter and the post -detection filter, an optimal IF filter

bandwidth can be found for which the sensitivity degradation is minimal. Further

increasing of the bandwidth results in an excess sensitivity degradation, since too much

(shot) noise is passed. On the other hand, if the IF filter bandwidth is too small, a BER

floor seriously degrades the receiver performance, due to the conversion of laser phase

noise into intensity noise and vice versa [9]. For ASK systems the BER floor depends on

the relation between the normalized IF filter bandwidth (BifTo) and the normalized laser

linewidth (Ll.vTo). The larger BifTo, the lower the BER floor.

For optimized IF and post-detection filtering, phase diversity ASK receivers have the

same phase noise performance as heterodyne ASK receivers, and they can tolerate the

same amount of laser phase noise [4,10,11]. For ASK receivers, the sensitivity penalty,

due to laser phase noise versus the normalized laser linewidth, is given in Fig. 2.5 for a

11

BER=IO-9, and optimized IF and post-detection filtering [10,11].

3.00 ~--- ------1

2.50

~ 2.00

g >, 1.50

== III C OJ

Q 1.00

0_50

I , 0.00 ~ __ L-~ __ ~ __ ~ __ ~ ___ -L __ ~ __ ~ __ ~' ~

0 0.5 1 1-5 2 25 Normalized laser' linewidth

Fig. 2.5: Sensitivity penalty versus tJ.VTo for ASK receivers (SER = 10-9).

2.4.2 Impact of the laser intensity noise

Beside coherent light due to the stimulated recombination of Electron Hole Pairs (EHP's)

in the laser cavity, a semiconductor laser also emits socalled incoherent light generated by

the spontaneous recombination of EHP's in the laser cavity. For semiconductor lasers, the

spontaneous emission of photons in the laser cavity gives onset to noise problems. This

spontaneous emission is the main cause for intensity noise. Due to the small device size,

the optical energy storage in the cavity is small, and therefore, the sensitivity to the

perturbations of spontaneous emission in semiconductor lasers is usually significant. The

influence of the spontaneous emission could be reduced by increasing the size of the laser

cavity. Intensity noise manifests itself as a random fluctuation of the optical power, and is

especially pronounced near the threshold and diminishes as the laser bias current is

increased. Statistically, intensity noise behaves like shot noise, after detection by a

photodiode which implies that it is also Gaussian distributed and has a flat power

spectrum (white noise) over a wide frequency range. Intensity noise is usually character-

12

ized by the so-called Relative Intensity Noise (RIN). The RIN spectral density is defined

as the variance of the optical power fluctuations, divided by the mean optical

power squared in 1Hz bandwidth (dBlHz).

The sensitivity degradation for the {2x2} and {3x3} phase diversity ASK receiver has

been analyzed in [9]. The sensitivity penalty has been computed with respect to the

performance of an ideal heterodyne ASK receiver. The bit rate is taken to be ISO Mbitls

and the influence of the phase noise has been neglected. The results are depicted in

Fig. 2.6 for three practical RIN levels -ISO, -135, and -125 dB/Hz, respectively.

30 -------_. __ ._---------

ill 25 -125

D

g 20

>. " ~

" <Il C Qi 15 0.

-- U5

>. ~

;; 10 .,

iii c Q!

(j) -150 ...... ----

5 -----I

~.-~--=---:::.- - --I 0 "---__ '-_---' __ ---' __ ----.l. ___ --"-___ J -30 -25 -20 -15 -10 -5 o

Local oscillator power (in dBm)

Fig. 2.6: Sensitivity penalty versus P, for several RIN values for a {2x2} ASK receiver

{ __ I and a {3x3} ASK receiver '- __ I with both an optimal threshold level.

For both the {2x2} and the {3x3} phase diversity ASK receiver an optimum value for the

local oscillator power (PJ is found, for which the sensitivity penalty with respect to the

ideal heterodyne ASK receiver is minimal. This value depends on the position of the

threshold level and the local oscillator laser RIN. For each combination of PL and a RIN

level, the threshold level ought to be optimized in order to minimize the BER, implying a

minimum sensitivity penalty. For an optimum threshold value and a BER= 10-9, the {3x3}

13

receiver outperforms the {2x2} receiver for a local oscillator power larger than the

optimum value of P" The reverse is true for values of PL smaller than this optimum

value.

2.4.3 Impact of phase mismatch and gain imbalance

The key component of all phase diversity receivers is an optical hybrid that provides the

means of recovering the modulated optical signal. All phase diversity receivers are

sensitive to gain imbalance and an aberration in the phase relations between the different

receiver branches. The aberration in the phase relations can originate from the optical

hybrid, and the gain imbalance in the receiver branches can be caused, firstly, by an

unequal amplification at the different IF stages, or secondly, by an unequal power

distribution over the outputs of the optical hybrid. In [9] the BER's have been calculated

for a {2x2} and a {3x3} phase diversity ASK receiver, using non-Gaussian Probability

Density Functions (PDF's) for the signals at the threshold comparator, and various values

for the percentage of gain imbalance and the phase mismatch.

The gain imbalance has been modeled by placing an additional amplifier in each receiver

branch. One of the receiver branches is taken as a reference with an amplification factor

equal to one. The amplification factor of the other branch in the {2x2} phase diversity

receiver is given by (1 +0), introducing a gain imbalance of 100*0% (-1 <0< 1). For the

{3x3} phase diversity ASK receiver, the gain imbalance is introduced in the same way

except for one difference. The percentage of gain imbalance is assumed to be symmetri­

cal.

The phase mismatch for the {2x2} receiver is modeled by introducing an extra phase shift

term in one of the two relations for the photocurrents at the IF stage, taking the phase of

one receiver branch as a reference. For the {3x3} receiver, the phase mismatch is

modeled by introducing an extra phase shift term, in two out of the three relations for the

photocurrents at the IF stage, taking one receiver branch as a reference.

In [9] upper-bounds for the gain imbalance and the phase mismatch in both receiver

structures have been derived. The criteria was given a certain percentage of gain

imbalance the maximum allowable phase mismatch, which increases the signal power P

required to achieve a BER = 10·' by maximal 0.5 dB. In the Tables 2.1 and 2.2, maxi­

mum values for the phase mismatch are presented, for a gain imbalance of +/-5% and

14

+1-10%, respectively.

Table 2.1

Maximum allowable phase mismatch for a {2x2} and a {3x3} phase diversity ASK receiver as a function of the gain imbalance.

BER = 10-9, non-optimal threshold, penalty margin = 0.5 dB

Gain imbalance 0% 5% 10% -5% -10%

{2x2} ASK receiver ±6.0° ±7.0° ±S,SO ±4.0° -----

{3x3} ASK receiver non-symmetrical ±9.0° ± 11.8° ---- ±4,SO -----configuration

{3x3} ASK receiver 5's" 6.2° ---- 4.0° -----

symmetrical -5.0° _7.0° _1.5° configuration ---- -----

---- = not possible within a penalty margin of 0.5dB

± = for positive and negative values of the phase mismatch

Tabel2.2

Maximum allowable phase mismatch for a {2x2} and a {3x3} phase diversity ASK receiver as a function of the gain imbalance. BER = 10-9

, optimal threshold, penalty margin = 0.5 dB

Gain imbalance 0% 5% 10% -5% -10%

{2x2} ASK receiver ±8.5° ±8.0° ±6's" ±8.0° ±6.2°

{3x3} ASK receiver non-symmetrical ±13.0° ±13.0° ± 12.0° ±12.0° ±10.0° configuration

{3x3} ASK receiver 8.0° 7.0° 5's" 9.1 ° 9.0° symmetrical

-7.8° _7.8° -7.8° -5.8° _4.0° configuration

± = for positive and negative values of the phase mismatch

15

In [9] has been concluded, that in comparison with the ideal situation, without any of

these imperfections, the signal value at the input of the threshold comparator becomes a

function of the radial offset frequency Wif. This results in a time varying character of the

BER, which leads to a sensitivity degradation. The sensitivity degradation has been

calculated with respect to the ideal {2x2} or (3x3} phase diversity ASK receiver for a

BER= 10~9. Since ASK demodulation highly depends on the threshold value, the BER's

have been calculated for a threshold value optimized for the case without any imperfec­

tions, and for a threshold value optimized for a certain combination of gain imbalance and

phase mismatch. By optimizing the threshold level, the sensitivity penalty can be

significantly reduced.

2.4.4 Impact of the extinction ratio of the external modulator

The ASK modulation of the optical signal wave transmitted is usually obtained by means

of an external amplitude modulator [12]. The modulator switches the amplitude of the

optical wave transmitted between two levels, corresponding to the information signal. In

case of ideal operation, the amplitude of the wave transmitted should be switched between

zero and a certain maximum level. However, for practical available amplitude modula­

tors, the lowest level of zero (no power transmission) is difficult to obtain at high bit

rates. The ratio of the amplitude of the signal part of the IF photocurrent in case a space

is sent and in case a mark is sent is an important parameter. This ratio is defined as the

Extinction Ratio (ER) of the external amplitude modulator. In [9] the sensitivity degrada­

tion, due to the use of an amplitude modulator with nonzero ER, has been analyzed for a

{2x2} and {3x3} phase diversity ASK receiver. The sensitivity degradation has been

computed with respect to the ideal (2x2} and {3x3} phase diversity ASK receiver,

respectively, and special attention is given to the optimal threshold level in dependency of

the ER, and the impact of nonzero ER's on the shape of the IF modulation spectrum.

In Figure 2.7, the results for a {2x2} phase diverisity ASK receiver are presented for a

(non-)optimized threshold value, and I) varying from 0.0 to 0.25.

From this figure it can be concluded that the degradation due to nonzero values of the

ER, can be significantly reduced by optimization of the threshold level. For the {3x3}

phase diversity ASK receiver, the results are depicted in Figure 2.8.

m u

.~

>-~

III c QJ [L

16

5 ------------------.-.--------- --- /j non-optimal/I threshold

4

3 optimal threshold _______

2

o~~~~l-----L----L----L---~

0.0 0.05 0.1 0.15 0.2 0.25

The Extinction Ration

Fig. 2.7: Sensitivity penalty versus extinction ratio for a {2x2} phase diversity ASK receiver,

and a BER ~ 10".

5

non-optimal

threshold 4

m u

3 g >- optimal ....

threshold iii c 2 QJ

Q.

o~~~~-~-~--~--~ 0.0 0.05 0.1 0.15 0.2 0.25

The Extinction Ratio

Fig. 2.8: Sensitivity penalty versus extinction ratio for a {3x3} phase diversity ASK receiver,

and a BER ~ 1 0".

17

From this figure, it can be concluded that similar as for the {2x2} receiver, the degrada­

tion due to a nonzero ER can be reduced by optimization of the threshold level.

The IF ASK spectrum for nonzero extinction ratios.

After detection of the composite wave and appropriate IF low-pass filtering, the signal

values in both receiver structures are given by [9]

sp) = ~ RVPLPS b(t) cosrt;(t)],

where

N = 2,3

jj(t) = ... j/2 + c/>(t)

jj(t) = 2?rj/3 + c/>(t)

(the number of receiver branches),

for the {2x2} phase diversity ASK receiver,

for the {3x3} phase diversity ASK receiver,

(2.7)

and b(t) is the signaling waveform. It is a known feature of two lasers with a Lorentzian

power spectrum, that after combining and mixing, the resulting power spectrum is also

Lorentzian. It has a FWHM linewidth which equals the sum of both laser linewidths

separately [9]. This implies that for the computation of the ASK modulated laser

spectrum, it is sufficient to perform the following convolution

(2.8)

where Sk,,,,(j) is the equivalent baseband laser power spectrum, which has a Lorentzian

line shape. Moreover, since the transmitting and local oscillator laser have equal

characteristics, the FWHM bandwidth is twice the laser linewidth of a single laser. M(j)

represents the modulation spectrum of b(t). The modulation spectrum M(j) can be

calculated to be [9]

18

(2.9)

Substitution of M(f) in Equation (2.8) finally results in the (equivalent baseband) ASK

modulated power spectrum Sm(f) for an ER of a.

(2.10)

A computed baseband ASK modulation power spectrum is given in Figure 2.9.

~

CQ -0 c v ~

~ .e

0

·1

-2

-3

-4

-5

-6

-7

-8

-9

-10 -1 -0.8 -0.6 -0.4 -0.2 0 0.2

Frequency (in Hz)

0.4 0.6 0.8

1 1

xl(}'l

Fig. 2.9: The computed equivalent baseband ASK modulated power spectrum for a PRBS of

140 Mbitls, an extinction ratio of 118, and FWHM laser line widths of 30MHz each.

19

For an increasing value of the ER, the FWHM bandwidth decreases. This can easily be

explained, since the power spectrum Sm(/) is the convolution of the modulation spectrum

and the laser spectrum, the spectrum Sm(/) is broader than both power spectra separately.

However, for values of the ER approaching towards one, M (/) becomes a delta function.

The convolution of Sm(/) and M(/) then results in the un modulated laser spectrum, with a

FWHM bandwidth equal to the sum of the FWHM laser Iinewidths.

In conclusion, the sensitivity degradation for the {2x2} and {3x3} phase diversity ASK

receiver, due to the use of an external amplitude modulator with a nonzero ER, highly

depends on the position of the threshold level. By optimization of the threshold level the

sensitivity penalty can be significantly reduced. For an optimized threshold level and a

BER = 10··, the {3x3} receiver is less sensitive to a nonzero value of the ER than the

{2x2} receiver. For the ER of 118 and a BER = 10··, the sensitivity penalty for the {2x2}

and the {3x3} phase diversity ASK receiver has been calculated to be 2.25 dB, and 2.13

dB, respectively (see Figure 2.7 and 2.8).

For ASK modulation, the FWHM bandwidth of the IF power spectrum is broadened in

comparison with the FWHM bandwidth of the un modulated laser spectrum. Increasing the

value of the ER leads to a smaller modulation index and therefore to a smaller value of

the FWHM bandwidth.

20

Chapter 3

The transmitter and local oscillator lasers

In the transmitter as well as the receiver the same kind of laser, constructions for the

laser fiber coupling, optical isolation and control circuits for temperature and DC current

where applied. Therefore these subjects are described in a separate chapter.

3.1 The DFB lasers

For coherent detection single mode lasers have to be used. The same type of laser diode

is used for both the transmitter laser and the local oscillator laser at the receiving end.

The laser diodes are GaInAsP DFB (Distributed Feed Back) lasers from Philips. A part

number or other designation is unknown.

The lasers had to be provided with a temperature sensor, a thermo-electric heatpump and

a laser fiber coupling mechanism. The laser itself is approximately 0.3 mm in square and

is soldered on a small heatsink. This small heatsink is mounted on a second gold

metalized copper heatsink of about 2 x 6 mm. Since the optical frequency of the lasers is

susceptible to variations in temperature (10 - 20 GHz/K), it is stabilized by a temperature

control circuit. The temperature sensor (Siemens K19 thermistor) is mounted as close to

the laser as possible. The sensor is provided with thermo conducting paste and glued

between the heatsink, where the laser is soldered on, and the ridge of the copper heatsink.

The copper heatsink with the laser diode is mounted on a Peltier thermo-electric heat­

pump.

At 25 degrees C the wavelength of the transmitter laser is 1539.2 nm and 1538.2 nm for

the local oscillator laser. The wavelengths or optical frequencies of the lasers are matched

by varying the temperature of the lasers. In Appendix 2 the data and characteristics of the

laser diodes are given. The maximum output power of the lasers is 5 mW. The linewidths

of the lasers are measured with a self homodyning interferometer. With self homodyning

21

the optical output signal and a delayed optical output signal of a laser are combined. This

signal is detected by a photodiode. The spectrum of the photocurrent is measured by

means of a spectrum analyzer. With this method the bandwidth of the measured spectrum

is twice the bandwidth of the spectrum of the laser because the laser output signal is

combined with a delayed signal which has the same frequency and spectrum. In Fig. 3.1

the spectrum of the transmitter laser is shown at 4 m W output power. The horizontal

frequency scale must be divided by two so the 3 dB bandwidth of the spectrum is equal to

the FWHM (Full Width Half Maximum) linewidth of the laser which is approximately

20 MHz.

~ L -21. 00 dBm - -- --- --~~ - - -TTE 10 [B .00 dB/DI V - -- -~- ---

-~ ----~ -----

I~ ~

)-~ \ ~ .. ~ _____ 1 _____

-----I

f----- - i I 1"'~~\l1

l ----

,~

STRRT 0 Hz 'RB 1.00 MHz -VB 3.00 kHz

-----~ -~~~ MIC1~ ---- -----~- -----~-

-

i -----+- -~- --~- ------- ~

I

~

-I--~-- -~I---I __ J_ . ---

1

I

i STOP 50.00 MHz

ST 50.00 msec

Fig. 3.1: Spectrum of the transmitter laser at 4 mW output power.

The Relative Intensity Noise (RIN) is measured as well, using the HP 71400 Lightwave

Signal Analyzer. RIN is defined as the variance of the optical power fluctuations, divided

by the mean optical power squared. This noise of the lasers highly depends on any

reflected light returning into the laser cavity, among other things. In Fig. 3.2 the relative

22

intensity noise and the noise floor of the analyzer are shown. The optical fiber connectors

were provided with index matching liquid, to minimize the displayed intensity noise. Both

spectra are rather flat and the traces hardly differ so the RIN is smaller than -128 dBlHz.

RL -9 81 dBm RTTE 0 B 5.00 dB/D V RVG WR - .8 dEm MARK R R. . N. 1. 00 GHz -62. 1 dB 1

START 0 Hz RS 3.00 MHz

(1 ~ z )

'VB 10.0 kHz

MKR #lA~KQ 1 000 GHz

..L

-6 .61 8e (1 Hz) L GHTW VE Of TICRL

SRMF LE

~

STOP 2.900 GHz ST 290.0 msee

Fig. 3.2: Relative Intensity Noise and the noise floor of the analyzer.

3.2 Laser fiber coupling

Since the DFB lasers were not packaged and pigtailed, the coupling of the laser light into

a fiber had to be realized in our laboratory.

Because of the rather large dimensions of the optical isolators and to avoid backreflection

into the laser cavity, the isolators could not be placed in the optical path directly.

To avoid a change in the linear state of polarization of the laser light, the .coupling

between the laser and the isolator was made with a single-mode polarization maintaining

23

fiber. The fiber had a length of approximately 30 cm.

For the laser fiber coupling, the hemispherical taper-end method was used [32]. The

fabrication process consists of the following steps :

1 - Elongation of a fiber by heating and pulling it in an electric arc discharge. The

biconical shape formed this way, has a region of minimal diameter, the waist. The

process is stopped when the desired diameter of the waist is reached.

2 - Cutting the fiber near the waist.

3 - Heating the top of the taper on which a hemispherical surface is formed by the surface

tension of the glass.

Low coupling efficiency would be obtained if one would apply above method directly to a

Panda fiber because of the presence of the SAP's (Stress Applying Parts ).

To solve this problem a Panda fiber was fused to a standard single-mode fiber, see

Fig. 3.3.

panda standa,d

1 2

L __ -L-I ==== 3 4

Fig. 3.3. Fabrication of hemispherical tip.

The waist was drawn in the standard fiber close to the fused splice at a distance of 2 to 3

cm. In the short section of the standard fiber the state of polarization of the light coupled

from the laser into the standard fiber will not change significantly. The manufacturing

was done with the setup that we normally use for making fused biconical taper couplers.

For easier handling and aligning, the uncoated part of the fused fibers was fixed in a

silica capillary, with the tapered-end leaving at one side.

The curvature of the radius of the hemisphere tips is about 20 to 30 /lm. The maximum

coupling is obtained at distances from fibertip to laser varying 20 to 40 /Lm. The coupling

24

efficiency slightly depends on the divergence of the laserlight.

We measured efficiencies of about 21 to 28 %. Improvement can be obtained by making

the radius smaller. For smaller radii of the fibertip, it should be positioned closer to the

laser to get efficient coupling. However, it will increase the risk of damaging the laser

facet during experiments.

The capillary with the protruding hemispherical fibertip is poSitioned before the DFB

laser by a precision translation stage for the z-axis and two adjustment constructions, both

for x and y direction. Each adjustment construction possesses a way to move in the x and

y direction, see Fig. 3.4. A translation in the y-axes is done by tilting the upper arm with

an adjustment screw, so that it pivots about a point (q). The arm acts as a lever, so that

about one fourth of the screw translation is effected on the taper. Tilting the lower arm

results in a x-translation caused by pivoting about point p. In fact the two movements are

not in orthogonal planes, but are displacements on the outline of two rather large circles.

The adjustment construction, closest to the laser, serves for the coarse alignment. The one

clamping the rear side of the capillary is meant for fine adjustment, see Fig. 3.4.

Likewise the block on which the laser is mounted, the construction is also made of alumi­

num to minimize the effects of thermal expansion on the coupling behaviour.

,-taper )

r-;~Hq

Fig. 3.4: x-v manipulator and Laser-fiber manipulator.

3.3 Optical isolation

If laser energy, travelling in a fiber optic system, is reflected back into the cavity of the

transmitting laser several unwanted effects can arise. For instance, amplitude instability.

To reduce and eliminate effects of optical feedback, it is necessary to optically isolate the

laser.

25

In this system it is done the following,

1 -Preventing reflection from the laser pigtail, see chapter 3.2.

2 -Preventing reflections from the downstream system by using optical isolators.

Ad .1 The end of the Panda fiber is polished to get a flat endface with an angle of 10

degrees to the normal on the optical path, see Fig. 3.5. In this way returned power from

Fresnel backreflections at the end of the pigtail will be launched in the fiber cladding and

fade out. A collimated beam is necessary to pass through the optical isolator. Therefore a

small spherical lens is positioned right behind the pigtail. The spherical lens is partly

flattened and this flattened plane is fitted at an angle of 6 degrees to avoid backreflection

into the pigtail. The lens has been provided with an antireflection V-coating for 1550 nm.

The collimated beam diameter is about 500 I'm. A theoretical reflection loss of 90 dB can

be reached in this construction, also called lensferrule. These lensferrules were fabricated

at Philips Research Laboratories in Eindhoven.

.---------~ r---------, I I I

I

I

I I _________ J L _________ ---.J

-90 dB lensferr'"ule Isolator 1 isolator 2 lensferrule

Fig. 3.5: The optical isolation.

Ad.2 The optical isolator is placed in the collimated beam behind the 90 dB lensferrule. It

consists of two cascaded Faraday rotators and polarizers in the same housing, and has a

measured isolation of 64 dB at 1550 nm. They were manufactured by OFR (USA). The

collimated beam is coupled into another lensferrule, which is connected to another short

piece of polarization maintaining fiber (Panda). This second lensferrule has no angle nor

is the lens flattened.

The measured optical spectrum of the DFB-Iaser was still unstable and so it was decided

26

to place a second isolator in series with the first. The second has an isolation of 42 dB

and has a single Faraday rotator. Much of the unwanted effects are eliminated this way.

Application of two optical isolators makes it also possible to position each one at a small

angle with respect to the optical path to avoid backreflections. The second isolator

compensates the parallel displacement of the collimated beam caused by the first one.

The polarizer of the first isolator is placed parallel to the plane of polarization of the

beam to maximize the transmission. As the double Faraday rotator in this isolator rotates

the polarization 90 degrees, the second isolator has to be adjusted to the new orientation.

The second isolator changes the plane of polarization 45 degrees and the orientation of the

receiving lensferrule is fitted to launch the polarized beam in the Panda fiber parallel to

the SAP's.

The lensferrules are clamped in V -grooves of a massive block and cannot be tilted. The

transmission loss caused by the lensferrules is 1.4 dB. The loss of the lensferrules and

both isolators is 2.6 dB, transmission 55%.

The isolation of the composite isolating system was beyond our measurement ability and

was found to be better then 75 dB.

3.4 The temperature controllers

The temperature controllers regulate the temperature of the transmitting and the local

oscillator lasers, see Fig. 3.6.

The lasers are selected for a matched optical frequency but, in order to get the frequency

difference within the locking range of the frequency controller, the lasers should be

tunable.

As the optical frequency of both lasers is dependant of the temperature of the laserchip, it

is possible to tune the frequency so that the frequency difference between the local

oscillator laser and the transmitting laser equals the receiver intermediate frequency.

This implies that the temperature must be adjustable over a range large enough to tune the

lasers within the frequency controller locking range and that temperature stability should

be adequate to keep them there.

27

TRA: ISM I TTER RECEIVER

IN

Fig. 3.6: The temperature controllers in the system.

DATA ~ OUT

CLOCK .- ... OUT

The temperature Gontroller consists of a temperature sensor, a control circuit and a Peltier

thermo-electric heatpump which can cool or heat the laserchip, see Fig. 3.7.

The sensor is a Siemens K19 thermistor that was chosen for its large sensitiyity and small

dimensions. This sensor is mounted on the laser heatsink as close to the chip as possible.

The temperature information from the sensor is used by the control circuit to drive the

thermo-electric heatpump.

SENSOR ~.

(L.O ) LASER •

T.E .HEATPUMP .. TEMPERATURE CONTROLLER ,- ------- -------- ,

I

~ 0 ~. I

I

-K I

I I I I

~, I" I

~ I

I I I I I BRIDGE AMPLIF. SETPOINT COMPAR. DRIVER I I I L ____________________________________ ~

Fig. 3.7: Schematic diagram of the temperature controller loop.

28

Resistance variation of the thermistor is converted to a voltage variation by the bridge

circuit. This voltage (approximately 47 mY IK) is amplified by an amplifier designed

around the Burr Brown INA 10 I low drift operational amplifier, bringing the signal level

to I V/K.

The output of the setpoint circuit, which is adjustable on a front panel dial, is compared

with the I V IK output of the amplifier.

The comparator can be used in a proportional mode, or in an integrating mode. The

proportional mode is used during the startup of the system in order to avoid large

temperature overshoots which could damage the lasers.

After an initial temperature stabilization the integrating mode is switched on by means of

a front panel selector switch.

The comparator controls the driver circuit. In order to be able to provide a positive

current (=cooling) or a negative current (=heating) to the Peltier device, a bridge design

of the driver circuit has been chosen. The driver also incorporates a current limiting

circuit to avoid overloading of the Peltier device.

3.5. The current control circuit

A block diagram of the current control circuit is shown in Fig. 3.8.

DISPLAY

~r.

REFERENCE ~

CURRENT ... I

SOURCE SOURCE p + ~

MODULATOR • ...

LASER DIODE

Fig. 3.8: Block diagram of the current control circuit.

29

The schematic diagram is given in Fig. 3.9.

LD

Rv

MOD. REF Rb 1-41----1+

Re

Fig. 3.9: Schematic diagram of the current control circuit.

The DC current through the laser Diode (LD) is detected by means of resistor Re and fed

to the input of the OpAmp. This voltage is compared with the voltage of a reference

source (LM385Z2.5). This source has a low temperature coefficient. The current can be

controlled with the variable resistor Rv. The resistor Re defines the maximum laser

current. The voltage across this resistor Re is the input signal for the laser current circuit

and is also coupled with a digital display. The LD can be damaged because of peak cur­

rents when for instance the current control circuit is switched on or off. Therefore the

circuit will increase the DC current through the laser slowly (20 Sec) when it is switched

on and vice versa when it is switched off.

The modulator is made up with a differential amplifier and consists of two HF transistors.

The current of one branch of the amplifier is added by the LD current. A third transistor

is used as a current source for the differential amplifier. This current can be adjusted with

a variable resistor and so controls the modulation depth.

The input signal for the modulator is an ECL-signal.

30

The temperature adjustment is displayed with a second digital display, which is connected

with the temperature control circuit. Two variable resistors are used, one for the zero

point adjustment and one for the gain of the detected voltage respectively.

31

Chapter 4

The transmitter

Both lasers and control circuits are equal for transmitter and receiver. These parts are de­

scribed in chapter 3. The additional parts of the transmitter, the external ASK modulator

and the FSK modulator, are described in this chapter.

4.1 The optical amplitude modulator

In case of ASK modulation direct modulation of the laser bias current it is not possible in

coherent detection systems because of laser chirping. In this case an external amplitude

modulator must be used. Because of the relatively low driving voltage the integrated

electro optical 1550 nm amplitude modulator from GEe Y-35-5370-01 is chosen which

has a bandwidth of 2.5 GHz. In Appendix 3 you find the data sheets of the modulator.

The modulator operates according to the Mach Zehnder travelling wave principle and is

made in lithium niobate. In Fig. 4.1 a schematic diagram of the modulator is shown.

RF,owce

Optical input Optical output

Fig. 4.1: Schematic diagram of the optical amplitude modulator.

32

Polarized light is launched into the optical waveguide, where it is split into two equal

parts. When a voltage is applied to the central electrode, opposing electric fields cause the

electro optic effect to advance one wave and retard the other. This is optimized by using

a travelling wave electrode structure so that the microwave and optical wave propagate

together. The light is recombined in the Y junction. In phase waves are transmitted

through the output guide and out of phase waves are configured into a higher order mode

which is lost into the substrate. In Fig. 4.2 the normalized optical transmission of the

modulator is shown as a function of the DC input voltage.

0.9

0.8

0.7

0.6

0.5

0.4

0.3

0.2

0.1

o~~~~~~~~~~~~~~~~~rr~~~~~~ -2.50 -2.00 -1.50 -1.00 -0.50 0.00 0.50 1.00 1.50 2.00 2.50 3.00

DC input voltage [V)

Fig. 4.2: Normalized optical transmission of the optical amplitude modulator

as a function of the DC input voltage.

The insertion loss of the modulator when the transmission is maximum (bias voltage is -

1. 7 V) is 7.1 dB and the extinction ratio or the maximum onloff ratio (bias voltage is 2.5

V) is 26 dB. To obtain maximum onloff ratio of the optical output in case of an AC

coupled modulation, a bias voltage of 0.4 V is necessary and the amplitude of the input

signal must be 4.2 Vtt.

33

In order to avoid reflections, the electrical input cable of the modulator must be matched

loaded. According to typical specifications the input impedance should be 50 Ohms.

However, by Time Domain Reflectometry measurement, this turned out to be about 27

Ohms. A series resistor of approximately 23 Ohms is installed to accommodate. A bias

tee is used for AC/DC decoupling.

In Fig. 4.3 the frequency characteristic of the electrical to optical responsivity is shown,

measured with an HP 8702A lightwave component analyzer. The 3 dB bandwidth is

approximately 2.4 GHz.

CH1 B/R

W/A

Cor E/O

TU

t

~ L-

log MAG

E-EC

/ III V vv

1 dB/ REF -55 dB

pJv '\ /\.

\ A/

\" / \ ~ ~

tv...

I

yw V

START .300 000 MHz STOP 3 000.000 000 MHz

Fig. 4.3: Frequency characteristic of the electrical to optical responsivity.

4.2 The FSK modulator

FSK modulation can be obtained by direct modulation of the transmitter laser bias

34

current. The frequency sensitivity of the transmitter laser used is 0.25 GHz/mA. For a

modulation index of 7, the variation in current has to be 4 rnA. This results in a ampli­

tude modulation index of less than 10 %. First a simple resistor network with a capacitor

for AC coupling was used. Since it was difficult with this solution to control the modula­

tion index, a electronic driver circuit was developed. The circuit diagram of the FSK

modulator can be found in Appendix 4.

35

Chapter 5

The receiver

In this chapter the parts of the receiver will be discussed except for the local oscillator

laser, which is dealt with in chapter 3. These parts are; the optical network, the OlE

converters, the IF section, the frequency controlling and the clock and data recovery

circuit.

5.1 The optical network

The optical network in the receiver is used to mix the received optical signal from the

transmitter with the light of the local oscillator (DFB-Iaser). In this phase diversity

receiver a {3x3}-opticaI fused fiber coupler has been used.

5.1.1 The 3x3 fiber coupler

Such a coupler can be made of optical fiber with the fused biconical taper technique.

The fabrication process is as follows:

In our coupler manufacturing setup, three matched cladding single mode fibers were

aligned after removal of the primary coating. The three fibers are stacked like a triangle

with two fibers parallel and one on top in the groove, formed by the former two. The

fibers are fused together to forme a circular area by heating them with an electric arc

flame placed below them. Then the fibers are pulled and elongated to obtain optical

coupling. During the fabrication, laserlight of 1540 nm is coupled into the upper fiber.

The optical power at the three outputs is monitored and the tapering process is halted

when the desired coupling ratio is achieved, i.e. the optical power is equally divided over

the three output fibers. Good results will only be achieved if the fibers in the fused

section have equal distances to each other.

Of even more importance are the phases between the three outputs which should be 120

36

degrees for the designed phase diversity system. For the output power of the three

branches an extra gain correction can be made in the amplifiers following the PINFET

modules. Such a correction is not possible for the phase relations. Therefore it should be

more effective if during the fabrication the phase at the output ports is monitored instead

of the optical power. In a lossless coupler, the phases between the outputs will automati­

cally be equal if the optical power is equally distributed. After packaging the coupling

ratio at the fiber output ports is checked for each branch acting as an optical input and

vice versa. In our {3x3}-coupler (No. 176) one of the outputs found to suffer from severe

optical attenuation when launching optical power in the same fiber at the input side,

resulting in an unequal light distribution at the outputs. This is shown in Fig. 5.1.

~

~ "-..

a; 3 0 "-~

:J "-~ :J 0

OPTICAL POWER DISTRIBUTION 3 x 3 COUPLER No. 176

50

40

30

20

10

0 ,A 2A

Input fibers

3A

_ OUTPUT

18

IIIiI!IIIIIIIIl OUTPUT 28

~ OUTPUT 38

Fig. 5.1: Power distribution of 3x3 coupler No. 176.

However, for the other two input fibers, the power division was equally. Since a phase

diversity receiver only uses two input fibers the latter should not be a problem. Unfortu­

nately, the measured phase differences between the three output ports were found not to

be exactly 120 degrees. The optical power launched at the input of a coupler from Sifam

(UK) was fairly equally divided over the three output fibers. This applied for all of the

three inputs. Measurements showed 33 % + 5, as is shown in Fig 5.2. The excess loss

37

was 0.1 dB.

~

*' ij; 3 0 "-~

~

"-~ 8

OPTICAL POWER DISTRIBUTION 3 x 3 COUPLER SIFAM

50,-----------------------------,

40 -30 mIIIIlIIiIl

20 ~

10

0 1A 2A 3A

Input fibers

OUTPUT 1B

OUTPUT 2B

OUTPUT 3B

Fig. 5.2: Power distribution of the 3x3 coupler of Sifam.

The phase differences also showed good results namely 120 + 9 degrees, see Fig. 5.3.

PHASE MEASUREMENT COUPLER SIFAM

360,------------------------------,

1 270

m 180

~ :ll &. 90

o 1A-2A 2A-JA 1A-JA 18-28 28-38 18-38

Irput fibers

_ phase

1B->2B

IIIIIIIIII!III! phase 2B->3B

~ phase 3B-> 18

I2i88:3 phase 1A->2A

f22Zl phase 2A->3A

~ phase 3A->1A

Fig. 5.3: Phase differences of the Sifam 3x3 coupler.

38

For phase measurements the setup from Fig. 5.4. was used.

-3 dB coupler Q:>tlcal Network: Photo-Olodes

Fig. 5.4: Phase measurement setup.

The lightsource is a HeNe-laser for 1520 nm followed by an optical isolator of 32 dB.

The AOM shifts the optical frequency by 80 MHz. The output currents of the photodetec­

tors contains the difference frequency of the two input wavelengths. The phase differences

between the output ports were measured by a Vector Volt Meter (VVM), each time using

two input ports.

5.1.2 Phase diversity network measurement set-up

The phase diversity network is an important component in a fiber optic phase diversity

coherent communication system.

Combining both the received optical signal and the output of a local oscillator laser results

in multiple output signals with a certain fixed phase relation, for example 7r12 between the

outputs in the case of a four branch network, while the power distribution should be

equal.

Fig. 5.5: The Phase Diversity Network in (a part of) a coherent receiver.

Imperfections in the network could lead to both phase and splitting-ratio errors. Ratio

39

errors however can be corrected for in the electronic processing part of the system, but

phase errors wi1llead to receiver performance degradation.

For this reason the measurements on the phase diversity networks focussed on the phase

relations of the output signals.

Phase diversity networks can be manufactured by several methods, for example by means

of a fused biconical taper technique. These networks feature a low excess-loss and are

easy to use in a fiber optic system, as the optical signals stay inside a guided(fiber)

system.

Measuring the phase relation of the outputs is fairly straight forward. A simplified layout

is shown in Fig 5.6.

LASER

sa~

Sb

Fig. 5.6: A phase-relation measurement method.

S1

S2

S3

S4

The output of a laser is fed both direct to an input branch of the phase diversity net­

work(Sa), as well as via an acoustic optic frequency shifter, that shifts the optical

frequency by 80 MHz with respect to the other input branch(Sb). The outputs branches

(SI..S4) are connected to photodetectors, where the optical signals produce a difference

frequency. The resulting photodetector currents are amplified and measured by means of

an oscilloscope or a vector voltmeter.

The Phase Diversity Networks where realized by T.U.D. by means of a planar technolo­

gy. Although the measurement is straight forward, the practical implementation with

regard to the planar network is somewhat more complicated:

40

1) This planar technology doesn't allow to couple fibers direct to the network

branches, therefore we used a prism coupling method at the input side, and

a lens coupling at the output side of the network.

2) Although earlier experiments have been done with (visible) 632 nm ReNe

laserlight, the current planar circuit are designed for (invisible) 1523 nm

Infrared light, that makes optical alignment a lot more difficult.

3) Due to high loss in the coupling area, we encountered problems with

regard to the signal-to-noise ratio of the photodetector signals.

Besides we found that optical reflections degraded the stability of the

measurements.

The most suitable method to couple light into the planar circuit is the prism coupling

technique. Coupling light out of the planar circuit to the photodetectors can be done by

means of a lens coupling, see Fig. 5.7.

Microscope objective

Planar circuit

P, ism COUp I I ng Lens coup ling

Fig. 5.7: Planar circuit coupling methods.

If we want that adjusting the output coupling does not influence the input coupling we

have to fix the planar circuit itself and move the input beam (both angle and location on

the planar circuit) and the output lens coupler. This implies that the total optical process­

ing should be mobile, or that a flexible interface, between the processing setup and the

prism coupling optics, should be used, see Fig. 5.8.

41

Fig. 5.8: Input coupling set-ups.

As the mechanical alignment of the optical processing part is rather critical, the last

solution has been chosen to avoid alignment stability problems.

Optical processing.

The light source, a Melles Griot type 05-LlP-171 infrared HeNe laser, delivers a 1523

nm. beam via an optical isolator to a lens-ferrule equipped single mode fiber.

The lens-ferrule can be aligned to the laser beam by means of a custom made alignment

facility, in order to obtain a maximal efficiency.

The fiber is connected to an input branch of a 3dB coupler that splits the power in two

equal parts. The other input branch is connected to a 632 nm SpectraPhysics HeNe laser

to couple some visible light in the optical path for alignment purposes. One output of the

coupler is connected to an acoustic optical frequency modulator, that shifts the optical

frequency by 80 MHz.

Both branches pass an optical attenuator and a polarization controller.

The attenuators reduce measurement instabilities due to optical reflections, as reflections

have to pass the attenuator more than once before entering the photodetectors.

The fibers leaving the polarization controllers are led to the beam delivery system which,

is mounted on the vertical rotation stage. This beam delivery system consists of two

microscope objectives that collimate the beams, a 45° mirror, a beam-combiner and a

cylindrical lens (f=50 mm).

Initialy a non-polarizing beam splitter was used as beam-combiner but as this splitter

introduced a 3dB excess power loss, we replaced it by a beam-combiner that consists of

selective gold coated prisms that are cemented together.

42

By proper aligning, both beams will focus through the cylindrical lens and the prism on

the coupling area of the planar circuit, see Fig. 5.9.

.. . j;..

Fig. 5.9: Input and output coupling area's.

output area

The prism that couples the light in the planar circuit is positioned by hand and not fixed.

In order to obtain coupling an index matching liquid has to be added between the prism

and the planar circuit surface.

The light that leaves the planar circuit at a cleaved edge is collected by a microscope

objective and, enlarged, projected.

A movable mirror that is mounted behind the objective makes it possible to project either

on the photodetectors, or on a infrared sensitive CCTV camera (Hamamatsu) for

alignment purposes, see Fig. 5.10.

As the distance between the projected output spots is small, in spite of magnification, the

spots are spatial separated by means of an image-splitter. (a 45° mirror, one spot on the

mirror, the other three just passing it).

So one spot is always focussed on photodetector no. 1 while the other three can be

scanned by moving photodetector no.2. This arrangement allows phase measurements

43

between output 1 and output 2,3 and 4, see Fig. 5.11.

I mage-sp I j t ter

Fig. 5.10: I.R. camera and photodetectors.

o L Photodetector 1 ~ 1111111111'

(3 , ~< : : -11 111111111 [I]

Photodetector 2

Planar circuit Image-splitter

Fig. 5.11: Image splitting.

Electronic layout.

The lasers have their own powersupplies and did not need further modifications. The

optical frequency modulator is connected to an Isomet driver. The frequency modulator

and the driver had to be adjusted for maximal optical output power.

44

The photodetectors consist of a Philips CPF30 PIN photodiode and a Signetics NE 5211

Transimpedance amplifier.

These two devices are build together inside a casing that provide protection against

interference from outside.

The transimpedance amplifiers are followed by Trontech type WIGE 1O-PS-22 amplifiers

that provide ample signal to the following Hewlett-Packard HP 8508A vector voltmeter.

Due to the high optical losses in the setup and in the planar circuit the signal-to-noise

ratio turned out to be to small to lock the vector voltmeter reliable. Inserting bandpass

filters (BW=20 kHz) between the Trontech amplifiers and the HP vector voltmeter

improved the "locking-range" of the vector voltmeter by more than 30 dB, see Fig. 5.12.

Attenuators Pol. Control

Coup I er

000

I~ Iver !

Photodetcctcr 2

In'Ia~e-SPlltter

.",'d,.c·1 V II V I >, ".cs I Jl I [J[J

Vector Voltmeter

Osc; I loscape

Fig. 5.12: The measurement set-up.

S.2 The PINFET OlE convertor module

This module is situated between the optical network (three optical outputs), and the

I.F.section (three electrical inputs). The module also provide a signal to the frequency

controller. The task of the module is to convert the optical signals to electric signals

45

which can be processed by the following modules, see Fig. 5.13.

TRANSMITTER RECEIVER

~=-l ~

~ E:J

DATA IN

Fig. 5.13: The PINFET DIE converter module in the system.

DATA '~OUT

CLOCK ~OUT

Three optical FC/PC connectors, which are mounted on the front panel, facilitate the

transfer of the light from the patch cords to the pigtails of the B. T &D. PINFETS type

PDC 2200 where the Opto/Electric conversion takes place, see Fig. 5.14.

PINFET OlE CONVERTER ----------------------------1

I

[D> I [I i SPL I I I Ell 1 MSI\ 0 18~,

I

I

-~i~1 II ~ IE]' ~ ___ 2=i _____________________ J

Fig. 5.14: Schematic diagram of the PINFET module.

..... ~

.. ~

.... ~

,.~

More information about the PDC 2200 module can be found in the appendices.

The PDC 2200 module is followed by amplifiers based on an Avantek MMIC type MSA

0785.

46

These amplifiers increase the system gain by 12 dB, and are connected to resistive power

splitters. Their outputs are located on the front panel.

So each of the three branches have two outputs, from which one is used to drive the

I F section. The remaining outputs are used for the frequency controller and for monitor

purposes. Fig. 5.15 shows the overall frequency response of the channels.

Branch I

..

.- '" I_I.,,, . , .... ,. '''1

,

i I

',.",'

, ..

"UI , """ """ u'"'- "" ..

Branch 2

Fig. 5.15: Frequency responses.

o;H, ~'" loO ~.r,

;J"I'! I 1

• I I I

Branch 3

In order to compensate for unbalance between the optical branches, for example due to

the optical network, it is possible to adjust the gain of each channel independently by

means of an internal gain control.

5.3 The J F section

The IF section receives the three signals from the PINFET OlE converter module, see

Fig. 5.16.

After filtering, amplification, detection, and addition of the three branches there IS a

baseband signal available to the clock and data recovery module, see Fig. 5.17.

The first stages of the I F section are the lowpass filters. These filters, having a cutoff

frequency of about 750 MHz, reduce the noise contribution of the preceding stages, but

don't degrade the signals. All three filters are adjusted for an equal response.

Fig. 5.18 shows the filter frequency responses.

After filtering, the signals are amplified by Avantek IVA 05118 variable gain amplifiers,

47

I ~M·I::,Iv111 T l~ RECEIVER

6---.-reMP

1 l~~lr_' _

DATA IN

~~~~v I

Fig. 5.16: The I F section in the system.

I.F SECTION

DATA .OUT

CLOCK · ... OUT

r---------------------------------------~

IL P FILTER AMPL I F I ER SQUARER ADDER P. D. FILTER :

I I I I I I I I I I I : .. ~

i C ';:Jt .... ® ~1~c~iT P DAMPLIFIERi I I 1 ______ ----------------------------------"

Fig. 5.17: Schematic diagram of the I F section.

providing a gain between 0 and 26 dB to the signals. The gain of the amplifiers is

controlled by an automatic gain control circuit which will be described later on. The

outputs of the amplifiers, both inverted and non-inverted, are connected to the R.F. and

L.O. inputs of Avantek lAM 81018 active double balanced mixers, which are used here

as square-law devices. Fig 5.19 shows the input signals of the I F section (simulated 140

Mb/s ASK 0-1 bit sequences) and the output signal of one squarer. (lower trace)ln order to

restore the original ASK signal, the output of an three squarers have to be added. This is

simply done by means of a resistor network. The network is designed for a maximum

isolation between inputs because, the mixcr outputs turned out to be load sensitive. A post

48

i--i--r-t-- .. - . +-+---f-f-----i

:;,-.\--\ f- --+\ .. --- -'-r- lI' : ; I' • t-- --+--- '

__ LL.l, __ LL 1'1.1 STAAT .300 000 I<H1 SlOP t 000.000 DO" _,

Fig. 5.18: I F filter response

A"5,000 no -l~ - -,--------- --- 5"4.000 nS , ~44 "'

_ "on non, '''' AA I £V' ,,"""", '"V""," """" \MAl ,AMnMI nAnnAn ""

,'" ,,,'Annl rUllVv,VVI ~., onl\l'J\J\ l~MM ~AMM fiA~~'v .'." ' I U~JLlllIV'

~IM V\fiIWV IIiMnoi """: - "" " "ru

J V vvv, rVVUUU' uUVUUV' '" '

---

- ,,00 u 0.00 o , o "'

1-- - n~~~\ ~\tt1 - --- - ~\~~I NM~)I~ . __ .

'", 1~1 "' nv ' ~~ l"IU-~ J!iIW ~ ~vw 1\t\i1A~ . . .AlliVVI - ",\1 IVV I J ,I I rvv----,

.- - - _.

- -- .... . - --~- --_._-_. - - .-~

Fig. 5.19: Simulated I F input signals.

detection filter follows the addition, providing an optimal signal-to-noise ratio to the

signal.

Restricted space on the J F section printed circuit board allowed only a rather simple

filter configuration, for which the cut-off frequency has been chosen at about 150 MHz.

If necessary more elaborate filtering can be done between the J F section and the clock

and data recovery module.

Fig. 5.20 shows the post detection filter frequency response.

49

PDF REF -10 0 dBm 10dB!

ATT 10 dB A norm.>a B blank

AVG A 50

~ MKR 155.0

RBW I MHz

VBW I MHz

SWP 50 ms

"0 '\

Hz \

1\ \

\

CENTER 250.0 MHz

- -

MK I 5.0 MHz - .97 d

\. 1\ \ ./

SPAN 500 MHz

Fig. 5.20: Post detection filter respons.

f'

The output of the post detection filter is connected to the post detection amplifier.

This amplifier is necessary to bring the signal to a level which is compatible with the

clock and data recovery module. (0.4-\.5 Vtt).

The design is simple, an Avantek MSA 0785 MMIC is followed by an Avantek 0385

MMIC. The output data is available at a front panel SMA connector.

The D. C. components from the mixer outputs are added and used by the automatic gain

control circuit. This a.g.c. circuit has two front panel controls, a switch to select either

a.g.c. mode or manual mode, and a dial to adjust the data output level.

In the a.g.c. mode the added d.c. components from the mixer outputs are compared with

the setpoint from the level-dial. The result is a constant output level that is, within the

gain range of the variable gain amplifiers, independent of fluctuations of the receiver

input signal and fluctuations of the local oscillator laser powerlevel.

The gain of the amplifiers is proportional to the level-dial setting when the manual mode

is selected.

To check the performance of the I.F. section a few measurements have been carried out

by means of a simulated 140 Mb!s ASK modulated P.R. B.S. data signal.

The plot in Fig. 5.21 shows the eye pattern of the signal at the input channels of the I F

section. These signals simulate the situation where the optical frequency of the transmit-

50

ting laser differs 548 kHz from the optical frequency of the L. O. laser. The pseudo

random generator is synchronized with the difference frequency (548 kHz) in order to

get a stable eye pattern.

;' ',(Jooo wI () 00000 ;' >,0000 [J'.

I., ..... '· ........ ··'!'"

Fig. 5.21: I F input 'EYE' pattern.

The next plot (Fig. 5.22) zooms in on one of these inputs, and gives a more detailed look

at the eye pattern, the lower trace shows a magnified part of the upper trace.

Fig. 5.22: Detailed look at the 'EYE' pattern.

The output of the I F section, before post detection filtering, is shown in Fig. 5.23. This

plot demonstrate that the reconstruction of the baseband signal has been done correctly.

The eye is 'open' and there is no noticeable channel unbalance. Fig. 5.24 shows the eye

51

10.000 I1S 0.00000 s 10.000 n5

Fig. 5.23: I F output before post detection filtering.

pattern after post detection filtering. The signal-to-noise ratio is increased, the eye is

'open', some intersymbol interference can be noticed. The last plot (Fig. 5.25) shows the

-10.000 n13 0.00000 s 10.000 flS

I ..• . . ... :. ;.:.:;<;~;.';';::" .---+-: .-: .. ::.-.. ::-+ ;:\;«-<-. -+-----+----1

Fig. 5.24: The 'EYE' pattern after post detection filtering.

spectrum of the output signal after post detection filtering.

PRBS 2'7-1 NA PDF REF -19.9 dBm !OdB/ I

i

\ I iii 1 kHz I

SWP L' i 10 5 .. 1.

, ! :

J __ J CENTER 250.0 MH7

52

SPAN 100 MH7

Fig. 5.25: The spectrum of the output signal.

5.4 The frequency controller

The frequency of the optical signal of the lasers which are used, is very susceptible to

variations in temperature (10 - 20 GHz/K) and to variations in current (0.5 GHz/mA).

Therefore it is not sufficient to stabilize the temperature of the lasers and the current

through the lasers to obtain a stable difference frequency or Intermediate Frequency (IF).

In the receiver a Frequency Locked Loop (FLL) has to be used in order to synchronize

the frequency of the local oscillator laser with the frequency of the received optical

signal.

The FLL was originally meant to operate at an IF of zero which gives certain specific

problems. Later on the IF is changed to 300 MHz. The design is however very flexible

because it is software controlled, so only the program had to be altered. First the FLL for

zero IF will be described [13].

53

5.4.1 Principles of the frequency locked loop

In this case a frequency discriminator for controlling the local oscillator laser is not

suitable. With zero IF the following problem occurs. If the frequency of the local

oscillator laser is too high, the IF is too high. When the frequency of the local oscillator

laser is too low, the IF is also too high. A frequency discriminator cannot distinguish

between these two situations, so in this way it is not possible to obtain an unambiguous

control signal for the local oscillator laser.

This problem is solved in the following way.

IF signal

... -~o:;:;-ptical"l---t network trans~tter laser

-L-__ --'-photodiode

frequency controller

~I DAC. micro- k---I

controller H '---_---' L __ -'

counter ,--------, I

presca.ler

,---------,II

r------------

local oscillator laser

Fig. 5.26: Block diagram of the frequency locked loop.

To detect the IF, a prescaler and a digital counter is used, see Fig. 5.26. A prescaler is a

high speed frequency divider with an analog input and a digital output. The output signal

of the prescaler is applied to the clock input of a digital counter. The counter is controlled

by a microcontroller (single chip processor). By counting the clock transients in a certain

time interval a value for the IF is measured. The microcontroller can read the value of

the counter and the microcontroller controls the frequency of the local oscillator laser via

a Digital to Analog Convertor (DAC).

The program of the microcontroller consists of two procedures. A capture procedure and

a tracking procedure. By means of the capture procedure a control voltage for the local

oscillator laser is obtained for approximately zero IF. After this procedure the tracking

loop will minimize the IF by means of a trial and error algorithm. In Fig. 5.27. a flow

diagram of the program is shown.

54

capture procedure

Ian niue counter

t.dd .tep to

.. lue lor DAC

rnd counler

K----"YE:::.'-<: haa vallie of counter becOme .maller?

NO

iuverillep

Fig. 5.27: Flow diagram of the program for zero IF.

After the capture procedure the value of the counter is saved. Then the frequency of the

local oscillator laser is changed by a small step by changing the digital input signal of the

DAC. Again the value of the counter is read. If the value of the counter has become

smaller and thus the IF has become smaller, then the frequency of the local oscillator

laser is changed by the same step. If not, the step is inverted and thus the direction is

changed. This procedure is repeated in an endless loop.

5.4.2 The realized frequency locked loop

The described principle is realized with a divide-by-64 prescaler, an 8-bit counter, an

8-bit microcontroller and an 8-bit DAC, so only 4 integrated circuits in total. With this

FLL, the two DFB lasers are synchronized, see Fig. 5.13. Because of the speed of the

controller, the frequency of the local oscillator laser is controlled, via the current through

55

the laser, which in its turn is controlled by the DAC. The tracking range of the FLL is

0.5 GHz. This results in a variation of the optical power of the local oscillator laser of

0.1 dB, which is not dramatic. The frequency steps are about 2 MHz. These steps are

small compared to the difference spectrum of the lasers which is about 40 MHz

(FWHM). The tracking procedure of the program of the microcontroller consists of 20

lines of assembly language. The average time of a loop of the tracking procedure is about

35 (.Is, so maximum about 28,000 iterations per second can be made.

5.4.3 Conclusions

It has been shown that it is possible to make a Frequency Locked Loop with a

microcontroller for a optical receiver with zero Intermediate Frequency. The use of a

microcontroller makes the FLL fast, simple and flexible. It is possible that after long

periods of time the difference of the frequencies of the lasers becomes too large, so that

the FLL gets out of range. It is easily possible to modify the FLL so that the

microcontroller also changes the temperature of the local oscillator laser for long term

stability. Because a trial and error algorithm is used which searches for a minimum, the

FLL is independent of the modulation of the received optical signal or linewidths of the

laser diodes and there are no adjustments necessary in the hardware of the FLL.

By changing software it is also possible to choose an IF unequal to zero. In Appendix 5

the listings of the 8048 source code for zero IF and an IF of 300 MHz are given.

5.5 The clock and data recovery circuit

The baseband signal at the input of the clock and data recovery circuit is received from

the IF section, while at the output regenerated data and clock signals are available, see

Fig. 5.28.

The concept of the module is to use a phase-frequency comparator to compare a reference

frequency to a V.C.O. frequency and generate a signal proportional to the phase

difference between the to sources. After lowpass filtering, amplification, and integration

this output will provide an error signal to correct the V.C.O ..

The baseband signal from the IF section provide the reference, and the V.C.O. will be

TRANSMITTER

DATA IN

56

EL L;-r

TEMP CONTR,

RECEIVER

Fig. 5.28: The clock and data recovery circuit in the system.

DATA ..• OUl

CLOCK ... OUT

locked to it. The V.C.O. output then becomes the clock for the following data processing

equipment.

A difficulty arise due to the fact that the data receiyed has no continues zero to one

transitions, but series of zeros and ones are possible.

A conventional phase comparator will interpret the absence of data transitions as a

V.C.O. leading the reference in phase and will attempt to decrease the V.C.O. frequency

in order to obtain phase-lock. This will result in severe phase-jitter in the case of a long

run length.

In order to avoid this we used a clocked scheme (Fig. 5.29).

The incoming data is applied to a pair of differential comparators followed by a pair of

flip-flops followed by the phase-frequency comparator.

The input signal to one of the comparators is delayed a quarter bit with respect to the

other comparator input. The undelayed signal is applied to the 'D' input of a flip-flop.

This flip-flop is clocked by the V.C.O. and regenerates at its output the data input

delayed in time by the V. C. O ..

The delayed comparator output is applied to the 'set' input of the other flip-flop, which

generates the reference input of the phase-frequency comparator. The output of both flip­

flops are applied to the Motorola MC 12040 phase-frequency comparator. When the

inputs are in phase it will result in equal amplitude and duration pulses on the comparator

outputs, which, when subtracted and integrated by the loopfiiter, will yield a zero error

signal for the V.C.O ..

57

B Oi fferen

.~ tial

~ 5 F lip

compar Flop q

~-~ : c

OATA IN Phase-fr-eq, .

Differ-en : V_:~~~ar ,

.~ tial .. ~ d

Flip Flop q ..

compar. e

~

d

0 nEeov, ~\~~ c

v. c.o: ," Loop-

DATA ~ .. q ampl if. ~. OUT In +f i Iter

CLOCK OUT ~ ..

Fig. 5.29: Schematic diagram of the clock and data recovery circuit.

In this situation the clock will sample the data midpoint in its stable state. If the data

remains at either the low or the high state both outputs of the phase-frequency comparator

remain at the non error, low, state, providing no correction signal to the V.C.O .. In this

situation the V.C.O. frequency is maintained.

Data regeneration is done by clocking the received data in a D-type flip-flop. The output

of this flip-flop is available at a frontpanel connector. The positive clock transition is

timed at the middle of the regenerated data bits, to provide a maximum timing margin.

The V.C.O. circuit consists of a Motorola MC 1648 ECI voltage controlled oscillator,

and an external parallel tank circuit consisting of an inductor, a capacitor and a varactor

diode. The inductor and the capacitor were selected for a V.C.O. tuning range of

approximately 135 to 145 MHz. The output of the V.C.O. is available at a connector at

the front of the module.

The loopfilter was designed according to the design guides given in the Motorola data

sheets. After lowpass filtering the Up and Down signals are summed by a LF 356 wide­

band amplifier and integrated by an other LF 356 amplifier that controls the V.C.O.

tuning voltage.

During system tests it turned out that the MC 12040 phase-frequency comparator did not

58

function well on the given bitrate. The frequency comparator gave ambiguous frequency­

error information to the V.C.O. when the V.C.O. was near phaselock.

The MC 12040 had to be temporary replaced by a phase comparator consisting of an

exclusive-or gate. Due to this modification the V.C.O. has to be manual adjusted to

obtain phaselock; for this a frontpanel frequency control is provided.

The phase comparator can be exchanged as soon as we obtain a phase-frequency compara­

tor which is functional at 140 Mb/s.

59

Chapter 6

Experimental results

In this chapter the measured IF spectra for ASK and FSK modulation, the Bit Error Rate

measurements and the TV codec are described successively.

6.1 IF spectra

The linewidths of the DFB lasers have been separately measured by means of a self

homodyning set-up [33]. These results are discussed in section 3.1. At 4 mW output

power the linewidth was approximately 20 MHz FWHM, for both lasers the same.

Difference spectra of both lasers have been measured also at 4 mW output power of each

laser with zero IF and an IF unequal to zero.

The results are in Fig. 6.1.

The output signals of the lasers were mixed with a {2x2} fiber coupler and detected by a

PINFET DIE converter of BT&D. The spectrum of the output signal of the PINFET was

measured by a HP 71400 spectrum analyzer. In Fig. 6.la the optical frequency of both

lasers differs approximately 70 MHz so the IF spectrum lies around this frequency. The

width of the spectrum at 3 dB is 50 MHz. Theoretically this should be 40 MHz but

probably because of optical reflections the spectrum has become widen. In Fig. 6.lb the

beatspectrum is shown with equal optical frequencies. The 3 dB bandwidth is at 30 MHz

while it should be 20 MHz. This can be explained by the fact that the impact of reflec­

tions is greater with homodyne detection than with heterodyne detection. When we have

fore instance an isolated laser with a long fiber, refections can cause a spectrum around

zero because of (self) homodyne detection of the reflections with the original optical

signal.

a)

b)

60

-RL 33 ee dB m RTTE lB B

.00 dB/O V MIe OWAVE

I ~\j"

II',

I~ ~J ~ ~

~ START 0 Hz RB 1.BB MHz -VB 3,00 kHz

STOP lBB. B MHz ST lBB.0 osec

RS 1.00 tlHz 3.00 kHz ST 100.0 msee

Fig. 6.1: Difference spectrum of the lasers a) with equal optical frequencies

bl with difference frequency of approximately 70 MHz.

61

In Fig. 6.2 the spectrum of the IF signal is shown for the zero IF case, with and without

FLL. The spectrum analyzer was in the maximum hold mode for 1 hour. In this mode

only maximum values are saved and displayed after the time period of the measurement.

There are hardly differences between the measured IF spectrum with FLL of Fig. 6.2a

and the spectrum of one of the lasers with a self homodyning measurement.

-RL 27 BB dB 0

RTTE lB B HB dB/O V

a) ~ ~ I A Jl II v' y~

r'tu~ , I~

.~

STRRT B Hz RB 3,B8 MHz -VB 18.8 kHz

-RL 27 00 dB m ATTE 10 B 3,00 dB/O V

b)

~ I~,

~

~, ..... ~ i' ~

"

START 0 Hz RB 3.00 MHz -VB 10.0 kHz

f\. '''f,

~

MIC O~RVE

~ ...... ~,

STOP 500.0 MHz ST 50.80 IDsec

MIC O~AVE

low". " ~

STOP 500.0 MHz ST 50,00 osec

Fig. 6.2: IF spectrum for the zero IF case a) without FLL and b) with FLL.

62

In Fig. 6.3 IF spectra are shown in case of 140 Mb/s PRBS (Pseudo Random Binary

Sequence) a) ASK modulation and b) CPFSK modulation with modulation index of 3. The

3 dB bandwidth in case of ASK modulation is 40 MHz.

a)

b)

RL 30 00 dB_ -, -

~TTE 10 B .00 dB/O V Mie OWAVE

~ ~,

r I' ~

~ -

--START B Hz RB 1.00 MHz -VB 3.00 kHz

-RL 26 00 dB _

ATTE 10 B 2.00 dB/Q V

\ \ ""'~

1\ I '-\ J \ ·rr· "'"

START 0 Hz RB 3.00 MHz -VB 10.0 kHz

1 \

~ STOP 100.0 MHz

ST 100.0 _sec

MIe QUAVE

STOP 1.000 GHz S1 100.0 _sec

Fig. 6.3: IF spectra is case of 140 Mb/s PRBS a) ASK and

b) CPFSK mOdulation with modulation index of 3.

63

6.2 BER measurements

In Fig. 6.4 and 6.5 the eye-diagram after post detection filtering and the Bit Error Rate

versus the optical power received For the ASK case are shown, respectively. The

sensitivity at a BER of 10.9 is -47.2 dBm, which is 4.2 dB better than the value in the

specifications.

,----,----,.----,.----,---- ~-------,----- ---- -~-- ~~----

I----+--+----+--+----+----t---+---j--~ -.- ---.-~

TiII,e!) ••• Me1" 2.00 n./dlv

Fig. 6.4: Eye-diagram after post detection filtering.

64

8ER versus OPTICAL POWER RECEIVED

-3 x

x -4

-5 x x

'" It w m '-'

-6 (Jl

0

-7 x

-8 x

-9

-10 '--_..L-_---'-_-'-_---'-_--' __ L----_..L-----l

-55 -54 -53 -52 -51 -50 -49 -48 -47

OPTICAL POWER [d8m 1

Fig. 6.5: Bit error rate versus the optical power received in case of ASK modulation.

6.3 The video codec

For demonstration purposes a 140 Mb/s video codec (coder-decoder) has been developed.

The coder part converts a standard analog video signal into a 140 Mb/s data signal. This

65

signal can be used as an input data signal for the transmitter of the phase diversity

system. The clock and data output signals of the receiver of the phase diversity system

are to be used as input signals for the decoder part of the video codec. The decoder part

converts the received 140 Mbps data signal back to the original analog video signal.

DC Video

Sync_ Scram-Video Multi- Data

res-ADC bIer out In plexer

torer AudlO----';

CLKB ,

CLK 14MHz 14QMHz

Timing

Fig. 6.6: Block diagram of the coder.

In Fig. 6.6. a block diagram of the video codec is shown. The analog input video signal

is amplified and clamped by a DC-restorer circuit and then fed into the video Analog to

Digital Converter (ADC). The sample rate of the ADC is 14 MHz (CLKS). The 8 bit

parallel output of the ADC is given into the multiplexer. Here one bit of a 14 Mbps serial

data stream for audio signals and one synchronization (sync) bit are added. After parallel

to serial conversion the 140 Mbps serial data stream is scrambled by a 5-th order basic

scrambler to avoid baseline wander in the receiver and to give the baseband signal always

enough transitions for the clock recovery circuit. In the decoder part of the codec the

reverse operations take place.

Figure 6.7. shows a block diagram of the video decoder part. The 140 Mbps serial data is

first descrambled by a 5-th order descrambler and then shifted into a shift register. An 11

bit word is loaded into flip flops and a sync detection circuit detects whether the first and

the last bits are sync bits. If so the frame is synchronized and after 10 clock periods a 11

bit word of the shift register is loaded again. If not the frame is not synchronized and

after II clock periods an II bit word is loaded into the flip flops. This shifting of the

frame continues until synchronization occurs. The video bits of the flip flops are shifted

66

into the Digital to Analog Converter (DAC). The output of the converter is buffered.

De- De- h-sync.P.

Data Video Vide o scram- multi- Buffer

out In DAC bIer plexer !-">Audi

eLK I

140MHz

eLK In

Timing

Fig. 6.7: Block diagram of the decoder.

Some rough specifications of the video codec are:

Analog video input/output voltage

Analog video input/output impedance

Digital sampling rate

Coding

Digital word structure

Scramblingl descrambling

Digital transmission rate

Digital input/output voltage

1.0 Vpp

75 ohms

14.0 MHz

8 bit linear PCM

8 bit video, I bit audio, I bit sync

5-th order

140.0 Mbps

ECL-Ievel NRZ

67

References

[1] Siuzdak, J. and W. van Etten

BER evaluation for phase and polarization diversity optical homodyne receivers

using noncoherent ASK and DPSK demodulation,

Journal of Lightwave Technology, Vol. 7 (1989), no. 4, p. 584-599.

[2] Kazovsky, L. and P. Meissner, E. Patzak

ASK multi port optical homodyne receiver,

Journal of Lightwave Technology, Vol. LT-5 (1987), no. 6, p. 770-79l.

[3] Garrett, I. and G. Jacobsen, E. Bodtker, R. Pendersen, J. Kan

Weakly coherent optical systems using lasers with significant phase noise,

Journal of Lightwave Technology, Vol. 6 (1988), no. 10, p. 1520-1526.

[4] Betti, S. et al.

Effect of non-Lorentzian lineshape of a semiconductor laser on a PSK coherent

heterodyne optical receiver,

Electron. Lett., vol. 23 (1987), no. 25, p. 1366-1367.

[5] Franz, J.

Optische iibertragungssysteme mit iiberlagerungsempfang,

Berlin: Springer, 1988.

[6] Foschini, GJ. and G. Vannucci

Characterizing filtered Jightwaves corrupted by phase noise,

Trans. on Information Theory, vol. 34 (1988), no. 6.

[7] Kazovsky. L.G.

Impact of lasel· phase noise on optical heterodyne communication systems",

J. of Optical Communications, vol. 7 (1986), no. 2, p. 66-87.

[8] Shanmugam, K.S.

Digital and Analog Communication Systems,

New York: Wiley, 1985.

[9] Krom, W.H.C. de

Optical Coherent Phase Diversity Systems,

Eindhoven University of Technology, 1992. Doctoral Dissertation.

68

[10] Kazovsky, L.G. and O.K. Tonguz

ASK and FSK coherent lightwave systems: a simplified approximate analysis,

J. Lightwave Technol., vol. 8 (1990), no. 3, p. 338-352.

[II] Foschini, G.J. and L.J. Greenstein, G.V. Vannucci

Noncoherent detection of coherent lightwave signals corrupted by phase noise,

IEEE Trans. on Commun., vol. 36 (1988), no. 3, p. 306-314.

[12] Thylen, L.

Integrated optics in LiNbO recent developments in devices for telecommu­

nications,

J. Lightwave Technol., vol. 6 (1988), no. 6, p. 847-861.

[13] Boom, H. van den

A Frequency Locked Loop for a Phase Diversity Optical Homodyne Receiver.

Proc, 16th European Conference on Optical Communication, Amsterdam,

September 16-20, 1990. Volume 1.

Amsterdam: PTT Nederland and Philips Research Laboratories, 1990. P. 377-380.

[14] Siuzdak, J. and W. van Etten

Heterodyne ASK multiport optical receivers using postdetection filtering,

Journal of Lightwave Technology, Vol. 8 (1990), no. 1, p. 71-77.

[15] Krom, W.H.C. de

Impact of local oscillator intensity noise and the threshold level on the perfor­

mance of a {2x2} and {3x3} phase-diversity ASK receiver,

Journal of Lightwave Technology, Vol. 9 (1991), no. 5, p. 641-649.

[16] Krom, W.H.C. de

Sensitivity degradation of an optical {2x2} and {3x3} phase-diversity ASK

receiver due to gain imbalance and nonideal phase relations,

Journal of Lightwave Technology, vol. 9 (1991), no. II, p. 1593-1601.

[17] Etten, W. van and 1. van der Plaats

Fundamentals of Optical Fiber Communications,

London: Prentice Hall, 1991.

69

[18] Krom, W.H.C. de

Impact of laser phase noise on the performance of a {3x3} phase and

polarization diversity optical homodyne DPSK receiver,

Journal of Lightwave Technology, Vol. 8 (1990), no. 11, p. 1709-1716.

[19] Etten W. van and H. van den Boom

Fused Biconical Taper Optical Couplers. Procedings symposIUm Sensors and

Actuators, Enschede, The Netherlands, November 15-16, 1990. Ed. by A.

Driessen.

Deventer: Kluwer, 1990. P. 213-216.

[20] Etten, W. van

Coherent Optical Fibre Communication,

Tijdschrift voor het NERG, part 55, Volume 3 (1990), p. 89-97.

[21] Siuzdak, J. and W. van Etten

BER performance for CPFSK phase and polarization diversity coherent optical

receivers,

Journal of Lightwave Technology, vol. 9 (1991), no. 11, p. 1583-1592.

[22] Soldano L.B. and M.K. Smit, A.H. de Vreede, 1.W.M. van Ufelen,

B.H. Verbeek, P. van Bennekom, W.H.C. de Krom, W. van Etten

Newall-passive 4x4 planar optical phase diversity network. Proc. 17th European

Conference on Optical Communication, Paris, September 9-12, 1991. Post­

Deadline Papers.

Paris: Societe des Electriciens et des Electroniciens, 1991. P. 96-99.

[23] Henry, C.H.

Phase noise in semiconductor lasers,

1. Lightwave Technol., vol. 23 (1986), p. 298-311.

[24] Kuwahara, H. and M. Sasaki, N. Tokoyo

Efficient coupling from semiconductor lasers into single-mode fibers with

tapered hemispherical ends,

Applied Optics, vol. 19 (1980), no. 15, p. 2578-2583.

70

[25] Groten, M. and W. van Etten

Laser line width measurement in the presence of RIN and using the

recirculating self heterodyne method,

Eindhoven: Faculty of Electrical Engineering, Eindhoven University of

Technology, 1992. EUT Report 92-E-262.

71

List of abbreviations

ADC

ASK

BW

BER

CCTV

CPFSK

DAC

DFB

EHP

ER

FLL

FSK

FWHM

IBeN

IF

IMIDD

lOP

LO

LPF

OlE

PDF

PRBS

RIN

SAP

SMF

SNR

TUD

TUE

UT

VVM

Analog to Digital Convertor

Amplitude Shift Keying

BandWidth

Bit Error Rate

Close Circuit TeleVision

Continuous Phase Frequency Shift Keying

Digital to Analog Converter

Distributed FeedBack

Electron Hole Pairs

Extinction Ratio

Frequency Locked Loop

Frequency Shift Keying

Full Width Half Maximum

Integrated Broadband Communication Network

Intermediate Frequency

Intensity Modulation/Direct Detection

Innovatief Onderzoek Programma

Local Oscillator

Low Pass Filter

Optical/Electrical

Probability Density Function

Pseudo Random Binary Sequence

Relative Intensity Noise

Stress Applying Part

Single Mode Fiber

Signal to Noise Ratio

Technische Universiteit Delft

Technische Universiteit Eindhoven

Universiteit Twente

Vector Volt Meter

72

Appendix 1 Specifications

The specifications of the phase diversity are:

optical wavelength : 1535-1540 nm

receiver : {3x3}/{4x4} phase diversity

penalty compared with an

ideal heterodyne receiver : {3x3} : < 0.4 dB / {4x4} : < 0.25 dB [I]

modulation : ASK / CPFSK

bitrate : 140 Mb/s

modulation-index CPFSK :>7

demodulation CPFSK : single filter method

IF : 300 MHz

IF-bandwidth : 650 MHz

LP-bandwidth : 140 MHz

laser linewidth : 30 MHz FWHM penalty < 1.4 dB [14]

RIN : < -135 dB/Hz penalty < 7.0 dB [15]

front end noise : 50.10-24 A2/Hz

responsivity front end : 10 V/mW (R".., = 50 Ohm)

polarization mismatch < 10 degrees penalty < 0.15 dB

extinction ratio ext. mod. > 10 dB penalty' < 1.2 dB [9]

phase mismatch {3x3} hybrid < 5 degrees penalty" < 0.5 dB [16]

gain imbalance {3x3} hybrid < 10 % penalty" < 0.8 dB [16]

phase mismatch {4x4} hybrid < 20 degrees penalty' < 7.3 dB [16]

gain imbalance {4x4} hybrid < 10 % penalty" < 1.2 dB [16]

BER : 10-9

sensitivity {3x3} ASK/CPFSK : -43 dBm / -40 dBm (5 dB margin included)

sensitivity {4x4} ASKlCPFSK : -37 dBm / -34 dBm (5 dB margin included)

'can be optimized when mismatch is known.

2f" I" 'I" 'I T" 'I" I" OJ, " "'1 ~ ] :,;1.5'1" ~

~- ~ ~ Iy --------- 1 > " ]

0.5 ~

o o

1 I !""",!",., •• 't'~~

2a 40 60 ea Current (mA)

~ ,

-d3id'$[] I 6 I

Au HETA.L1Z£O ALUHINA

Gl

(L~ ;\1 1

1 , ~

100

J: E

~ , Q. ~ , o ~

" C1

-'

:I~""'" I~I' "d~,/ ""'~>1 , I ~

3 ! / , ' / /

21:" / ,/ i : I i

/ / -; / ' 1 ' I / '

I / ~ , '

a II I i "3 o " .. ! I",! t t, It, I ,j

(") > :r -= ~ -= " tD ft ::I ::I, Q,. '" _. ::t,

~ " '" N 0 -. -:r tl <>

q po § S '" §, </l - ::r ft tD '"' (\l

or Ul" '"

20 10a 40 60 Current (mA)

ea

~ 0 ...., g. (\l

"0 -J

'" ::r _ . ...... -. -= </l

Temp. (C) : 25 65

tl cil

Ith (rnA) : 18.8 40.4

Es (mW/A) : 108 74 ...... po </l Rs (Ohm) :7.4 7.6 (\l '"1 </l V30 (V) : 1.07 1.06

lop (rnA)

74

Optical spectrum of the transmitter laser

..... \l I

Cap) A'+ sua,+uI

[S) m If)

[S) I\-If) .....

" E

[S)[ If)'-' If) ..... ..c:

3: E

-f1~ mlf) [

[S)(l)CI: E [T) ~.

If)(l)en ..... > .

to I'-If)

:s: u

[S) ..... If)

If) (\j

.....

(f)

:::J 0

If) ..... :::J I z

H

f-Z 0 U

E en c -0

(\j ~

en (\j

(T) '" If) I ~

s:. +' m c Q) c

0 Q)

> (I)

to (I)

3: Q)

s... -0 D-

Q) Q..

+' ;:)

s:. (f)

m Q)

Q) -0 3 0

:L s... Q) Q)

3 -0 0

(L (f)

Linewidth of the transmitter laser

LLJ :::> a:: = = ~=

U 0------<

=

:j

~ ~

E 1 = LJ = :::>

LJ 0------<

F CSJ 0 CSJ CSJ '-

-J .,-i = .,-i LJ ru

[J I LLJ CSJ I-- CSJ

---.J I--

'I ,-,-"

75

~

...::i V ~ ~

~ ~

;;:

~ ~

k#

.. N

i = = CSJ CSJ

CSJ lJJ

D-

= I--(f)

N

= = I--cc a:: I--(f)

U Q.J

(J)

E

CSJ CSJ

CSJ lJJ

I--(f)

N

= ...::,::

CSJ CSJ

= :::>

*

N

= = CSJ CSJ

.,-i

= cc *

2["1" ",i,," "I'''''''-ry~iil'''" 'I~

>' 1.5

• 01 • +'

g

0.5

at .... ..... .. .. , o 20 40 60 se 100

Cu,..,..ent (mR)

~ ,

£1 kd'$[] I 6 I

Au HE TA.UlEO ----ALUMiNA

Q)

~

J: E

+' , Q. +' , o

+' .<: 01

-'

5 r .. 0" .. 0 " 0" ." • '0)' , 0 ' .... 'j

4J / ~ .I /~ I / 3

3 / // i ~ ; / 1

t / / ,

2 / ./ -= L / j

J=(~< .. " .. o .1 o 20 40 60 S0 100

Cu"',..ent (mR)

n ::r ., ~ .. ::J. '" ::to (')

'" 0 ...., ~ ::r " ~ e:-o '" D. = ., 0 '"' 0;-'" ~

-.l 0\

Temp. (C) : 25 65

Ith (rnA) : 23.4 49.3

Es (rnW/A) : 151 103

Rs(Ohm) : 5.0 5.5

V30 (V) : 1.01 0.99

lop (rnA)

" rn 0

t-

~ -,J ; ~

L

~ -

201 I'

[ -30 I

H E II -40

~I!.LI~JjMlMlulkaelllll' mIllllIlIUN!lIMIII· -50 -

1490 1510 1530 1550 1570 Wave 1 ength (nm)

CONTINUOUS 25 C 67.9 rnR 5.0 mW

Power weighted Wavelength 1538.2 nm

Side Mode Suppression -33.2 dB

~ g . ., -~ q c: 3 0 ....,

~ -::r '" g e 0

'" £ ;" 0-....

~ .... -J -J

1590

78

Linewidth of the local oscillator laser

! ~ I ! = I

! = l I = ~ -_. -cc.- .-~----- .... ,-- - - . i u

rT.u:~ I I cs: ~

= I ~= Lfl

~·--f.---I---.J---+----+---+----'~ -----~-. u_ ~ ~ , i , ! ,

r--+--1i- ) ~ i I ~---+--+---------. --- -~-~--.- --- --3~----··- .­

~

,

! , I ,

, I I ~ ,

i

I ?~ I

r----t---t----t----t---;.F=~-+----r---j-.--i_--- ---

! ! , I , I , , i , :

N = ~ = = = ::> * ~I I I ##=-

-01= ::> I ~ N N

~1~·~=4f.----+--~~~~~-+---~---.J---+I----+---4---4:: ::

..,-;1 -0 .~ = ~~_.~-~_-:-'~----_._=_.-.. _---r:,~-•...... -.~. -. - ! -]---1= ~ ~

79

Appendix 3 Data of the GEe optical amplitude modulator

FEATURES

• Wide bandwidth

• Low insenion loss

• Low drive voltage

• EMC/EMI-free transmission

APPLICATIONS

• Microwave frequency optical links

• Optical signal processing

• Optical sensor systems

• Coherent transmission

Integrated optical modulator with lid removed

The usc of external modulators with CW laser diodes have considerable advantages when compared with direct modulation; integrated optical modulators eliminate dynamic laser chirping, have gigahertz bandwidths with nailer frequency responses and allow greater signal to noise ratios. This makes them panicularly suitable for usc in communication systems where transmission and manipu1ation of high bandwidth data rates arc required.

TYPICAL OPERATING PARAMETERS AND SPECIFICATIONS

Operating wavelength Bandwidth Drive voltage (Vrr) Electrical impedance Electrical return loss Extinction ratio Optical insenion loss

RF connector Package dimensions

Standard Fibre: Input Output

Environment

Notes

~

(I) (2)

(3)

Y-3'i-5170-01 .L!.ni.ts

1.3 ~m 2.5 GHz 3.0 V 50 Ohms 10 dB 25 dB 6 dB

SMA SMA 6Ox35xl5 42x25xl5 mm

I metre York HB 1250 polarisation maintaining I metre GPT SMOg standard monomode

Normal laboratory conditions

(I) Also available for 1.5J.lm wavelength. Specification as above with exception of the drive voltage which for Y-35-5370-02 is 4.0 V and Y-35-5600-02 is 8.0 V.

(2) Electrical bandwidth ± 3 dB measured using a modulated optical microwave link. (3) Devices can be supplied with other fibres. The devices are polarisation dependant and if

monomode input fibre is chosen, a fibre Polarisation Controller is an accessory available through GEC Advanced Optical Products.

80

Appendix 4 Circuit diagrams

I Laser current control and modulator circuit

2 display cicuit temperature, laser current and -SV source

3 Temperature control circuit

4 Frequency control circuit

5 Optical/electrical converter module

6 IF section

7 Clock and data recovery circuit

8 Rear panel connector and laser connections

9 Front panels

27 > -< Eel s

R22 T R23

SK S1

2 ")

02 DF8

"¥'" ~ ~ lASER 03

lcs

OS

.17 BAV ~ ~ BAV ~ ~ } 99 ~

~ r2n 99 ~

~ ==

R1 ...:: ...; 22n

l1 ,. C3 11 22pF R6 • A 1r 4<7

C1 • 3?.. 01 .470uH 04 ~

,. J;s os ~ Ii ~B :;: 5 01 k BAV

'- BAY -' '- : ,. L.4 L.4 E K '.5 99 ~ ~ r2 99 ~ ~ 6uB 02 - 22n

R5 I~ I~

JlO2~17>-' - +

6 0 j:;IEF ) ,. BCW~

12,5Y 2

" BAY 99 608 Be B46B I C 1 03 04 R19 - - -- FIJ~ I; \i TCA 520B R12,,-20> BFR 5.6

L M LM C

G ,.5 r 93 (I

00 -

385 03

93 ~ 1K R9 ~

R20 Sk BFR

10 ,I S.

C4 • '.5 Q5

Ie2 :~ H ~> P

1Su R4 R7 RB R14 R16 R17 021

R10 R15

,.5 T6U8 33K 1k 22 " 1K 1< 560

51 - 27 "")

1 Laser current control and modulator circuit

82

LASER CURRENT

TEMPERATURE

.11V 101

101

13

R~6

101

IOn U

Cli t24 114~ R~' R~I R~1 ..

lOOn 22n 1. 10K 19K1 1[21 31 101

- 15V 25V 1

... 01 1N400S

-5 VOLT~GE SOURCE -------------~---

-11V L~ 317 0 -SV

31 21

Ie 2 R25 08 lN4005

G

-3, ]V Cl0 ell R29

4 1Su R2G 68n 101 15. 68n

2lV e12 en 1K 25V

• 6ua 68n • m 2N412" 15.

25V m ~)

1

2 display cicuit temperature, laser current and -5V source

83

2 C34

cn l8L05 0

Ie 5 R50 18n T

-L R5l G R51 "!' C30 10K

6K8 6K8 lu

3

C31 R52 lK82

R66

~JOK .BV .15V 2

31

Ai?

10K 1 C~l

1DOnF I .,. -6V -15V

3 Temperature control circuit

84

16

2' ~~.1~5~V ______ -r __________ -r~

RB2 CSO

Ie 12 '"" Ie 13 RBl

100n 410 -470 22u

in 6 out iI'r RF ~AR RF >,:::...L-l-il--. __ -I-~ SP B 7 5 5 r--,- '" /

9111 GRO GRD

MIA 1101 100n

.,. ~AR 6

o v

("2l. ,15V '-"-J

lcss

lDn 16

.. .....!... ~ 01

H

...i. S os 13

..l... 12 C DC

-1... f.!.LiI 0 DO

f.!.L RCO ,......!... ENP

10 EN1

~ 2 W CLK

~ 9 LOAD

C67 r ~ CLR

10n1 Ie 16174HC161

.,.

""v 16

1..2... flLl-A 01 I...i. 13

B DB 12.. 12

C DC 1-1... 11

0 00 74HC04 RCO ~I-

L....2.. 10

INO ENT

2 CLK

9 ~ LO~D

L-...!., CLR

Ie 17114HC161 -----.-.-----

1 o V

31 -15V

ln

..

mT 1D'l

i I

"11-

12

13

14

1S 1S

11

1S 19

hold " reset 22

.1L

.1!... ~ .2.!.. 22. 2!...

....!.

...1!.

....!. ,.!!.

RBJ

1S'

PROG

OB.O OB.l

OB.2 OS.J

DB.4

OS.5 OB.6

OS.l

P2.0

P2.1 P2.2 P2. J

P2 .•

P2.5

P2.S P2.1

-RO

iR PSEN

AlE

Ie 19 - ----

10 Ie 14

7 12 13

C65 1n C66

40

X1 WJ 187481

3 X2

4 RESET

55 h1-,..L TO ~ T1

o-!--INT

Pl.D 17

2B P1.1

P1.2 P1.3

29 3D 31

P1.4 12

P1,5 33

P1.6 P1.1

14

EA f-l-20

4 Frequency control circuit

C70 II33p

.ill 78L 10 ~ IF" E!] ,15V T C:

11 IC 19

I~on II 10n ;!r-

,~

13

~ 12 10AC 081

14 RSS 11 C14

10 10' :~ 9 ~22n B

RBS

7 f!--Q kJD 6 5 ~

10' Ie 20

RB< 2 116~~ 014 -.... ~ ~ C75

1N914

'-~ 10,

51 i :~ Cl2

1"

~

<- • .9V

FCI PC

co<

,

R1no

"

~ 100n

91&0

R1Q04

"

Rl01

"

• POC

85

A1Dl 510

"

~ ~J.. 100n

~ 100n

-,-..... " ",

10 12 ,C 21 co. "SA

22110 15 Ri09

5 IC 20 16,5

CB7

<- • - SV -'- " R1D'5

I 10k. ' 1 _____________________________________________ 1

<- A .911 ..J.... I

Rl1i Rl1l A1'" Al1' 510

" " " " R120

fel PC

eBB

~ ~...L ~ ~ " .,.. 100n

~ "T iDDn 100n 22, 100n

1 • 10 12 c •• 8T&0 poe 2200 15 1178:5 R121

5 IC 22 16,5

~oon C95

~

tt:: GA IN

<~u_=-~5V~~=--'--__I:!:jill-fu1!.J__j1 <- B A117 R119 I I

-r- 10k i

<- A .9V

Fel PC

<- • - 5V

31 -15V

C1D4

10, •

---------------------------------------------j

C10

R12-t

"

~ 100n "!'"

Bl&O

6

"

R1:?)

" ~

100n "'"'!"'

A12S

"

~ 1Don

10 12

A 127

"

poe 2200 OSA

15r----,0785

5 Ie 24

~,oon ... "

GAIN

..,..

R12!! A13l

Ie 25

"

'"

510

R132

C 102

C1aa

,

A1::!!!

: _____________ ~~ ______________________________ J

2

5 Optical/electrical converter module

s., R11D

18,5

R1ll

111,5 SO,

..,..

.. ,

R123

16,5 ""

86 1---------------------------------------------------------------

------------j IN A I lUl2 .. l3/l2f1

v' , , -----------_ ..

Lor P~SS F I L TE~

.- ...................... i IN A. I l4/l21l LS/J211

, , +:

Lor P~SS F I L TE~

100'~ RUO ,

~

10a'Q AH5 ,

or

C119

~H 10

C122 1DOn 10(]n

~GC1out H'0' e116 560

HfTlll-l ...... ~~"-' 7 1Dn ,-" .............. -,

R1H ,.

I Ie 29: 7

1--- IC29 IVA -----

... 0511B

AGC AMPLIFIER

, , ' .. I£. 11!!7

I IVA or 05118

AGC AMPLIFIER

, 4' I C 29' "8r;;'-1--"6r

C120 I ... TC121

100, .. 1 . .'00' V WIXER -:r

C12~ ::: • 10:0n *

tHC",' r",'0=,-",",:;--'

(;:136

100, ,LJ 10,===, 'f1 AGe2ou! ,......--,

7 .- ................ -. R149

1K

1SWA

e137 out B

100n or

1----------------------------------------------------- __________ ~

1- ...................... ,

IN' , L?132.'

,

or : I'" , v , ,-----------_ ..

Lor PASS FILTE~ CH2

100'~ R1S0 ,

..

, 4, , I r 00,7 -r· ~: J.. 05118 ...

AGC AMPLI F I ER

tH1

10lln

10

C 150

,LJ 10, 'fl AGC30ut

7 ________ _

IAW9tD1B

R15" ,.

C154::: C 1110n 1

or

, , , 1 ______ ----------------------------------------------- __________ _

6a IF section

m

10K

AGC offset

A163

1K

C153 1u

Ie J5

6

c- - A f-'---11

21:: 10K

R1 DB

'"

2K 10K

R19B

'"

51

R167

"

'"

87

110K

"

7 Ie 36 IIV2 SAT B5

2 015 016 017 019

AGCln'

C154

1u

GAIN

10K

3,

2

2

AGeln2

C165

I.

AGelnJ

C166 6b

1u IF section

88

La L9 L 10 L 11

SUA 120 17n+33n 56n+3911 5611+39n 1811+33/1 S~A

INPUT A

1157 1159 "='"

A176

61 ·

· LOW PASS fiLTER

S~A 120 ---------------

INPUT B

A17a

· 61

· S~A 120 lK 590

INPUT C

A1ao C160

· 61

· 10n + lK 1C163

590

- -"='" 1 On L13

Ie 31 L 12 Ie 38 1. 1. -----

I SMA SUA SUA

f:~, moo,,"", C162 C164

100. lOOn "='" "='" "='" lOOn

. .

6c IF section

R213 R211

." r-...L:' __ "',' 1

1

U U42

~~~':"r-~-~JC_+"'!1' ~ , P ',',10 ~'.;.".;2_-+-,,'<t' )" , ..... ~71 V

~ 1'" .:' l" i, .0. I A214

I H7 Lf " 5'll

IN Jl E, Li Ai

'i , , , , 'M our 1-

11215

A230 ..

89

~ ,

hel 'cd

D' 10,

~ 'I ;;

" ~ '---r

r-Uff '---'

"" • .. cunl

f-A2l3

'" A212

I ~C179 " , . , 10.

I

CLOCK t'M -If"" " '/1--"",+...1.1

_ '-I 14~ c ~_

.1!..

~ 10n ~nH

'00

' .. ATA~IIII. lIel)8

"' 11 10n R229

10'

'."

~f-!:++-' IIIC10116

12 ~ 10 t-- ~ 15

11 8"~ 4 1) ,WL,

-;:-' ".

IICtlU"

U46 1

IIC1648

R2B

'"

,

,

" I~ • .!........ U

~ ~O ..!!lll... r-' 'r;H •

l~~ u~s 1

IIC12(HO

'F

'--

'" ~ '" 11

~ YAII

". " I-

m

i ~ R2H

".

"" ~C1Rl ~ I

r! , I'O' R235 ,,. , ..

m

(182

'" '~H-!Eiil

-rifu:J.. Ll. .~' , 1,

- U18

11215

11.

'"

1 I F156

."

,

7 Clock and data recovery circuit

~I Rl22

" ~ R221 R226

'" S60

e17]

Hf t+ 1000p 1000p

RZ25 R22?

'" '"

H H C1H e176

lDDOp 1000p

GND 1

+1SV 2

NTC/GND 3 BROWN -4

NTC/- 5 GREEN

6

NTC31 + 7 WHITE -8

9

10

GND 11

Vamp 12

13

14

15

+5V 16

17

1B

19

+ IILD 20 WH ITE -21

22

23

-I/LD 24 BLUE

25

GND 26 BLACK -- 5V I LD 27 BROWN -PELTIER 2B WHITE

29

PELT I ER 3D PINK -- 1SV 31

GND 32

-I

-

r - ~r

-I

I

-'-~

r

I

~

90

"" 5 90

eOr-3

70 2

60 1

V l.

r-- '-___ +-__ NTC

W

8 Rear panel connector and laser connections

. · · · · SET UP VA.LLE

TEMPERATURE

D · U3

· U2 0

TEMPERATURE CONTROL

D · ._, . """ CO , · U, . . · .- ""

, CURRENT I rrVII

- . ~ I II I

. . · · · ·

· · · · . • . • . • • • • • . . . • AS< EA.-iS"lI: CLOCK8dlllTA

SET UP VALUE FREQUENCY J )( 3 COJPLER I FI PDF RECOVERY

TEMPERATURE

c=J aJNTROL

0 D 0 , ''''"'

, · U, T ....... 0 • · · U2 0

, , 1 •• - 0 1 • · 1

D 0 0 ~:

TE~ERATURE '"" • • 0 /'

CONTROL •

D 2 5 • 2 0

- · .T_ Do\TA OUTPIJ

" .. D 0 , . ON ,

· U, · 0 0 ""'M . . · 0 , ,

· ''''''" "" , - , • • 3 0

CURRENT J mA INPUT 0"""'" Wi< 0 0 " - 0

""" - . ~ I II II II II II II II I

. · · · · . . . . . . . . . . . .

9 Front panels

92

Appendix 5 Source code listings of the frequency locked loop

Listing for the zero IF case:

ORG 0

MOV R2,lOH ; stapgrootte

MOV R3,80H ;instelling voor DAC

MOV R4,80H ; tellerstand

LO: MOV A,R4 ;vorige tellerstand (RS)

MOV RS,A ;wordt tellerstand

MOV A,R3 ;R3 wordt R3 + stap

ADD A,R2

MOV R3,A

OUTL Pl,A ;DAC

MOV A,lH ; reset teller

OUTL P2,A

MOV A,3H ;enable teller

OUTL P2,A

MOV A,2H ;hold teller

OUTL P2,A

IN A,PO ; lees teller

MOV R4,A

CPL A

INC A

ADD A,RS

JC LO ;jrnp indien juiste richting

MOV A,R2 ;stap wordt -(stap)

CPL A

INC A

MOV R2,A

JMP LO

END 0

93

Listing in case the IF has to be 300 MHz

ORG 0

MOV R2,lH ; stapgrootte

MOV R4,lBH ;gewenste tellerstand

MOV R3,80H ;initialisatie DAC

MOV A,R3

OUTL Pl,A

MOV A,R2 ;berekening -(stap)

CPL A

INC A

MOV R5,A ;-(stap)

LO: ANL P2,OIB ;reset teller

ORL P2,llB ;enable teller

NOP

NOP

ANL P2,lOB ;hold teller

IN A,PO ; lees teller

CPL A

INC A

ADD A,R4

JC Ll ;verlaag stroom indien

MOV A,R3 ;teller te hoog, anders

ADD A,R2 ;verhoog stroom

MOV R3,A

OUTL Pl,A

JMP LO

Ll: MOV A,R3

ADD A,R5

MOV R3,A

OUTL Pl,A

JMP LO

END 0

Eindhoven University ,:,f Technol<xlY Re,3ean;h Reoorts 155N 0167-9708 Ccx:ien: TEUEDE

Fl1cu1ty of Electrical F.nqineering

12441 . Maril" and Ton van den Boom. Ad Haruen OF CLASSICAL AND MODERN CONTROLLER -DESIGN: A case study.

EUT Report 90-E-244. 1990. ISBN 90-6144-244-3

12451 Berg. P.H.G. vande ON THE ACCURACY OF RADIOWAVE PROPAGATION MEASUREMENTS: Olympus propagation experiment. EUT Report 90-E-245. 1990. ISBN 90-6144-245-1

12461 Maaqt. P.J.!. de A SYNTHESIS METHOD FOR COMBINED OPTIMIZATION OF MULTIPLE ANTENNA PARAMETERS AND ANTENNA PATTERN STRUCTURE. EUT Report 90-E-246 1990. ISBN 90-6144-246-1

12471 Jozoiak. L. and 1. Spmova-Kwddltaal DECOMPOSITlONAL STATE ASSIGNMENT WITH REIISE OF STANDARD DESIGNS: Using counters as sub­.achines and uSln9 the method of maXimal ad]acensies to select the state chains and the state codes. ElIT Report 90-E-247. 1990. ISBN 90-6144-247-8

11481 Hoellmakers. M"'))",a:~n:d,,;,).;.~MA', ,~~~¥", DERIVATION AND UI, THE SYNCHRONOUS MACHINE WITH RECTIFIER WITH TWO DAMPER WINDINGS ON THE DIRECT AXIS. EUT Report 90-E-246. 1990. ISBN 90-0144-146-6

12491 Zhu. Y.C. and A C.P.M. Bach. P. Eykhoff MULTIYARIABLE PROCESS IDENTIFICATION FOR ROBUST CONTROL. EUT Report 91-E-249. 1991. ISBN 90-6144-249-4

mOl PfaftenhOfer. P.M. and P J.M. Cluitmans. H.M. KUlpers £MDABS: Design and formal specification of a datamodel for a clinical research database system. EUT Report 91-E-250. 1991. ISBN 90-6144-250-8

12511 EiJndhoven. J.T.J. van and G.G. de Jonq. L Stok THE ASCIS DATA FLOW GRAPH: Semantics and texiual format. EIIT Report 91-E-251. 1991. ISBN 90-6144-251-6

12521 Chen. J. and P.J.!. de Maagt. M.HA.J. Herben

1253)

(2541

12551

WIDE-ANGLE RADIATION PATTERN CALCULATION OF PARABOLOIDAL REFLECTOR ANTENNAS: A compar,tlvP study. EUT Report 91-E-252. 1m. ISBN 90-6144-252-4

Haan. S.W.H. de ! PWM CIIRRENT-SOURCE INVERTER FOR INTERCONNECTION BETWEEN A PHOTOYOLT!IC ARRIY IND THE UTILITY LINE. RUT Report 91-E-253. 1991. ISBN 90-6144-253-2

Velde. M van de and PJ.M. Cluitnans EEG ANALYSIS FOR MONITORING OF ANESTHETIC OEPTH. ElIT Report 9I-E-254 1991. ISBN 90-6144-254-0

Smolders. A. B. AN EFFICIENT METHOD FOR ANALYZING MICROSTRIP ANTENNAS WITH A DIELECTRIC COYER USING A SPECTRAL DOMAIN MOMENT METHOD. EUT Report 9H-255 1991. ISBN 90-6144-25H

Eindhoven University of Technology Research ReDOrts ISSN 0167-9708 Ccxien: TEUEDE

Faculty of Eiectrical Engineerioo

(2561

(2571

1256)

(2591

12601

12611

(2621

(2631

(2641

Bac'x. A.C.P.M. and A.A.H. Da.en IDENTIfICATION FOR THE CONTROL OF MIMO INDUSTRIAL PROCESSES EUT ~eport 9H-256. 1991. ISBN 9H>144-256-7

MOdgt. P.J.!. de and H.G. ter Morsche, J.L.N. van den Broek A SPATIAL RECONSTRUCTION TECHNIQUE APPLICABLE TO MICROWAVE RADIOMETRY. RUT Report 92-E-257. 1992 ISBN 90-6144-257-5

Vleeshouwers. J.N. DERIVATION OF A MODEL OF THE EXCITER OF A BRUSHLESS SYNCHRONOUS MACHINE. EUT Report 92-E-258. 1992. ISBN 90-6144-258-3

Orlov. V.B. DEFECT MOTION AS THE ORIGIN OF THE llF CONDUCTANCE NOISE IN SOLIDS. EUT Report 92-E-259. 1992. ISBN 90-6144-259-1

ROOllackers, J.E. ALGORITHMS FOR SPEECH CODING SYSTEMS BASED ON LINEAR PREDICTION. EUT Report 92-E-260. 1992. ISBN 90-6144-260-5

Boom, T.J.J, van den and A.I.H Damen, Martin Klompstra IDENTIfICATION FOR ROBUST CONTROL USING AN H-Infinity NORM. EUT Report 92+261. 1992. ISBN 90-6144-261-3

Groten. M. and W. van Etten LASER LINEWIDTH MEASUREMENT IN THE PRESENCE OF RIN AND USING THE RECIRCULATING SELF HETERODYNE METHOD. EUT Report 92-E-262. 1992. ISBN 90-6144-262-1

Smolders, U. RIGOROUS ANALYSIS OF THICK MICROSTRIP ANTENNAS AND WIRE ANTENNAS EMBEDDED IN A SUBSTRATE. EUT Report 92-E-263. 1992. ISBN 90-6144-263-X

auditory evoked potential pal terns. EUT Reoort 92-£-264. 1992. I5BN 90-6144-264-8

(2651 Wellen, J.S. and F. Karouta. M.F.C. Schemmann, E, Sma lbrucrqe , L.M.F. Kaufmann MANUFACTURING AND CHARACTERIZ!TJON OF G!ASIALGAAS MULTIPLE QUAHTUMWELL RIDGE WAVEGUIDE LASERS. EUT Report 92-£-26), 1992. ISBN 90-6144-26)-6

(266) CluHmans. U.N. USING GENETIC ALGORITHMS fOR SCHEDULING DATA FLOW GRAPHS. EUT Report 92-E-266 1992. ISBN 90-6144-266-4

(2671 ,J6zmk. L. and U.H. van Oil. A METHOD FOR GENERAL SIMULTANEOUS FULL DECOMPOSITION OF SEQUENTIAL MACHINES: Algoritlims and lmplenent,tl"''!. EIIT Report 92-E-267. 1992. mN 90-6144-267-2

!268j Booo. H van den and W. van Etten, W.H.C de Krom. P. van Benllekom. F. HIlI]S,ens L. Niess,n, F. de Lener ~ =

AN OPTICAL ASK !NO FSK PHASE DIVERSITY TRANSMISSION SYSTEM. EUT Report 92-E-266. 1992. ISBN 90-6144-268-0