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    Application Note AN-EVP001Rev 4 - Aug 2006 How to Design Low Cost, High Efficiency, SMPSs

    Copyright 2006 - ASIC Advantage, Inc. All rights reserved - Revision 4 August 22, 2006 1 / 17

    INTRODUCTIONThis application note, AN_EVP001, was created to

    complement the IN-PLUG product datasheets andhelp SMPS designers select the right AAI Controller ICfor building low power, off-line flyback SMPS.

    The goal is to build the most forgiving and cost-effective state of the art SMPS using AAIs patentedapproach that eliminates the necessity for severalstandard SMPS components.

    The input mains line voltage can be either 110 voltsAC (Domestic US) or 85 to 265 volts AC

    (International.) The line frequency can be 50, 60 or400 Hertz nominal. Most designs will also operatefrom high-voltage DC with any polarity, like 140 voltsor 120 to 400 volts.

    The emphasis is on low-power, low cost SMPS withone isolated low-voltage output. However, please notethat AAI ICs can also be used in higher power SMPSapplications, especially when high output voltages andmultiple outputs are required by applications as forexample in CRT monitors.

    When designing an SMPS, please check whichgovernmental regulations your product will have to

    comply with, such as the following: Power Factor Correction (PFC),

    Electro-Magnetic Interference (EMI or EMC),

    Safety issues

    This application note deals with electro-magneticinterference and safety issues, but not with power

    factor correction which is not yet required for SMPS

    below 75 watts of input power. Other IN-PLUG prod-ucts like the IPS101 are offered for higher powerSMPS with power factor correction.

    Excluding the PFC, an actual SMPS will include:

    The mains AC line input with the Diode Bridge,Storage Capacitor(s), EMI-reduction Choke as

    well as safety and protection components likeFuse or Fusible Resistor, MOV and X-Capacitor;

    The high-voltage Power MOSFET with itsassociated protection and EMI-reductioncomponents;

    The line-side controller with its associatedcomponents;

    The secondary Power Diode, Tank Capacitors andRipple-reduction components;

    The load-side controller with its associatedcomponents.

    The components between line-side and load-side,e.g., the Switching Flyback Transformer with its

    isolated secondary, the feedback Optocoupler andthe Y-Capacitor.

    Corporation

    AApppplliiccaattiioonn NNoottee AANN--EEVVPP000011 RReevv..44

    HHooww ttoo BBuuiilldd CCoosstt--EEffffeeccttiivvee LLooww--PPoowweerr

    FFllyybbaacckk SSMMPPSS wwiitthh AAAAIIIINN--PPLLUUGGCCoonnttrroolllleerrss aanndd IIXXYYSS PPoollaarrHHVV PPoowweerr

    MMOOSSFFEETT aanndd SScchhoottttkkyy DDiiooddeess

    Please, see Figures 9 and 10 page 12, for detailed

    application schematics.

    L

    O

    A

    D

    IPS20,

    IPS21,

    IPS25

    FEEDBACK

    controllers

    IPS15,

    IPS15H,

    IPS18

    FLYBACK

    controllers

    Line

    Output

    Diode

    DC OUTAC IN PRIMARY SECONDARY

    Patented

    Modifiedsnubber

    Optocoupler

    Fig.1

    Rectifier

    bridge

    MOSFET

    IPS101

    PFC

    Controller

    Transformer

    Note: If the reader is unfamiliar with the SMPS concept or withsome of the terminology, he may refer to APPENDIX 1 page 15.

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    Application Note AN-EVP001Rev 4 - Aug 2006 How to Design Low Cost, High Efficiency, SMPSs

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    Line Sid e Flyback Cont r ol l ers:The IN-PLUG IPS1x series (IPS15, IPS15H, IPS18)

    was designed for low-cost , high efficiency, low-powerflyback off-line switching power supply solutions up toapproximately 50 to 70W. It contains a shunt-regulator, a precision oscillator, a PWM with itsassociated comparator and loop compensationcomponents as well as all the necessary biasing andprotection circuitry (thermal shutdown, undervoltageand overcurrent).It is optimized to operate with an optocoupler toprovide the feedback from the secondary but couldalso be used with a bias winding which couldsometimes be more economical. Typical applicationsinclude domestic and international power supplyfeaturing and AC input ranging from 90 to 264V andDC from 100 to 350V.IPS15H and IPS18 incorporate a special circuitry thatinvolves a counter and some other digital blocks. Thisfeature has been added to avoid delivering a highcurrent to the load on an overload condition, whichmay result in damages to the SMPS, the load or both.IPS18 has been optimized for stand-by applicationswhere the SMPS has to deliver a minimum amount ofpower while being required to comply with tight "green"regulations. When no load or a minimum load isconnected to the SMPS, the controller enters a cycleskipping mode to keep the power consumption as lowas possible.

    Note: In low-power applications, the IPS1x series caneliminate MOV, Common Mode Choke, X and YCapacitors and still enable SMPS to meet the strictestgovernment regulations for EMI and input surgevoltages.

    Load Side Feeback Cont r ol l ers:For applications requiring current limiting & voltageregulation, or voltage limiting & current regulation,select AAIs feedback controllers: IPS25, IPS20 orIPS21. Otherwise, select an appropriate zener diode

    or a TLC431 adjustable shunt regulator as shown inFigures 2 and 3 below.

    If the output current is below 700mA, select IPS25.

    If the output current is 700mA and above, select IPS20 or IPS21.

    Note: Both the IPS20 and 21 incorporate the same

    trimmed temperature compensated 1.2Vreference to set the output voltage. They only

    differ by the characteristics of the current sensingreferences. The IPS20 features a trimmed 50mVreference with a positive temperature coefficientwhich closely matches that of a PCB copper trace.This copper trace sensing method can be usedinexpensively with very low associated powerlosses. The IPS21 incorporates a constant 100mVreference for more conventional resistors

    The IPS25 can be used in high current applicationwhere efficiency is non-critical. (A 1.3 volts drop isrequired across the current-sensing resistor).

    The IPS20 and 21 can be used at any outputcurrent for maximum efficiency. However, anauxiliary supply is required to power the chipunder a dead output short-circuit condition.

    MOSFET Br eakdow n Volt age:State of the art low-cost, high-voltage MOSFETssuitable for off-line flyback SMPS are available inbreakdown voltages of 400, 600 & 800 volts. For Domestic applications, 400 volts is sufficient,

    though 600 volts are sometimes more readilyavailable and equally suitable.

    For International application 600 volts can be usedwhen the transformer turn-ratio is less than seven.800 volt MOSFETs should be used with higherturns ratio transformers which generate largervoltage spikes due to higher leakage inductanceand increased reflected voltages. With a carefuldesign, an 800 volts MOSFET could somewhatincrease the SMPS efficiency.

    (See also transformer and snubber sections)

    D1

    SB160SCHOTTKY

    U2

    15H

    VOUT+

    GNDSECONDARY

    L2

    4.2uH

    R5

    100

    C4

    2200pF 250V

    TX1

    C7470uF16V

    D4

    9.2V

    C3220uF16V

    VOUT-

    1.5mH

    Fig. 2

    Fig. 3

    SELECTION OF FLYBACK

    AND FEEDBACK CONTROLLERS

    SELECTION OF THE MOSFET

    15H

    TL431

    1

    2

    3

    REF

    ANODE

    CATH

    C7470F

    16V

    TX1

    R7

    10k

    VOUT-

    L2

    4.2H

    U2

    R6

    30k

    1.5mH

    C3

    330F

    16V

    VOUT+

    D1

    5A 60V

    SCHOTTKY

    R5

    10kC4

    2200pF250VGNDSECONDARY

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    Application Note AN-EVP001Rev 4 - Aug 2006 How to Design Low Cost, High Efficiency, SMPSs

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    MOSFET Cur r ent :For International applications, a good approximationon how to rate a MOSFET is as follows:

    Ip (MOSFET) = Input Power / 20.

    Ip = peak current in the MOSFET

    The input power can be deducted from the outputpower, assuming an efficiency of about 75%accounting for losses.

    Input Power = 1.25 x Output Power.

    Please add some safety margin when choosing theMOSFET. Modern high-voltage MOSFETs have verygood switching characteristics, but tend to de-saturateand behave like a current source when the applicationslightly exceeds their nominal current.

    Example:80W of output power requires 100 x 1.25 = 125W ofinput power and a 125/20 = 5A MOSFET. Use a 5A orpreferably a 6A MOSFET.

    MOSFET Gat e Char ge:The total gate charge is an important parameterwhich governs how much operating current will beneeded to drive the gate. The IPS1x series isdesigned to drive MOSFETs with up to 50nC of totalgate charge, but modern MOSFETs are available withtypically 5nC for a 1A 600V device and 8-10nC for a2A 600V device. The lower the gate charge, the fasterthe device and the lower the switching losses. There-fore, choosing a state of the art MOSFET with a lowtotal gate charge has many advantages, usually atlittle or no extra cost.

    MOSFET Ron and Capaci t ance:Choosing a MOSFET with a low Ron resistance willreduce the conduction losses, but this is of littleconcern with a low-power flyback SMPS. Greaterlosses occur when the MOSFETs capacitancecombines with the transformers primary distributedcapacitance. Therefore, it is better to compromise onRon to chose a MOSFET with a low total gate chargeand smaller Drain & Miller capacitance.

    Note: Recent evolution in high-voltage MOSFETtechnology has achieved a substantial reduction of theRon without increasing the drain capacitance.

    MOSFET Ava la nche Rat i ng:The avalanche rating of a high-voltage MOSFETdefines how much energy the MOSFET can safelyabsorb when its maximum voltage is exceeded. TheSMPS should be designed to never cause avalancheunder normal operating conditions.

    This is achieved by:1. Choosing the right breakdown voltage for the

    MOSFET,

    2. Designing the transformer for minimum leakageinductance,

    3. Using a suitable Snubber Network4. Clamping the spikes for higher power applications.

    When all these requirements are fulfilled, theavalanche rating becomes an additional feature thatincreases the reliability of the SMPS and preventsaccidental destruction in abnormal conditions like LineSurges, Lightening, etc.

    Refer to IXYS Polar N - Channel MOSFETs family

    to select the suitable MOSFET.

    Type of Diodes:

    Schottky Diodes:

    Schottky Diodes are preferred due to their unique

    combination of low Forward Drop and extremespeed. Their utilization will increase the efficiencyof the SMPS by reducing the losses in the diodeitself and reducing the switching losses in theMOSFET. They can be used in Flyback SMPSoperating in either Continuous or Discontinuousmodes. Unfortunately, their breakdown voltagescannot economically exceed 60 volts, which pro-hibits their utilization in certain applicationsinvolving large input line voltages, transformerswith small turn ratio, large output voltages.

    Fast Silicon Diodes:

    Fast Silicon Diodes, preferably with low series

    resistance (FRED Diodes), are very economicaland can be found in a broad range of current andvoltage ratings. They can be used in FlybackSMPS that do not operate in continuous mode.

    Ultra-Fast Silicon Diodes:

    Ultra-Fast Silicon Diodes are required for FlybackSMPS that are allowed to operate in continuousmode.

    Out put Diode Curr ent r at ing:To maximize the efficiency, the diode will have tohandle a peak current equal to the peak current in theMOSFET, multiplied by the turn ratio of the

    transformer:

    Ip(d) = Ip(MOSFET) x Np / Ns

    Where Ip(d) is the peak current in the output diode,Ip(MOSFET) is the Peak Current in the MOSFET, Npis the number of turns of the primary transformerwinding, Ns is the number of turns of the secondarytransformer winding.Using a diode for which the voltage drop is notguaranteed at the intended operating peak current will

    SELECTION OF THE OUTPUT DIODE

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    Application Note AN-EVP001Rev 4 - Aug 2006 How to Design Low Cost, High Efficiency, SMPSs

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    result in excessive power dissipation. Due to thetriangular shape of the current and the duty cycle, thiswill result in choosing a diode for which the nominalcurrent could be as high as 2 to 3 times the outputcurrent of the SMPS.

    Looking at the forward drop on the bench or on acurve tracer is also a good way to check the suitabilityof a given diode. The forward drop measured at thepeak current should not exceed:

    - 0.7 volts for a Schottky Diode,- 1.2 volts for a Fast or Ultra-Fast Diode.

    Refer to IXYS shottky diode DSS family to select thesuitable diode.

    Out put Diode Vol t age Rat ing:The diode will have to block a voltage which is thesum of: The maximum output voltage (Vout),

    The maximum rectified line voltage reflected to the

    secondary of the switching transformer when theMOSFET turns ON. (VAC x 1.414 x Ns / Np)

    Some spikes and ringing not completelyeliminated by the snubber network.

    Import ance of a Good Tr ansf ormer :The transformer is a key component of a good SMPSsince it affects all aspects of the project, as well as theselection of other key components and the possibleelimination of others. In a flyback SMPS, theTransformer operates more like an inductor with two(or more) windings than a typical transformer:

    the Primary winding is used to magnetize the coreand store the energy on a cycle to cycle basis;

    the secondary winding demagnetizes the core andtransfers the stored energy to the load side.

    The energy is transferred sequentially from the lineside to the load side while ensuring proper isolationfrom dangerous line voltages. However, despite thefact that it operates like an inductor, the bi-directionalmagnetic coupling between primary and secondary

    reflects voltages and spikes that must be accountedfor in the breakdown voltages of associated switchingcomponents like MOSFET and diodes.

    Major Parameters Af fec ted by the

    Trans former:1. The size of the SMPS,

    2. The frequency of operation with effects on Size,Ripple and EMI,

    3. The efficiency through the losses in thetransformer itself and those induced in the

    MOSFET and the Snubber Network by theLeakage Inductance and the primarycapacitances,

    4. The voltage and current ratings of the MOSFET,

    5. The type, voltage and current ratings of the outputDiode,

    6. The Y-Capacitor and the loop stability through theprimary to secondary stray capacitance andcommon mode noise injection,

    7. EMI compliance,

    8. Compliance with Safety Agencies requirements.

    Impor t ant Tr ansfor mer Parameter s:

    1. Frequency of operation,2. Maximum permissible power at Maximum

    temperature,

    3. Maximum permissible duty cycle at Minimum inputvoltage and Maximum temperature,

    4. Primary Inductance,

    5. Primary Resistance,

    6. Secondary Inductance (or Np/Ns turn-ratio),

    7. Secondary Resistance,

    8. Leakage Inductance,

    9. Primary distributed capacitance,

    10. Primary to secondary capacitance,

    11. Isolation voltage and construction techniquesuitable for Safety Agencies approval.

    SELECTION OF THE TRANSFORMER

    Typical IN-PLUG

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    Application Note AN-EVP001Rev 4 - Aug 2006 How to Design Low Cost, High Efficiency, SMPSs

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    Designing a good transformer for low-power, low-costapplications can be counter-intuitive. The followinganalyzes in the order of their importance the criticalSMPS parameters that are directly impacted by thedesign of the transformer itself.

    Saf et y Agencies Requir ement s:The approval of SMPS by safety agencies creates tre-mendous constraints on the selection of thetransformers core, the transformers construction andsome electrical parameters.

    With larger size transformers, margin tapeconstruction is considered the most cost-effectiveways of complying with agencies requirements. Thistechnique is usable down to a 19mm E-core. But forsmaller E-cores, the margin tape constructionsubstantially degrades the power capability and the

    leakage inductance.

    For smaller and more compact SMPS, the onlysolution is to use Triple Insulated wires for thesecondary and make sure that the CreepingDistance requirements are respected either at theBobbin or at the PCB level. Using Triple Insulatedwires, high performance transformers can be built for:

    6 watts international applications with a 13mm

    E-core.

    15 watts international application with a 16mm

    E-core.

    Leakage Induct ance:Leakage inductance should be as low as practicaleven if it means compromising with other parameterslike copper losses.

    The leakage inductance of optimized transformerstends to be higher with E-cores than with PQ coreswhich unfortunately do not exist in very small sizes.Therefore AAI has designed transformers using E-cores that take full advantage of the IPS15 ability toeliminate a number of external components from thePCB.

    One technique to reduce the leakage inductance in E-core transformers is to use a turn-ratio Np / Ns thatdoes not exceed 7. This is easy to do for domesticapplications and practically always allows theutilization of a Schottky diode for the load side.It is also possible to use such low turn-ratios in manyinternational applications. This will in-turn allow usingof a 600V MOSFET; and in many cases, when theoutput voltage is 12 volts or less, a Schottky diode aswell.

    It is more difficult to design a low-leakage inductancetransformer for applications requiring larger turns-ratiowithout sacrificing several other parameters. In fact, aturn-ratio in excess of 14 is not practical and should beavoided.

    Note: AAI is developing a transformer intended todeliver 35 watts using a turn-ratio of 14 on a 20mmPQ core. This transformer shows excellent efficiencyin flyback mode. A custom bobbin is presently beingtooled in order to pass UL requirements.

    Pr imar y Distr ibut ed Capaci t ance:Reducing the primary distributed capacitance isimportant for the efficiency of the SMPS, as discussedin section MOSFET Ron and Capacitance p 3. Thebest results are obtained when the primary is woundwith either 1 or 3 full layers of magnet wire. The onelayer solution may concern people not used toswitching transformers, since they involve currentdensities in the order of 10A/mm2. However, suchvalues are perfectly acceptable for a single layerwinding which evacuates the generated heat moreeasily than multi-layer ones.

    Pr ima r y t o Secondar y Capacit ance:In applications where the Touch Current must be

    reduced to under 100A or in order to simply eliminateY-capacitor, the primary to the secondary capacitancebecomes critical.

    For such applications, optimized transformers can bebuilt that have acceptable leakage inductance and lessthan 16pF of primary to secondary capacitance.Unless the primary to secondary capacitance is 16pFor less, the Y-capacitor cannot be eliminated withoutcausing loop stability problems that becomes difficult ifnot impossible to manage even with AAIs feedbackcontrollers.

    Note: Adding an internal shield is not an optionbecause it will degrade the leakage inductance of asmall-size transformer by a large factor.

    Maximum Permi ssibl e Dut y Cycle when

    Hot :Most switching transformers are made of ferritematerials which tend to rapidly lose their magneticcharacteristics at elevated temperature. Since flybacktransformers are designed with an air gap, thepermeability of the ferrite material is of lessimportance than the magnetic induction at saturationand, to some extent, the losses at the operatingfrequency.

    HOW TO DESIGN A GOOD SWITCHING TRANSFORMER FOR LOW-POWER APPLICATIONS

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    Since the flyback operation involves operating in abroad range of duty cycles, the core should not beallowed to saturate in the worst-case conditions asfollows:1. Minimum input voltage (100VDC or 180VDC);

    2. 60% duty cycle meaning for example 6s whenoperating at 100 KHz;

    3. Core temperature at full load and maximumambient temperature.

    Frequency of Operat ion:

    Higher frequency does not always mean better SMPS,and therefore in our opinion the choice of theoperating frequency must be based on a number ofconsiderations discussed in section AAI patentedsnubber network p7.

    Maximum Permi ssibl e Power :As indicated in section safety agency requirementsp5, AAI has designed modern and optimizedtransformers for small, low-cost SMPS in the followingpower ranges:

    6 watts international applications with a 13mmE-core;

    15 watts international application with a 16mmE-core;

    35 watts international application with a 20mmPQ core.

    Larger power ratings are possible, especially whenmultiple-outputs and relatively high output voltages arerequired. One typical example is a 75 watts SMPS forCRT monitors which involves 3 low-voltage outputs

    and one 75 volts output.

    Turn-Rat io :The turn-ratio is important for:

    1. Reducing the leakage inductance as discussed insection leakage inductance p5,

    2. Taking advantage of Schottky diodes ininternational applications delivering 12 volts ormore,

    3. Increasing the maximum power with a given coreand MOSFET.

    Two possible optimizations: one with turn-ratiosnot exceeding 7; another with turn-ratios between 10and 14. Values greater than 14 are typically notpractical.

    Pri mary Induct anceEven though the primary inductance is key to theSMPS operation, it is not a high priority. Due to thegap in the core, the required inductance can beachieved in different ways, whereas there is lesslatitude for other requirements outlined above.

    The optimum primary inductance can be calculatedfrom the following formula:

    Lp = 800 / (P x F)Where:

    Lp is the primary inductance in Henrys,P is the Input Power in Watts, not the outputpower. (In doubt use 75% eff iciency),F is the frequency in Hertz.

    Note:1. Use the formula as is for international

    applications using a turn-ratio of 7.

    2. Use Lp = 1000 / (P x F) for internationalapplications using a turn-ratio of 14.

    3. Increase the inductance value by 30% for 220VAConly applications

    Pr ima r y Resist ance:As indicated in section important transformerparameters p4, contrary to general belief, the primaryresistance is not a concern in most applications. Infact, conduction losses which are small compared tothe switching losses are due to:

    1. the primary and secondary resistances,2. the MOSFET Ron,3. the MOSFET current-sensing source resistance.

    Seconda r y Resist ance:As indicated in section Primary Resistance above,the secondary resistance is not a critical parameter.However, when using magnet or triple-insulated wiresfor the secondary winding, it is important to adjust thediameter of the wire to entirely fill one full layer.Several wires in parallel in one layer are perfectlyacceptable since they increase the copper section andsomewhat decrease the leakage inductance at thesame time.

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    Origin:

    Originally, the purpose of the so-called SnubberNetwork was to reduce the switching losses of thebipolar transistors, which at the time were slow andwould otherwise have died from excessive losses.

    As bipolar transistors gave way to faster switches likeMOSFETs, reducing the switching losses became lessof an issue. However, designers found that a snubbernetwork was then required to slow-down fast voltagetransients that otherwise created unacceptable EMI.

    Though it is not its primary function, up to certainpower levels, the snubber network can reduce theover-voltage drain spikes created by the leakageinductance of the transformer. Power levels in excessof 15-20 watts, usually require the utilization of a

    clamp, especially when a 600 volts MOSFET is usedin international applications.

    Ty pical Snubber Netw ork:Figure 4 and Figure 5 show two typical snubbernetworks connected between the MOSFET drain andthe primary ground. It consists of a fast diode D1 anda capacitor C1 which voltage rating should equal thatof the MOSFET and two resistors R1 and R2.

    In the arrangement of Figure 4, R1 is in series with D1and can be omitted in many applications. When theMOSFET is ON, the time constant R2 x C1 should be

    low enough for C1 to be almost discharged at the endof the ON time. When the MOSFET turns OFF, C1charges through D1 and R1 and slows down thetransition, reducing the switching losses and improvingthe EMI at the expense of some power dissipation inR1 and R2.

    This is the snubber effect. R1, which value is usuallylow, increases the dynamic resistance of the diode D1in order to prevent possible ringing created by too fasta transition in the diode. This further improves theEMI.

    The arrangement of Figure 5 can be used when R1 in

    Figure 4 can be reduced to zero without harmfuleffects on EMI. Then, placing a high-value resistor R1in parallel with D1 creates a damping effect on thetransformer at the end of the demagnetization cycle.This can greatly attenuate the free oscillations thatwould otherwise normally occur and adversely affectthe EMI signature.

    AAI Pat ent ed Modif ied Snubber Netw or k:

    Since most SMPS require a snubber network, AAI hasmodified the classical schematics and patented thecircuit in Figure 6 (Patent No. US 6,233,165 B1 hasbeen granted by the US Patent Office on May 15 of

    2001.) The circuit consists of two diodes D1 & D2which are low-cost low-voltage signal diodes, one low-value capacitor and one resistor R1.As before, when the MOSFET turns OFF, C1 chargesthrough R1 to slow-down the transition; but C2, whichis part of the Line-side controller IC circuitry, is alsocharged through D2. When the MOSFET turns ON,D1 becomes forward biased and C1 discharges to beready for a new cycle.As switching occurs, there is enough powertransferred to C2 to energize the entire line-side

    SNUBBER NETWORK

    TRANSFORMER

    MOSFET

    +DC

    D1

    C1

    R1

    R2 R3

    Fig.4

    MOSFET

    +DC

    D1

    R2

    TRANSFORMER

    C1 R3

    R1

    Fig.5

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    Application Note AN-EVP001Rev 4 - Aug 2006 How to Design Low Cost, High Efficiency, SMPSs

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    circuitry without having to use an auxiliary powersupply.

    This arrangement totally eliminates the need for a thirdwinding on the switching transformer (sometimescalled Bias Winding) and results in cost and spacesavings. It also makes the transformer more forgivingto design and manufacture.

    Impor t ant Note:AAIs policy is to give a royalty-free, non-exclusive

    license to use the above-referenced patent to every

    user of IPS products who purchase their ICs either

    directly from AAI or through authorized distributors /

    representatives.

    By slowing down the transitions and creating somecontrolled losses, the snubber network tends toreduce, at the MOSFET drain, the voltage spikecreated by the transformer leakage inductance. Forhigh power applications, however, and especiallywhen using a 600 volt MOSFET in internationalapplications, the utilization of a snubber network maynot be sufficient to clamp the drain voltage to anabsolutely safe limit, and an actual Clamp Circuit maybe required to perform just that.

    The two types of clamps commonly used for this

    purpose are shown on Figure 7 and Figure 8. ClampFigure7 consists of a high-speed high-voltage diodeD1 in series with a Zener diode. The Zener voltageshould be 20 to 30 volts above the output voltagereflected through the transformer;

    the zener should be sized to dissipate thefull energy stored in the leakage inductanceof the transformer.Clamp Fig.8 resembles a traditional snubbernetwork with different component values,that returns to the +DC and not to GND.Here again R1 may be omitted in someapplications if little or no ringing occurswhen D1 starts to conduct.

    Note: Even though the clamp in Figure 8 has a lowerefficiency than the clamp in Figure 7, it is used moreoften because it is using readily available, low-costcomponents compared to high-voltage zener diodes,which have to be special ordered.

    to

    R2

    VCC

    D1

    C1D2

    IPS10/15

    +DC

    R1

    +

    C2

    MOSFET

    TRANSFORMER

    Fig.6

    +DC

    D2ZENER

    MOSFET

    D1

    TRANSFORMER

    Fig.7

    R1

    MOSFET

    R2

    D1

    TRANSFORMER

    +DC

    C1

    Fig.8

    DRAIN CLAMP

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    Application Note AN-EVP001Rev 4 - Aug 2006 How to Design Low Cost, High Efficiency, SMPSs

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    Input Capaci t or:

    The input tank capacitor serves a critical purpose: it isbeing charged to the peak line voltage at twice the line

    frequency and reduces the ripple of the rectified DCfeeding the switching circuitry.

    Value:Use a capacitance value sufficient to keep the lowerpoint of the rectified and filtered AC line above 100V.

    For SMPS up to 35 watts, 2 F per watt is usuallyenough. See also ESR and Temperature rating below.

    Volt age Rat ing:For Domestic Applications use 200 or 250 voltscapacitors. For International Applications use 400volts, or 450 should the SMPS be capable of surviving300VAC Line Voltage for some period of time.

    ESR and T empera t ure Rat ing:It is critical to use a low ESR capacitor. Check with

    the manufacturers rating to make sure that 2F perwatt is enough to handle the peak current. 85C rated

    capacitors usually require more than 2f per watt andtherefore capacitors rated 105C are often a betterchoice.

    EMI-reduction Tip: Splitting the required inputcapacitance between two smaller size capacitorsconnected by a small inductor creates a low-cost filterthat can pass the EMI requirements for low-power

    SMPS. This is especially true when combined withother EMI reduction techniques like: MOSFET gate-drive optimization featured by the

    IPS15; Optimum Snubber Network; Low leakage inductance transformer; Low-noise transformer design.

    St ar t -up Resistor :

    The IPS 10 and IPS 15 both feature a starting current

    of approximately 120 A. The starting current must beprovided by an external start-up resistor. Once thisresistor has pulled the VCC up to approximately 10

    volts, the SMPS starts and the IC, which then requiresmore than 120A, is powered through AAIs PatentedModified Snubber Network or other suitable means.

    A 900K resistor is necessary to ensure proper start-up at a line voltage of 85VAC (Minimum for the

    International range), whereas a 2 M resistor issufficient for 180VAC minimum line voltage.

    Bia s Resist or :

    This resistor allows the user to operate at theswitching frequency which is best for the application interms of SMPS size, efficiency and EMI compliance.Use the information provided in the IPS15, 15H and 18datasheet documents to select the appropriate value.

    Please rememberthat the switching frequency affectsthe following three parameters:

    The transmitted power in watts is:

    1/2 x L x I2

    x F,

    where L is the primary inductance of thetransformer, I is the peak current in thetransformer, and F the operating frequency.

    Within limits, the higher the frequency the smallerthe inductance and therefore the smaller thetransformer.

    The switching losses which increase proportionallyto the frequency and are described thereafter:

    Sw i t ching Losses:Switching losses in the MOSFET heat the device anddegrades its efficiency. In comparison, the conductionlosses in the MOSFET are usually small. This meansthat the Ron of the MOSFET is not as critical aparameter as one may think for Flyback SMPSoperating in discontinuous mode.

    Addit io nal Swi t ching Losses:Additional switching losses caused by the capacitanceseen by the drain of the MOSFET. Such capacitance

    consists of the distributed capacitance in the primarywinding of the transformer in parallel with theMOSFET capacitance. Depending on the exactwaveforms, this stray capacitance can generate up to

    an additional 1/2 x C x V2

    of losses. This furtherdegrades the efficiency and generates more heat inthe MOSFET. This effect underscores the importanceof the following: Using a state of the art MOSFET with low stray

    capacitance; Giving preference to a low-capacitance design for

    the transformer.

    EMI:

    EMI is the third important factor to take in accountwhen selecting the operating frequency. Since theenergy spectrum tends to shows that harmonics ofhigher order are less energetic, operating at the lowestpossible frequency for the application usuallysimplifies compliance with regulatory requirements.This is why, SMPS operating at high frequency oftenrequire some additional EMI filtering components toreduce the effect of harmonics of low order.

    CRITICAL DISCRETE COMPONENTS

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    Frequency Selecti on Ti p:A good compromise for low-power SMPS below 30watts is to operate around 70kHz. This usually resultsin a small enough transformer, and makes it easy tomanage EMI. Operating above 120 kHz tends tosignificantly degrade the efficiency and should beavoided. Going below 25 kHz which could becomedangerously close to frequencies that humans and

    pets can hear, should also be avoided.

    Gat e Dr iv e Resist or s

    The internal driver of the IPS1x series has beendesigned to drive various MOSFETs up to 50 nC oftotal gate charge. For many small power application,this driver is greatly oversized, the MOSFET is driventoo fast and results in an increase of the EMIgenerated by the switching.

    The IPS1x series feature a split gate drive that allowstwo different resistors to independently control of therise and fall time of the gate signal. When operating

    the SMPS in discontinuous mode, there is no currentin the primary inductance when the MOSFET turnsON. By slowing down the turn ON time of theMOSFET, it becomes possible to significantly improvethe EMI signature without noticeably degrading theefficiency. The turn OFF needs to remains fast butcould be somewhat adjusted to obtain an acceptabletradeoff between efficiency and EMI.

    Line Over volt age Resist or sThis feature is available with the IPS1x series where aresistor divider can be placed between the rectified ACline and the ground to drive the overvoltage sensing

    pin.

    The divider should be calculated for this pin voltage tobe 3.9 volts when the line voltage reaches themaximum value at which the SMPS is required tooperate. Above this voltage, theMOSFEToperation isinhibited allowing the SMPS to withstand higher linevoltages without being permanently damaged.Beware of possible non-linearity of SMD resistorswhen operating them at high voltage. Through-holeresistors might be a better choice. The totalimpedance should be very high (around 10 MEG) tolimit the power dissipation.

    Outpu t F i l t e r :

    All SMPS require an output filter and, as a minimumfor non-critical applications like replacement of wall-

    mounted transformers, one capacitor of 470f ormore. Most applications however are moredemanding and require a LC or a RC filter to reducethe output ripple below the required level.

    With the IPS25, which requires a voltage drop of 1.3volts across the current-sensing resistor, it is usuallypossible to meet the ripple requirements by using thisresistor as a filter element in the output and to avoidusing an output inductor. The same applies to theIPS20/21 when self-powered, but an output inductorwill usually be required when operated with anauxiliary supply for maximum efficiency at high outputcurrents.

    Value:Since the SMPS operates at high frequency, the totaloutput capacitance to be used depends more on the

    ESR and the manufacturers rating in terms ofmaximum current.

    Volt age Rat ing:The capacitor should handle the required outputvoltage and the repetitive spikes generated by ESR atthe peak diode current. Again give preference to a lowESR capacitor, especially for low output voltageSMPS.

    Temperat ure Rat ing:The ESR will generate additional power losses whichwill raise the capacitors temperature well above

    ambient. 105C rated capacitors are usually the bestchoice since they usually combine lower ESR withhigher temperature rating. Undersizing the outputcapacitors will result in performance degradation andreliability issues.

    Please refer to the individual data-sheets for more details.

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    Fuse or Fusibl e Resist or , In-Rush Cur r ent :

    The fuse is essentially a protection device whichshould fail safe to prevent a fire hazard. Its voltage

    rating should correspond to the maximum line voltagethe SMPS is rated to operate at, Its current ratingshould take into account the high peak current whichresults from the full-wave rectification as discussed inparagraph Diode Bridge below. It should be of atype approved by the relevant safety agencies.

    In low power SMPS it is not unusual to replace thefuse by a Fusible Resistor that has a dual purpose:

    Limit the in-rush current to comply with someregulations and prolong the life of a line switch if

    any. (10 is the value most often used in 5 to10W SMPS)

    Perform like a regular fuse should the SMPS fails.

    With higher power SMPS, the in-rush current shouldbe limited differently to prevent generating unac-ceptable losses once the in-rush current is over.

    Suitable techniques include:

    Negative Temperature Coefficient Resistors; Series resistor shunted by a relay or an active

    device;

    Surge Pr ot ector (MOV or Ot hers):Surge protectors are typically used right after the fuseor the fusible resistor in order to protect the SMPS

    from damaging transients that could instantly destroysensitive components like high-voltage MOSFETs.

    Most frequently used protectors are Metal OxydeVaristors (MOVs). Other types of protectors appearless popular than MOVs for low-power, low-costSMPS. They include Gas-Filled Protectors, PrecisionSpark Gaps and Semiconductor Devices.

    The surge protector may be removed from certain low-power applications by taking advantage of theovervoltage sensing circuitry included in the IPS15line-side controller.

    X-CapacitorThis capacitor is usually placed in parallel with theMOV and contributes to the low-pass filter which helpsassure conducted EMI compliance.This safety-agency listed capacitor which is bulky andexpensive, may be removed from certain low-powerapplications by taking advantage of the EMI-reduction

    features offered by the IPS15 line-side controller.Please refer to Section Gate Drive Resistor page 10.

    Common-Mode Choke:

    A common-mode choke that is used in conjunctionwith one X-capacitor is the most common way tocomply with conducted EMI regulations. Thiscomponent may be replaced by a small and low-cost

    inductor for low-power SMPS using the IPS15. Pleasealso refer to Section Gate Drive Resistor page 10.

    Diode Br id ge:

    The diode bridge performs the full-wave rectification ofthe AC line voltage. Despite the short conduction timeof its diodes (on the order of 2mS) which results inhigh peak currents, a 1A current rating is usuallyadequate for most SMPS up to 35 watts. The voltagerating should be in accordance with that of the MOVbeing used.

    An additional safety margin should be used when the

    MOV is omitted. Up to 800 volts for a diode bridgeand 100 volts for discrete diodes would provide a costeffective margin of safety.

    Y-Capaci tor:The Y-CAP is usually connected between the SMPSprimary and secondary to reduce the common modeinjection of noise into the secondary which has anadverse effect on loop stability. The IPS1x seriescombined with a suitable design of the transformerallows the elimination of the Y-capacitor in low-powerSMPS.

    The IPS 20, IPS21 and IPS25 have been designed forease of voltage and current adjustments. For theseICs, the output voltage can be adjusted with anexternal voltage divider that in turn allows remotesensing when desirable.

    The maximum output current is set by a resistoracross which a voltage of 50mV or 100mV should bedeveloped respectively for the IPS20 and IPS21 and1.3volts for the IPS25. These ICs feature:

    A maximum noise rejection through current modedrive of the optocoupler,

    Independent voltage and current feedback loopcompensations.

    Setting Output Voltage and Current Limiting

    OTHER DISCRETE COMPONENTS

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    APPENDIX 1 - AAI MODIFIED SNUBBER NETWORK PATENT INFORMATION

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    Sw i t ch Mode Power Supp ly (SMPS):

    Switch Mode Power Supply (SMPS) are smaller andmore efficient than their linear counterparts. The inputis usually the AC mains line and the output is low-

    voltage DC. Most of the time the output is fully isolatedfrom the mains line as required by applications anduserssafety.

    The relatively high frequency of operation allows aconsiderable size reduction of the magnetic compo-nents and tank capacitors while achieving highefficiency and low output ripple. These areprerequisites for reducing the size of SMPS.

    Flyback:

    Flyback converter, sometimes also called BoostConverter, is one of the oldest SMPS techniques. It ispresent in all TV sets where it generates several KV of

    high-voltage required by the CRT. The utilization of theflyback technique to generate isolated low voltagesfrom an AC mains line is somewhat more recent andreally took off when high-voltage Power MOSFETswith breakdown voltages of 600 volts and abovebecame commonly available and very affordable.

    There is a tendency to use the term Flyback whenthe output is isolated from the input. The isolation fromthe mains line voltage requires a switching magneticcomponent with a minimum of two windings whichlooks like a transformer but does not transfer theenergy like one.

    The term Boost Converter is generally used whenthe switching magnetic component is a simple inductorthat can only generate an output voltage greater thanthe input voltage and is not electrically isolated from it.

    This application note deals with isolated SMPS, andwe will therefore exclusively use the term Flyback.

    The flyback approach has a number of uniquecharacteristics which explain its wide acceptance. Ithas few drawbacks when dealing with low-powerapplications and can be used for higher powerapplications in certain conditions.

    The first advantage of the flyback technique is the

    ability of a given design to operate within a wide rangeof input and output voltages. A factor of 10 is notuncommon, allowing a given SMPS to operate with agood efficiency from 85 to 265 volts AC at its input andstill have an output voltage fully adjustable, forexample, from 5 to 12 volts.

    This type of performance is impossible with a linearpower supply and a regular transformer, and difficult ifnot impossible with other types of SMPS. This

    explains why flyback is the solution of choice forTravel Power applications.

    The second advantage of the flyback technique is itssimplicity. Only one high-voltage switching component

    is required, which cuts on size and cost and explainswhy Flyback is the solution of choice for minimum costSMPS.

    The first and main disadvantage of the flybacktechnique is the relatively large amount of electromagnetic interference generated by the intenseswitching which is taking place involving fast voltageand current transients. For low-power SMPS up to 10watts, this application note will help the designermanage these undesirable effects by using acombination of inexpensive techniques. Power levelsup to 35 watts can be managed with a few more EMIcomponents. Power levels up to 75-100 watts are

    possible with more EMI filtering, especially whendealing with multiple outputs SMPS. Differenttechniques are definitely better suited than flybackwhen trying to achieve higher power levels with a fullyEMI-compliant design.

    The second disadvantage of the flyback technique hasto do with the Power MOSFET and secondary PowerDiode which must be sized to handle the peak current.The same components could deliver a lot more powerwith a different, but not as simple arrangement. Thisused to be a problem when power components wereexpensive, but has now become a very minor issue.

    Flyback Tr ansfor mer :

    As indicated before, the switching magneticcomponent used in an isolated flyback SMPS has atleast two windings which makes it look like atranformer. This is why it is often improperly calledTransformer. For convenience we will also call it atransformer even though it does not transfer theenergy like a transformer does. With reference toFigure 1, the energy is transferred as follows:

    1. When the MOSFET is ON, the rectified ACvoltage is applied across the primary winding andthe current starts to flow. The current increaseslinearly with time and the winding magnetically

    stores an energy equal to 1/2 x L x I2. (L is theinductance of the primary winding of transformerTX1, I is the current in the winding and theMOSFET.)

    2. When the MOSFET switches OFF, its drainvoltage raises above the rectified DC voltage andthe core is de-magnetized as current flowsthrough the Secondary Power Diode and chargesthe output capacitor.

    APPENDIX 1 - SMPS OVERVIEW AND DEFINITIONS

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    Figures 12 and 13 show the following waveforms:voltage on the MOSFET drain and current in theMOSFET and the primary of the transformer. We willdiscuss the differences between the two figures in thenext paragraph but before we proceed lets emphasizea very important point which governs the selection ofboth the MOSFET and the Secondary Power Diode.The primary and secondary windings are coupled and

    there is a transformer effect taking place which cannotbe ignored. As indicated before, the transformer effectis not used to transfer the energy from Primary toSecondary but it reflects voltages which have to betaken into account when selecting the breakdownvoltage of both the MOSFET and the SecondaryPower Diode.

    When the MOSFET is ON, the full DC voltage dividedby the turn ratio of the transformer appears across thesecondary terminals and this voltage, plus themaximum output voltage of the SMPS plus somesafety margin defines the minimum reverse breakdowvoltage of the Secondary Power Diode. Similarly,

    when the MOSFET turns OFF, The secondary PowerDiode becomes forward biased which applies theoutput voltage less a diode drop across the secondarywinding. This voltage multiplied by the turn ratiobetween Primary and Secondary adds to the rectifiedDC and governs the breakdown voltage requirementsfor the MOSFET.

    Discont inuous and Cont inuous Modes of

    Operat ion, Loop St abi l i t y :Looking at the MOSFET drain voltage on Figure 12,one can see three phases in the repeat pattern:

    1. The MOSFET is ON and the energy is stored inthe inductor;

    2. The MOSFET is OFF and the energy istransferred to the load;

    3. The MOSFET is still OFF and the inductor beingtotally demagnetized oscillates freely before a newcycle starts. This is a dead time typical ofdiscontinuous operation.

    With the transformer core being fully demagnetized atthe end of each cycle, it is possible to change theenergy stored from Zero to Maximum within one cycleresulting in a very fast transient response.In Figure 13, there is no longer any dead time. Thevery first cycle is the only one to start with a fullydemagnetized transformer core resulting in a currentincreasing linearly from zero like before. In continuousmode however, a new cycle starts before the core isfully demagnetized resulting in not dead time in thecycle and a traperoidal shape for the currentwaveform. Please note that the current waveformbuilds up progressively taking several cycles to reacha steady state. The energy transfer cannot thereforevary as quickly as before making the loop response of

    the discontinuous mode much longer than that of thecontinuous one.A complete SMPS resembles a servo-mechanism interms of stability and must be designed to be perfectlystable over the entire range of input line voltages andloading conditions. At no load all flyback powersupplies operate in discontinuous mode with a fastresponse but may enter continuous mode when

    operating with minimum line voltage and heavy loads.Such a situation should be avoided since it is almostimpossible to compensate the loop to be stable forboth discontinuous and continuous modes of opera-tion. The current-limiting loop is usually somewhat lesscritical and could be slow enough to tolerate a certainamount of continuous mode as the SMPS operates inan overload or short-circuit condition.

    Flyback SMPS can be designed to operate indiscontinuous mode only where the energy transfer ishigher with identical components. But this is beyondthe scope of this application note.

    Besides the loop stability issues, operating indiscontinuous mode has three major advantages:

    1. The transformer core being fully demagnetized atthe end of each cycle, the MOSFET turns ONunder very favorable contitions: there is noinductive current and only a small current spikecaused by the stray capacitances. This results invery small switching losses at turn ON.

    2. Since the MOSFET turn ON is not critical, it canbe made relatively slow in order to reduce the EMIsignature and save on EMI-reduction components.This is very easy to achieve with the IPS15 line-side PWM controller.

    3. Since the transformer core is fully demagnetizedwhen the MOSFET turns ON for a new cycle,there is no current flowing in the Secondary PowerDiode which makes its reverse recovery time anon-critical parameter. In continuous modehowever, an Ultra-Fast Diode or, voltagepermitting, a Schottky Diode must be used tominimize the switching losses as the MOSFETturns ON when there is still current flowing in thediode.

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    Figure 12 - DISCONTINUOUS MODE:

    Figure 13 - CONT INUOUS MODE:

    APPENDIX 2 - VOLTAGE AND CURRENT MOSFET WAVEFORMS

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