IEEE TRANSACTIONS ON INDUSTRIAL INFORMATICS, VOL. 10, NO. 1, FEBRUARY 2014 461

An Active Power-Decoupling Method forSingle-Phase AC–DC Converters

Mei Su, Pan Pan, Xi Long, Yao Sun, Member, IEEE, and Jian Yang, Member, IEEE

Abstract—This paper presents an active topology for power de-coupling in single-phase ac–dc converters, featuring the effectivesuppression of low-frequency power ripple which is an inherentproblem in single-phase energy conversion systems. This topologyis composed of an H-bridge circuit and a ripple power-decouplingcircuit. The ripple power-decoupling circuit shares one bridgearm with the H-bridge circuit so it has less additional compo-nents: just one IGBT, one diode, and an energy-storage inductor.Its modulation strategy is properly devised to coordinate theoperation of decoupling circuit and H-bridge circuit. To enhancethe power-decoupling performance, a direct ripple power-cancel-lation method based on energy-storage inductor is proposed. Amultiresonant controller with feed-forward control is introducedfor fast and precise current tracking. The effectiveness of thistopology has been verified by detailed simulation studies as wellas the laboratory prototype experiment results.

Index Terms—Active power decoupling, direct ripple power can-cellation, multiresonant controller, single-phase ac–dc converter.

I. INTRODUCTION

W ITH the development of the power industry, variousforms of power converters are required to meet the de-

mand of different energy-utilization technologies. Recently, nu-merous dc loads, like batteries, LED lamps, or dc sources likefuel cells and PV panels, are increasingly used in power system.Thus, in order to exchange power with ac distribution network,they require AC/DC converters to connect with the network. Insome applications, the single-phase ac network is very common[1]. The instantaneous power flowing from the ac side of asingle-phase ac–dc converter is time-varying with twice the gridfrequency [2]. If the ripple power is not properly filtered, itwill propagate to the device on the dc side and further affectits performance. For example, dc bus voltage ripple caused bylarge low-frequency ripple power decreases the efficiency of PVpanels [3] or generates the low-frequency flicker of LED lamps.In addition, the related ripple current will also affect the capacityand lifespan of electrochemical devices such as fuel cells andbatteries [4], [5].Usually, a passive energy buffer circuit—a capacitor or LC

filter, which is demonstrated in Fig. 1, is installed across the dcbuses to buffer the ripple power. In the capacitor case, a bulky

Manuscript received November 26, 2012; revised March 05, 2013; acceptedApril 17, 2013. Date of publication May 01, 2013; date of current version De-cember 12, 2014. This work was supported in part by the National High-techR&D Program of China (863 Program) under Grant 2012AA051601 and Grant2012AA051603 and by the National Natural Science Foundation of China underGrant 61203031. Paper no. TII-12-0788.The authors are with the School of Information Science and Engineering,

Central South University, Changsha 410083, China (e-mail: yaosuncsu@gmail.com).Digital Object Identifier 10.1109/TII.2013.2261081

Fig. 1. Conventional single-phase ac–dc converter.

aluminum electrolytic (AE) capacitor is a common choice, forit has the required high capacitance to reduce the voltage rippleto an acceptable value. Considering the short lifetime and largesize of AE capacitors, it is far from being a satisfying solution[6]. Regarding the LC filter case, due to the required low res-onant frequency, i.e., 100 Hz at a grid frequency of 50 Hz, thelarge matched passive components also preclude reduction insize, weight, and cost [7].To meet the needs for high power density and long lifetime

of the single-phase ac–dc converter several active decouplingmethods based on H-bridge converters were proposed [8]–[12].Generally, in these active decoupling methods, auxiliary energystorage elements and power control circuits are used to decouplethe inherent low frequency ripple power. Therefore, only a smalldc bus capacitor is needed to filter the remaining high frequencycomponents. As a result, it is possible to replace the AE capac-itor with a long-lifetime film capacitor [13].Some active decoupling methods adopt the active decoupling

circuit with an energy-storage capacitor installed on the dc side,which functions as a bidirectional dc–dc converter and real-izes the power decoupling by properly controlling the capac-itor’s power flow [8], [9]. In this case, the voltage stress of theswitches and capacitor in the decoupling circuit is high.Some other active decoupling methods install capacitors on

the ac side, in which the capacitor voltage are sinusoidal [10],[11], or apply an buck-type converter on the dc side [6]. Themaximum voltage of the capacitor in power decoupling circuitsis equal or lower than the dc bus voltage. Therefore, the de-coupling performance will be limited in a lower dc bus voltagecondition.Another choice is using an inductor as the energy-storage el-

ement. Indeed, the inductor’s power density is lower than thecapacitor’s. However, with the recent development of magnetic

1551-3203 © 2013 IEEE

462 IEEE TRANSACTIONS ON INDUSTRIAL INFORMATICS, VOL. 10, NO. 1, FEBRUARY 2014

Fig. 2. Main circuit of the proposed topology.

materials, storing the energy into an inductor could also be re-alized effectively [14]. In [12], the current in the inductor iscontrolled according to ripple current in the dc side. In thistopology, additional one pair of switches is required and the con-trol scheme is relatively complicated.Also, there exist other topologies based on flyback-type in-

verter with a power-decoupling function [15]. However, unidi-rectional power flow and limitation to low power level makethese topologies not good candidates.In this paper, another power decoupling topology for the

single-phase ac–dc converter is proposed. In Section II, themain circuit configuration of the topology is presented. Theripple power is stored in the energy-storage inductor. Com-pared with other topologies, less additional switches andpassive elements are needed in this topology and the controlalgorithm is relatively simpler. In Section III, a direct ripplepower-cancellation method based on it is used to reduce the dcbus voltage ripple, which is easy to realize. In Section IV, therelated control scheme and modulation strategy are designed toenhance the decoupling performance. The topology’s operationis described in details and its effectiveness of ripple voltagesuppression is confirmed through simulations and experimentsin Section V. Finally, Section VI presents the conclusion.

II. MAIN CIRCUIT CONFIGURATION

The main circuit of the proposed topology is shown in Fig. 2.By adding a power decoupling circuit to the conventionalH-bridge circuit, the proposed single-phase ac–dc topology isconstructed.As shown in Fig. 2, the power-decoupling circuit draws the

ripple power at twice the grid frequency from the dc bus andstores the energy in an inductor. The power drawn by power-de-coupling circuit is equal to the ripple portion of the inputpower . Thus, power flowing into capacitor and load

is approximately constant. The H-bridge circuit consistsof bridge arm A (S1, S2) and bridge arm B (S3, S4). To makethe full usage of the H-bridge, the decoupling circuit shares onearm with it. Therefore, just one IGBT (S5), one power diode(D1), and an energy-storage inductor are needed to form

Fig. 3. Equivalent circuit of the proposed topology.

the proposed decoupling circuit. It is clear that this topology re-duces the switch number.The average equivalent circuit of the proposed topology is

shown in Fig. 3. As shown in Fig. 3, and are theratio duties of S1, S3, and S5, respectively. Ratio duties of S2and S4 are complementary with ratio duties of S1 and S3. Onthe ac side, three bridge arms are treated as three controlledvoltage sources and , respectively. On the dc side,the H-bridge circuit and the decoupling circuit are expressed ascontrolled current sources and . is the switching cycleaverage current flowing from the H-bridge to the dc bus. Simi-larly, is the current flowing from the dc bus into the decou-pling circuit. In addition, it is assumed that the current flowinginto the buffer capacitor and load is and the current flowinginto the buffer capacitor is . In the proposed topology, ifthe component of twice the grid frequency in is preciselycancelled by will only contain other high-frequencycomponents which can be easily filtered by a very small buffercapacitor .

III. POWER-DECOUPLING PRINCIPLE

A. Power Ripple and Passive Filter

Here, we will briefly discuss the cause of power ripple andthe size of the corresponding passive filter. The power flow inconventional single-phase ac–dc converters is shown in Fig. 1.Assume that the input voltage is sinusoidal with the am-

plitude and angular frequency . It is expressed as

(1)

With unity power factor, the input current is

(2)

where is the amplitude of the input current.As shown in (3), the converter’s instantaneous input poweris decomposed into two terms, namely the average input

power and the ripple power . The ripple power is a time-varying term with twice the grid frequency oscillation

(3)

For most applications, the load power is controlled to be con-stant. Ignoring the converter loss, it is equal to the average inputpower . In (3), the amplitude of the ripple power is ,which is as large as the average input power . Usually, to bal-ance the input and load power, the entire ripple power is buffered

SU et al.: ACTIVE POWER-DECOUPLING METHOD FOR SINGLE-PHASE AC–DC CONVERTERS 463

Fig. 4. Power flow in the dc bus capacitor.

by a passive filter such as a capacitor. As a consequence, the ca-pacitor is charged and discharged as Fig. 4.The capacitor voltage is then determined by

(4)

where is the average value of the capacitor voltage.Rewriting (4) leads to the capacitor voltage to be

(5)

Applying Taylor series expansion for (5), we obtain

(6)

In (6), obviously, is twice the grid frequency componentin . In Fig. 4, notice that whenis discharged. So the capacitor voltage decreases. when

is charged and thus the voltageincreases. Assuming that the maximum value of isand the minimum value is , then the exchanged energy

during charging or discharging process is

(7)

Since the average bus voltage is , theamplitude of the voltage ripple is simplified as

(8)

With unity power factor, the system’s average power willbe . Thus, in a 200-W case, provided that the dc busaverage voltage is 120 V and the desired ripple voltage is below5%, the required capacitance in passive methods is up to 0.884mF according to (8), which is quite large.

B. Direct Ripple Power-Cancellation Method

As discussed above, the biggest problem in the single-phaseac–dc converter is the large twice the grid frequency ripplepower. The researchers have proposed various decouplingmethods to cancel twice the grid frequency power ripple.Mostly, their power decoupling methods were based on thecontrol of additional injected bus current. However, they werenot direct or accurate ways to eliminate the ripple power. In

this paper, a more fundamental and accurate method is given toachieve better power-decoupling performance.Assume that the current of the energy-storage inductor takes

the form of

(9)

where is the amplitude of the inductor current and is theadvance angle with respect to the input voltage. The stored en-ergy is expressed in

(10)

and the corresponding power is expressed in

(11)

where is the inductance value of the inductor. Thus, as (11)shows, the inductor current specified in (9) results in power flowat twice the grid frequency. Additionally, the correspondingpower in the input inductor is

(12)

Ignoring the switch loss and line resistor, the total power flowin the converter is shown in

(13)

In most applications, load power is constant and equalto . In order to cancel the ripple power, the correspondingpower in the energy-storage inductor should be controlledto be identical to the sum of and , namely

(14)

We simplify (14) as follows:

(15)

where . Therefore, and can be cal-culated by

(16)

(17)

464 IEEE TRANSACTIONS ON INDUSTRIAL INFORMATICS, VOL. 10, NO. 1, FEBRUARY 2014

Fig. 5. Phasor diagram for the proposed topology.

Substituting (16) and (17) into (9), the required current for theenergy-storage inductor to decouple the ripple power is attained.While designing the decoupling circuit, (16) can be used to se-lect its parameters. From (16), to decrease the current stress ofS5, it is better to choose a larger inductor. However, an inductorwith large inductance value means the one with large volumeand high cost. Thus, a tradeoff in selection between the induc-tance value and the current stress of S5 must be made.Generally, is small. If the power factor is unity, the advance

angle will be approximately 45 . Fig. 5 illustrates the pha-sors of the related currents and voltages, in which all vectors arerotating with radian frequency .

C. Discussion on the Current Stress

The current in bridge arm A does not change, so its currentstress is also the same. Because bridge arm B is shared by therectifier and the decoupling circuit, its current stress should bepaid more attention. The current phasors and which rep-resent the currents flowing into bridge arm B during the twodifferent half cycles are added in Fig. 5. As shown in Fig. 5, thecompensating current changes the current . Neglecting thecurrent ripple, the current during the two different half cyclescan be calculated as

(18)

where . Considering is very small, the maximum valueof the current flowing into bridge arm B could be approximated

as: .Therefore, the current stress level of bridge arm B is higherthan that without decoupling circuit. According to the resultabove, can be reduced by selecting a larger . However,the trade-off between the value of and current stress shouldbe made.

IV. MODULATION STRATEGY AND CONTROLLER DESIGN

A. Operation Analysis and Modulation Strategy

The proposed topology can be divided into two functionalblocks: the H-bridge circuit and the decoupling circuit. Being avoltage source converter, its switch signals in the same bridgearm are usually complementary. The proposed topology can be

divided into two functional blocks: the H-bridge circuit and thedecoupling circuit.Being a voltage source converter, its switch signals in the

same bridge arm are usually complementary. Therefore, theduty ratios satisfy the constraints: and .Assuming that the switch frequency is sufficiently high to applythe switch average modeling, then the average output voltage

is given by

(19)

Fig. 6 shows the operation modes of the decoupling circuit.The decoupling circuit shares bridge arm B with the H-bridgecircuit. The direction of inductor current is constrained by thediode in the main current path. Thus, the circuit has four modes.In Mode 1, S5 and S4 are on, while S3 is off. The currentflows through S5, and S4, from the positive dc bus to itsnegative one. is used to denote the voltage across the en-ergy-storage inductor. In Mode 1, , so charges themagnetic energy. It is referred to be a charging mode. In Mode2, S5 and S3 are off, while S4 is on The inductor current isfreewheeling through D1, and S4. In Mode 3, S4 is off, whileS5, S3 are on. The inductor current is freewheeling throughS5, , and S3. In both Modes 2 and 3, , so the mag-netic energy in the inductor almost does not change. In Mode4, S5, S4 are off, while S3 is on. The inductor current flowsthrough D1, and the antiparallel diode of S3. In this mode,

, so discharges the magnetic energy. Thus, thisis referred to as a discharging mode. Taking all of these opera-tion modes into account, the average voltage across the inductorduring one switching cycle is expressed as

(20)

To facilitate the analysis of modulation strategy, (19) and(20) are normalized by . Then, the following equations areachieved:

(21)

where and are within the range of . In (21), thereare two equations and three control inputs. Thus, one degree offreedom exists. The choice of the freedom degree determines theperformance of modulation strategy. To achieve the maximumdc voltage utilization ratio, the duty ratios should be designedproperly. From (21), the duty ratios which are related toH-bridge circuit, are limited by

(22)

Solve the inequality constraints above. The feasible domainof is defined as

(23)

In (23), it is found that the solution is always feasible ifthe following inequalities hold:

(24)

SU et al.: ACTIVE POWER-DECOUPLING METHOD FOR SINGLE-PHASE AC–DC CONVERTERS 465

Fig. 6. Operation modes of the decoupling circuit.

Fig. 7. Control block diagram for the proposed topology.

Usually, according to (24), is selected as

(25)

Furthermore, through the solved in (25), the values ofand can be calculated from (21). For simplicity, the switchpatterns are determined by applying the carrier-based modula-tion strategy.

B. Controller Design

There are two main control objectives in the proposedtopology: one is to eliminate the voltage ripple on the dc busand the other is to guarantee low THD of the input current.Here, two steps are taken to achieve the goals above. Theoverview control scheme diagram is shown in Fig. 7.

Step 1 is about the control of the H-bridge circuit, whichshould guarantee the unity power factor as well as low THDof the input current. Fortunately, the control scheme for theconventional H-bridge ac–dc converter will also be effective. Itconsists of an outer-loop voltage controller and an inner-loopinput current controller. The feedback signal of the dc busvoltage is filtered by a 100-Hz notch filter, which can avoidthird-order current caused by the remained double-grid fre-quency voltage ripple. The voltage error is regulated by a PIcontroller. The input voltage’s phase information is locked by asingle-phase phase-locked loop (PLL). The voltage controller’soutput is taken as the amplitude of input current. Multipliedby the input voltage’s phase, it is used as the reference for theinner-loop current controller. To track the reference currentprecisely, a proportional-resonant controller for the inner-loopcurrent control is introduced.

466 IEEE TRANSACTIONS ON INDUSTRIAL INFORMATICS, VOL. 10, NO. 1, FEBRUARY 2014

TABLE ICIRCUIT PARAMETERS

Step 2 is about the control of the decoupling circuit. Its aim isto eliminate the voltage ripple on the dc side. As mentioned inSection III, if the current of the energy-storage inductor is con-trolled according to (9), the decoupling circuit can successfullydecouple the ripple power. However, referring to Fig. 2, the cur-rent is positive. It seems that this topology is not suitable forthe ripple power decoupling. In fact, the energy stored in an in-ductor depends only on the square of inductor current, thus thesign of the current does not affect the ripple power decoupling.Therefore, the current reference could be modified as

(26)

which will deliver the same instantaneous power as that withthe current reference shown in (9).According to the Fourier analysis of (26), the current refer-

ence could be decomposed into

(27)

where is the dc component. From (27), it is easy to find thatmainly consists of dc, second-, fourth-, and sixth-order

components and the components after sixth order are almostnegligible. To track the reference quickly and precisely, amultiresonant controller integrated with a feed-forward controlis proposed. The current controller for the decoupling circuit isdesigned as

(28)

(29)

where is the error signal of inductor currentand represents the inverse Laplace operation.

denotes the feed-forward term.

V. SIMULATION AND EXPERIMENT RESULTS

A. Simulation Results

The above control scheme for the proposed topology is ver-ified by the simulation in MATLAB/Simulink environment. Thekey parameters are summarized in Table I.The simulation results between the conventional single-phase

ac–dc converter and the proposed topology are compared. The

Fig. 8. Waveforms of the dc bus voltage.

Fig. 9. Waveforms of the input current and energy-storage inductor current.

dc bus voltage waveforms are demonstrated in Fig. 8. Since justa small capacitor is installed in the dc bus, the dc bus voltageripple (peak to peak) in the conventional ac–dc converter is upto 70 V approximately, which account for 58.3% of the averagevalue (120 V). In contrast, the voltage ripple in the proposedtopology has been reduced to about 4 V, which is only 3.3%with respect to the average dc bus voltage.Fig. 9 shows the waveforms of the input current and the en-

ergy-storage inductor current. The energy-storage inductor cur-rent is almost a rectified sine wave, indicating the excellentperformance of proposed current controller. The zero-crossingpoints of lead those of the energy-storage inductor current

by approximately 45 , which is in accord with the analysisin Section III.

B. Experiment Results

A laboratory prototype was developed to evaluate theproposed topology. The main circuit consists of five IGBTs(1MBH60D-100), a diode (DSEI60-10A), an input inductor(3 mH), an energy-storage inductor (10 mH), and a dc buscapacitor (75 F). The other main parameters are listed inTable I. Both switching and sampling rate were 10 kHz inthis experiment. The control algorithm was realized in theTMS320F2812 DSP from Texas Instruments. The parametersof the voltage controller are . The param-eters of input current controller are . Theparameters of the current controller for decoupling circuitare . In order toverify the performance of this topology and its related algo-rithm, some comparison experiments between the conventionalsingle-phase ac–dc converter and the proposed topology wereconducted.

SU et al.: ACTIVE POWER-DECOUPLING METHOD FOR SINGLE-PHASE AC–DC CONVERTERS 467

Fig. 10. Experimental waveforms without active power decoupling circuit.

Fig. 11. Experimental waveforms with the proposed power decoupling circuit.

Fig. 12. Spectral analysis of the dc bus voltage.

With the proposed decoupling circuit disabled, the prototypeworks as a conventional single-phase ac–dc converter. Thevoltage ripple (peak-to-peak) of the converter’s dc bus voltageis up to 70 V, which is shown in Fig. 10. In contrast, as isdemonstrated in Fig. 11, the voltage ripple is reduced to around6 V with the proposed decoupling circuit, which is nearly tentimes less than that in the situation without active decouplingcircuit. Under the same situation, if the passive filter is used, itwould require a 1.46-mF capacitor to reach such a low-voltageripple level according to (8). In Fig. 11, we observed thatthe input power factor is nearly unity and the input current issinusoidal. Thus, it is clear that the proposed topology showsa good grid side performance.

Fig. 13. Experimental waveforms showing the effectiveness of the proposedpower decoupling circuit.

Fig. 14. Experimental waveforms with load changing from 75 to 50 .

Fig. 15. Experimental waveforms with voltage reference changing from 140to 120 V.

Fig. 12 shows the spectral analysis of dc bus voltage underboth conditions mentioned above. The second-order componentin the dc bus voltage drops from 34.26 to 2.18 V due to theproposed decoupling circuit.Fig. 13 illustrates the experimental waveforms with the pro-

posed decoupling circuit being enabled and disabled. Before theinactive point shown in Fig. 13, the voltage ripple is negligible.When the decoupling circuit is inactivated, the energy-storageinductor current decreases to zero immediately, meanwhile, thevoltage ripple increases to a large value suddenly.To show the dynamic response of the proposed topology, the

experiments with varying load and reference were conducted.

468 IEEE TRANSACTIONS ON INDUSTRIAL INFORMATICS, VOL. 10, NO. 1, FEBRUARY 2014

As shown in Fig. 14, when the load increases from 75 to 50abruptly, the dc bus voltage drops slightly and restores to thereference value in a very short time. When the reference of dcbus voltage steps down from 140 to 120 V, the dc bus voltagetracks the reference quickly. As illustrated in Fig. 15, the tran-sient is as short as 0.1 s and the overshoot is less than 5 V.

VI. CONCLUSION

This paper proposes a novel ripple power-decouplingtopology based on the energy-storage inductor, which reducesthe required capacitance of the dc bus capacitor greatly. Com-pared with other active power decoupling methods, fewerswitches and passive elements are needed in the proposedtopology. To enhance the performance of the power-decou-pling circuit, a direct ripple power-cancellation method andthe related modulation strategy are proposed. A laboratoryprototype based on TMS320F2812 DSP controller has beenbuilt to verify the feasibility and transient performance of theproposed converter. The experimental results demonstrate thatit could eliminate the ripple power effectively.

ACKNOWLEDGMENT

The authors would like to thank the laboratory membersH. Wang, Y. Liu, H. Wang, H. Dan, and G. Zhang, for theirvaluable suggestions.

REFERENCES

[1] P. T. Krein, R. S. Balog, and M. Mirjafari, “Minimum energy andcapacitance requirements for single-phase inverters and rectifiersusing a ripple port,” IEEE Trans. Power Electron., vol. 27, no. 11, pp.4690–4698, Nov. 2012.

[2] H. Hu, S. Harb, N. Kutkut, I. Batarseh, and Z. J. Shen, “Power de-coupling techniques for micro-inverters in PV systems—A review,” inProc. IEEE ECCE, 2010, pp. 3235–3240.

[3] S. B. Kjaer, J. K. Pedersen, and F. Blaabjerg, “A review of single-phasegrid-connected inverters for photovoltaic modules,” IEEE Trans. Ind.Appl., vol. 41, no. 5, pp. 1292–1306, Oct. 2005.

[4] H. Kim and K. G. Shin, “DESA: Dependable, efficient, scalable archi-tecture for management of large-scale batteries,” IEEE Trans. Ind. Inf.,vol. 8, no. 2, pp. 406–417, May 2012.

[5] G. Fontes, C. Turpin, S. Astier, and T. A. Meynard, “Interactions be-tween fuel cells and power converters: Influence of current harmonicson a fuel cell stack,” IEEE Trans. Power Electron., vol. 22, no. 2, pp.670–678, Mar. 2007.

[6] R. Wang, F. Wang, D. Boroyevich, R. Burgos, R. Lai, P. Ning, and K.Rajashekara, “A high power density single-phase PWM rectifier withactive ripple energy storage,” IEEE Trans. Power Electron., vol. 26,no. 5, pp. 1430–1443, May 2011.

[7] R.Maheshwari, S.Munk-Nielsen, andK. Lu, “An active damping tech-nique for small DC-link capacitor based drive system,” IEEE Trans.Ind. Inf., vol. 9, no. 2, pp. 848–858, May 2013.

[8] A. C. Kyritsis, N. P. Papanicolaou, and E. C. Tatakis, “A novel parallelactive filter for current pulsation smoothing on single stage grid-con-nected AC-PV modules,” in Proc. IEEE EPE, 2007, pp. 1–10.

[9] K.-H. Chao, P.-T. Cheng, and T. Shimizu, “New control methods forsingle phase PWM regenerative rectifier with power decoupling func-tion,” in Proc. IEEE PEDS, 2009, pp. 1091–1096.

[10] T. Shimizu, T. Fujita, G. Kimura, and J. Hirose, “A unity power factorPWM rectifier with DC ripple compensation,” IEEE Trans. Ind. Elec-tron., vol. 44, no. 4, pp. 447–455, Aug. 1997.

[11] H. Li, K. Zhang, H. Zhao, S. Fan, and J. Xiong, “Active power de-coupling for high-power single-phase PWM rectifiers,” IEEE Trans.Power Electron., vol. 28, no. 3, pp. 1308–1319, Mar. 2013.

[12] T. Shimizu, Y. Jin, and G. Kimura, “DC ripple current reduction on asingle-phase PWM voltage-source rectifier,” IEEE Trans. Ind. Appl.,vol. 36, no. 5, pp. 1419–1429, Oct. 2000.

[13] F. Schimpf and L. Norum, “Effective use of film capacitors insingle-phase PV-inverters by active power decoupling,” in Proc. IEEEIECON, 2010, pp. 2784–2789.

[14] J. X. Jin, W. Xu, X. Y. Chen, X. Zhou, J. Y. Zhang, W. Z. Gong, A.L. Ren, and Y. Xin, “Developments of SMES devices and potentialapplications in smart grids,” in Proc. IEEE ISGT-Asia, 2012, pp. 1–6.

[15] G. H. Tan, J. Z. Wang, and Y. C. Ji, “Soft-switching flyback inverterwith enhanced power decoupling for photovoltaic applications,” IETElectr. Power Appl., vol. 1, no. 2, pp. 264–274, Mar. 2007.

Mei Su was born in Hunan, China, in 1967. Shereceived the B.S., M.S., and Ph.D. degrees fromthe School of Information Science and Engineering,Central South University, Changsha, China, in 1989,1992 and 2005, respectively.Since 2005, she has been a Professor with the

School of Information Science and Engineering,Central South University, Changsha, China. Herresearch interests include matrix converters, ASD,and wind energy conversion systems.

Pan Pan was born in Hubei, China, in 1981. Sheis currently working toward the Ph.D. degree at theSchool of Information Science and Engineering,Central South University, Changsha, China.Her research interests include the application of

power electronics and new energy power generation.

Xi Long was born in Hunan Province, China, in1988. He received the B.E. degree from the CentralSouth University, Changsha, China, in 2010, wherehe is currently working toward the M.E degree inelectrical engineering.His research interests include power electronic

converters and distributed power generating system.

Yao Sun (M’13) was born in Hunan, China, in 1981.He received the B.S., M.S., and Ph.D. degrees fromthe School of Information Science and Engineering,Central South University, Changsha, China, in 2004,2007, and 2010, respectively.He is currently with the School of Information

Science and Engineering, Central South University,China, as a Lecturer. His research interests includematrix converters, micro-grids, and wind energyconversion systems.

Jian Yang (M’09) received the Ph.D. degree inelectrical engineering from the University of CentralFlorida, Orlando, FL, USA, in 2008.He was a Senior Electrical Engineer with Delta

Tau Data Systems, Inc., Los Angeles, CA, USA, from2007 to 2010. Since 2011, he has been with Cen-tral South University, Changsha, China, where he iscurrently an Associate Professor with the School ofInformation Science and Engineering. His main re-search interests include control application, motionplanning, and power electronics.