HANCOCK

f T J 163 . A87 EP- RR13

A SURVEY OF ISOLATION AMPLIFIER

CIRCUITS

I. D. G. MACLEOD

February, 1967

Department of Engineering Physics

Research School of Physical Sciences

TUE AUSTRALIAN NATIONAL UNIVERSITY

berra, A.C.T., Australia.

's]

3 i

TJ163.A87 EP-RR13.

924135

A . N . U . L I B R A R Y

EP-RR 13

This book was published by ANU Press between 1965–1991.

This republication is part of the digitisation project being carried out by Scholarly Information Services/Library and ANU Press.

This project aims to make past scholarly works published by The Australian National University available to

a global audience under its open-access policy.

A SURVEY OF ISOLATION AMPLIFIER CIRCUITS

by

LD. G. MACLEOD

February, 1967

Publication EP-RR 13

Department of Engineering Physics, Research School of Physical Sciences,

THE AUSTRALIAN NATIONAL UNIVERSITY,

Canberra, A. C, T. Australia.

ft? R S P H Y S .S

Ll B RA R ^

" 6 FEB 1968

CONTENTSSUMMARY Page

L ist of Symbols

1. Introduction. 1

2. Am plitude m odulation with tra n s fo rm e r coupling 22 .1 Square wave c a r r ie r (chopper type) 3

2 .1 .1 M echanical chopper 42 .1 . 2 D ual-em itte r tra n s is to r chopper 62 .1 . 3 F ie ld -effec t tra n s is to r chopper 72 .1 .4 P ho to -e lec tric devices 82 .1 . 5 Sem i-conductor diode ring-m odulator 9

2. 2 Sinusoidal c a r r ie r 92. 2 .1 V ibrating capacitor m odulator 102 . 2 . 2 F. E. T. m odulator 112. 2. 3 V ariab le-capacitance diode bridge. 11

3. O ther fo rm s of modulation with tra n sfo rm e r isolation 123.1 P u lse or square wave c a r r ie r . 133 .2 Sinusoidal c a r r ie r 15

4. Iso lation with a d ifferen tia l am plifier 154 .1 Common-mode re jec tion without p r io r am plification 164. 2 Common-mode re jec tion with p r io r am plification 17

5. Iso lation by optical coupling 175.1 Photon-coupled isolation with a frequency m odulated 17

c a r r ie r5. 2 D igital light coupling. 19

6. Iso lation by sam pling and storing. 20

7. Iso lation by coupled flux 207. 1 P u lse relaxation am plifier 207. 2 Second harm onic type m agnetic m odulator. 22

8. O ther methods of iso lation 238 .1 "O verlapped” c irc u it 238 .2 Iso lation by e lec tro -m agnetic coupling 258. 3 A coustical coupling 258 .4 M echanical coupling 26

9. Conclusions 26

10. Appendices 2710.1 Floating am plifier 2710. 2 T em perature-dependent d rift reduction in the

F airch ild fiA709 C operational am plifier 2810. 3 Iso lated d rive c ircu it fo r MOST’S. 31

11. R eferences 33(i)

SUMMARY

The re jection of re la tive ly la rge common-mode voltages

and the sim ultaneous am plification of sm all signal voltages is a

common requirem ent in instrum entation and control. Conven­

tional d ifferen tia l am plifiers a re usually satisfac to ry , but in

applications involving large o r rapidly changing common-mode

voltages, m ore effective " iso lation” of the signal voltage is

required .

Methods of coupling a signal voltage between two d iffe r­

ing common-mode potentials a re examined, and an a ssessm en t

is made of each method’s su itab ility fo r p a rticu la r applications.

LIST OF SYMBOLS

A S5 a. c. coupled a m p lif ie r

A e= d. c. coupled a m p lifie r

D 6= dem odulato r

D r d 2s diodes

E .Slg

E = s ig n a l voltage

Ecmes com m on-m ode voltage

E + outs= output voltage

L P F er lo w -p ass f i l te r

M s m odulator

MM s r m onostab le m u ltiv ib ra to r

PS s= floating pow er supply

Q c= t r a n s is to r

RFO s= ra d io -freq u e n cy o sc illa to r

ST s= Schm itt t r ig g e r

T c= t r a n s fo rm e r

(iii)

1. INTRODUCTION1

The output voltage of a transducer is often accompanied by a large common-mode voltage to ground. This is particularly true of transducers associat­ed with e lectrical machinery.

The large common-mode voltage may not be a problem when the indicat­ing equipment, for example, an oscillograph, does not have to be grounded. In many cases, however, the transducer output voltage must be made available to equipment which is grounded. Provided that the common-mode voltage is only a few volts (for transistorized circuits) or a few tens of volts (for vacuum tube circuits), the tran s­ducer's output may be applied to a conventional differential amplifier, this amplifier being relatively insensitive to the common-mode v o l t a g e . 5

When the common-mode voltage is greater than a hundred volts or so, it becomes impracticable to use conventional differential amplifiers, particularly when the transducer output voltage is small and a high common-mode rejection ratio (CMRR) is required. In such cases an amplifier with an inherently greater CMRR and allowable range of common-mode voltages is required. Such amplifiers will be called "isolation amplifiers".

An examination of commercially available isolation amplifiers has re ­vealed only a small number which could be suitable. Typical of these are the Philbrick Researches SP2A^ which operates on the principle of section 2.1. 5, the Brush # 13-4215-80 high voltage d. c. amplifier5 which chops the input signal and isolates with a transform er, and Dynamics Instrument Corp. # 6050 wideband differential d. c. amplifier6 which achieves isolation by modulation and demodulation of a 100 kHz carrier.

In some applications, certain characteristics of the above amplifiers ( e. g. - 100 [x V/day drift for the SP2A, frequency response of only 0-200 Hz for the

#13-4215-80, and - 300 V peak common-mode voltage for the #6050), may be a limitation. In general, these amplifiers are also too expensive for installations which require a large number. Considering the wide range of possible applications and the accompanying range of performance requirements, a number of possible isolation amplifier circuits is examined below and relevant characteristics are given. This range of circuits should allow the simplest (and hopefully the cheapest) sa tis­factory circuit to be chosen for particular applications.

2. AM PLITUDE MODULATION WITH TRANSFORMER COUPLING

A tra n s fo rm e r can ach ieve a la rg e CMRR w hen a s ig n a l vo ltage w ith a com m on-m ode com ponent to ground is applied to the p r im a ry w inding and the s ig n a l voltage is re c o v e re d fro m th e seco n d ary winding. T h is re je c tio n is in c re a se d by sh ie ld ing th e tra n s fo rm e r , and fo r low com m on-m ode freq u en c ies . U n fo rtu n a te ly , th e fre q u e n c y re sp o n se of a conventional t r a n s fo rm e r does not extend down to d. c.If the s ig n a l to be iso la ted contains d. c. and low -frequency com ponents, the m ethod of m odulating a c a r r i e r frequency by the s ig n a l vo ltage, co u p lin g th e m odulated c a r r i e r to ground v ia a tr a n s fo rm e r , and then dem odulating th e c a r r i e r to ob ta in an iso la ted v e rs io n of th e input s ignal, m ay be u sed (see F ig u re 1).

Note: a m p lif ie r m ay p rece d e dem odu l­a to r if s ig n a l levels a r e low.

Feedback (if req u ired ) Iso la tio n w ith p r io r am p lifica tio n

E . s ig

E cm

■® Output

i M

Feedback (if req u ired )

Iso la tio n w ithout p r io r am p lifica tio n

F ig u re 1. ISOLATION WITH A TRANSFORMER

AMPLITUDE MODULATION WITH TRANSFORMER COUPLING 3

A common form of modulation is am plitude modulation, w herein the am pli­tude of the c a r r ie r waveform envelope is made to vary in sympathy with the am plitude of the signal. When am plitude m odulation is used, it is often n ecessa ry to employ negative feedback to overcom e n on -linearities and in s tab ilities in the tra n s fe r c h a rac t­e r is tic s of the am plifier and m odulators (see F igure 1). Typical reaso n s for th ese defects a re ; sensitiv ity of the demodulation method to varia tions of the m ark -sp ace ra tio of a m echanical chopper, changes in the effective tra n s fo rm e r ra tio with signal level, and non -lin earitie s in so lid -s ta te m odulators.

The provision of negative feedback a t the input to the iso lation am plifier involves the sam e problem s as those which it is being used to overcom e and if the feedback voltage is coupled by a tran sfo rm er, i t too m ust be modulated and dem odulat­ed. The overa ll linearity of the iso lation am plifier w ill be lim ited by the linearity of the feedback and the balance between the dem odulators. Techniques such as biasing and high signal levels can be used to im prove the linearity of the feedback c irc u it but the need for feedback can be a serious disadvantage.

Fortunately, if the perform ance requ irem en ts a re not too s tr ic t , ca re in the design of the forw ard signal path can som etim es avoid the need fo r feedback.

The CMRR achieved with the iso lating tra n s fo rm e r is very la rg e a t d. c. and d ecreases with frequency because of s tra y capacitances in the tra n s fo rm e r and capacitive cu rren ts flowing in both the shields and the windings. The re jec tio n achiev­ed w ill depend to som e extent on the d isposition of shields w ithin the tra n s fo rm e r and the im pedances of the windings but in one com m ercial c ircu it using a sc reened t r a n s ­fo rm er, a CMRR of approxim ately 120 dB a t 60 cycles p e r second is re p o rted 5. If the re jection ra tio at h igher frequencies is insufficient, the overa ll re jec tion can be in ­c reased by amplifying the signal before isolation.

If the frequency response of the d. c. coupled am plifie r is too re s tr ic te d , it is possib le to p a ra lle l an a. c. coupled signal path (using a wide band iso lation t r a n s ­form er) with the d. c. signal path, and combine th e ir output a fte r iso lation (as shown in Figure 2) to obtain a g re a te r band-width . O verall negative feedback may be r e ­quired to smooth out undulations in the frequency response n ear the c ro sso v e r f r e ­quency. The above approach can be adopted if any of the c a r r ie r- ty p e iso lation am plifiers to be described below is sa tisfac to ry for a given application ap art from a lim ited frequency response.

2 .1 Square-W ave C a r r ie r (Chopper Type)

A square-w ave c a r r ie r may be produced by "chopping" the input signal. This is usually achieved by a lte rnately applying and rem oving the input signal (half­wave chopping) o r periodically rev e rsin g its po larity (full-wave chopping). The chopped signal may be applied to the isolation tra n sfo rm e r d irectly , o r a fte r som e p re lim in ­ary am plification.

4 AMPLITUDE MODULATION WITH TRANSFORMER COUPLING

A. C. Signal pathInput $ Output

> \y• I

i

D. C. Signal PathFigure 2: PARALLEL A. C. and D. C. SIGNAL PATHS

The input signal can be recovered from the modulated ca rrie r by phase- sensitive rectification and low-pass filtering (see Figure 3). In this case, depending on the acceptab le ripple amplitude, unless complex filters are used, the frequency response is limited to about one-tenth of the carrier-frequency for half-wave chopp­ing and about one-quarter of the carrier frequency for full-wave chopping. It is often thought that there is a theoretical limitation to the band-width of the input signal recoverable from the chopped signal, of some fraction of the chopping frequency. It has been shown that this limitation is practical rather than theoretical*8 and the’bver- lapped" circuit examined in section 8. 1 is a case of a chopper-type amplifier whose frequency response does not depend primarily on the chopping frequency.

2.1.1 Mechanical Chopper

Refs. 9, 10, 11, 12, 13 (see Figure 3)

The mechanical chopper has the inherent advantages of a very high "off” to "on" resistance ratio, virtually no offset voltage or current and very good isola­tion of the driving signal. Unfortunately, being mechanical, it has a limited life

As an example that (in the ideal case) bandwidth is not a function of the fundamental carrie r frequency, consider an ideal full-wave chopper amplifier sim ilar to that shown in Fig. 3. The amplified modulated signal, Vm(Xj = - kjA V depending on the polarity of the input chopper (with k = input transform er ratio and A = amplifier gain). The demodulated signal = - vmo^ depending on the polarity of the output chopper (with k2 = output transform er ratio). If the choppers are in perfect syn­chronism and take only a small time to change polarity, the output signal, VQu =+ ( + ^ k2 Vin) and - (- k^ k Vin ) respectively as the choppers are in the positive and negative positions, i. e. Vout = k^ k2 Vin continuously, complete with all fre ­quency components.

AMPLITUDE MODULATION WITH TRANSFORMER COUPLING 5

Output

Modulated Signal

Mark SpaceHalf Wave Circuit

Full Wave CircuitNote: Choppers can be mechanical or solid state.

FIGURE 3. "CHOPPER" TYPE CIRCUIT (SQUARE WAVE CARRIER)

and reliability, suffers from contact bounce (which can be reduced with mercury wett­ing), and is relatively slow in operation.

Conventional mechanical choppers normally have a maximum chopping rate of about 400 Hz, but choppers which will operate at up to 2, 000 Hz have been report-

6 AMPLITUDE MODULATION WITH TRANSFORMER COUPLING

9 10ed ’ . These high speed choppers can be used with conventional c ircu its toobtain a useful frequency response of 100 Hz o r so.

An advantage of the m echanical chopper is that the driving signal need not be isolated from ground. M echanical choppers can be used e ith e r before or a fte r am ­plification of the tran sd u cer output but, depending on the method of demodulation, overall feedback is often requ ired , thus com plicating the c ircu it.

Because of its low "on" re s is tan ce and high "off" re s is ta n c e the m echan­ical chopper is suitable for a wide range of load im pedances. With care , long-term stab ilities of the o rder of sev e ra l pi V can be achieved.

2 .1 . 2 D ual-Em itter T ra n s is to r Chopper

Refs. 14,15, 16

The use of junction tra n s is to rs in chopping applications has been trea ted extensively in the lite ra tu re . 1^» 1®» 21,22 The d u a l-em itte r tra n s is to r is anatural extension of the junction tra n s is to r used as a chopper and in th is application it is superio r to the conventional device, which w ill not be considered further.

When tra n s is to rs a re being operated as sw itches, th e ir "on" voltage may be, reduced substantially if they a re operated in the inverted mode, i. e. , by in te r ­changing the norm al co llecto r and e m i t t e r . 20 jf two tra n s is to rs a re o p e ra t­ed "back to back", th e ir "on" voltages balance to produce a much lower offset voltage. The varia tions in the combined offset voltage with tem p era tu re a re a lso substantially less than those of the individual tra n s is to rs , if the p a ir is physically and therm ally matched. The dua l-em itte r tra n s is to r is functionally equivalent to such a pair. C urrently available devices have offset voltages of - 50 f iV maximum, leakage cu rren ts of 20 pA, tra n s fe r re s is ta n c es of 25 m ß , offset voltage d rifts of -hzV/ °C, tran sitio n capacities of 2 pF and "on" re s is ta n c es of 100 ß 14, 15. These devices a re suitable sw itches for load im pedances of i= lMß and voltages down to m illivolts, and can in many cases be substituted for the m echanical choppers in the c ircu its of section 2 .1 .1 with the attendant advantages of fa s te r operation and longer life. If a com m on­mode voltage is associated with the signal, the d rive voltage fo r turning these t r a n s is ­to rs "on" and "off" has to be iso lated from ground. A facto r which can be a serious disadvantage in low -level c ircu its is that at the instan t of turning the tra n s is to rs on (and off) the sudden change in d rive voltage is coupled into the signal c ircu it because of the dynamic m ism atch existing between the two em itte r tran s itio n capacitances.The "turn-off" tran sien t is usually w orse than the "turn-on" because during "turn-on", these capacito rs charge through the t ra n s is to r 's re la tive ly low "on" re s is tan ce w here­as during "tu rn-off" the capacito rs d ischarge through the higher s ig n a l-c ircu it r e s i s ­tance. The effect of these tran s ien t spikes is slightly reduced by using driving signals with a fast r is e - tim e , which reduces the period of dynamic m ism atch and thus shortens the transien t.

AMPLITUDE MODULATION WITH TRANSFORMER COUPLING 7

If the spikes are smoothed by a low-pass filter, they produce an equivalent offset voltage which will vary as the characteristics of the spikes vary with changes in rise-tim e, drive voltage and transition capacitance. The magnitude of this offset, calculated for 100% dynamic mismatch,+is approximately 2mV for a 100 kß input impedance, 5 kHz chopping rate (with - IV drive signal) and 2 pF transi­tion capacitance. There is a reasonable degree of matching between the capacitances and in practice this offset voltage can be a small fraction of a millivolt and have reasonable stability so that it can be compensated. The feed-through of switching spikes can, depending on the applied signal voltage and load impedance, limit the chopping rate to several kHz and thus limit the useful band-width to a few hundred Hz. In the case where the transistor chopper is used prior to any amplification of the transducer output, care will need to be exercised in the design of the isolating tran s­form er since the transients tend to excite oscillations in the transform er stray re ­actances which may be troublesome in the following circuits.

Because of the fast operation of the dual-emitter transistor, the mark-space ratio stability depends mainly on the drive-voltage timing accuracy. For this reason, feedback may not be necessary.

The dual-emitter transistor will be suitable as a chopper after some preliminary amplification of a low level transducer’s output or when acting directly on a high level (say > 50 mV) low impedance (say < 50 ß ) transducer output. In these cases the isolating transform er design will not be as difficult because of the lower circuit impedances and higher signal level compared to the spike amplitude.

2.1. 3 Field Effect Transistor Chopper

Refs. 23-29. (See Figure 4)

Source electrode Drain electrode

Gate electrode

Figure 4: FIELD EFFECT TRANSISTOR SCHEMATIC

The field effect transisto r (F. E. T .) is roughly equivalent to a re s is t­or whose resistance between the source and drain is variable over a wide range of from a few hundred ohms to hundreds of megohms by a control voltage applied to the gate. This device has very low offset voltages ( < . 1 ^iV)^ and low leakage currents in the tumed-off state. By the use of a square-wave gating voltage, the F. E. T. can be used in a series or shunt divider circuit as a chopper. In this application, the F. E. T. is comparable to the dual-emitter transistor in that it is a satisfactory

8 AMPLITUDE MODULATION WITH TRANSFORMER COUPLING

chopper for modulation of the low -level tran sd u ce r outputs p r io r to any am plification, ap art from the feed-through of switching spikes, which in th is case a re la rger. This is because th e re is no balancing between a p a ir of devices and, while the in te r ­electrode capacitances a re of the sam e o rder, a la rg e r gating voltage is required .By coupling an antiphase gating w aveform to the d ra in via a capacitance equal to Cgd of F igure 4, a la rge reduction in the spikes can be achieved. This reduction is lim ited, however, by the dependence of Cgd on applied voltage and tem peratu re. In the case where the modulation is to be perform ed d irec tly on h igh-level low- im pedance tran sd u ce r outputs o r a fte r am plification of the low -level transducer outputs, the F. E. T. is again com parable to the dual em itte r tra n s is to r in that because the spike feed through is not as se rio u s, it is a sa tisfac to ry chopper.

The gating voltage w ill have to be isolated for a s e r ie s chopper, but advantage can be taken of the insu lated-gate F. E. T. ’s low gate cu rren t to sim plify the design of the gate voltage iso lating tran sfo rm er. A technique which can be used fo r th is sim plification is outlined in Appendix 10. 3. The F. E. T. is suited for load re s is tan ces of from kß ’s to Mß ’s depending on the device and w hether it is connected as a s e r ie s o r shunt chopper.

2 .1 .4 P hotoelec tric Devices

Refs. 30, 31, 32, 33

The p h o to -tran s is to rs and photo-F. E. T. ’s which a re available can be used in much the sam e m anner as the norm al devices except that in this case, the d rive voltage is not requ ired and is rep laced by a light source which is turned on and off as required . Except fo r the case of p h o to -re s is to rs , the tran s ien t spikes a re not avoided, because the photo-junction, in effect, supplies the energy necessary to tu rn the device on and the changes in e lectrode voltages a re s ti l l coupled by in te r­electrode capacitances to the signal c ircu it.

P h o to -re s is to rs a re alm ost ideal as chopper elem ents, having good "off” to "on” re s is tan ce ra tio s and low offset voltages and drifts ( - 1 ^ V / °C and - 1 fi V / week) together with excellent iso lation of the driving signal, but they a re com paratively slow in operation (for solid s ta te devices) and they a re not generally used at chopping ra te s above a few hundred Hz22. Using a conventional half-wave c ircu it, th is lim its the attainable frequency response to about twenty Hz. The "on" and "off" re s is ta n c es of p h o to res is to rs can be varied over a wide range and they a re suitable for load re s is tan ces of tens of kß ’s to Mß ’s.

P h o to -re s is to rs a re ra th e r uncerta in in th e ir chopping because of slow operation and they a re norm ally used in se r ie s -sh u n t p a irs with overall negative feedback.

P h o to -re s is to rs need not be considered fo r chopping a fte r p re lim in ­a ry am plification of the tran sd u ce r outputs since d u a l-em itte r tra n s is to rs o r

AMPLITUDE MODULATION WITH TRANSFORMER COUPLING 9

F. E. T. 's a re superio r in th is application because of th e ir fast operation.

2 .1 . 5 Sem iconductor Diode R ing-M odulator

Refs. 12, 13, 34. (See Figure 5)

Output

AAAASwitching

drive

5. SEMICONDUCTOR DIODE RING -MODULATORFigure

The diode bridge is made to conduct a lte rnate ly on one side and then the o ther by the switching drive. The signal voltage is thus tran sm itted on a lte rnate cycles to opposite sides of tra n s fo rm e r T2 p rim ary . To avoid feed-through of the switching drive, very good matching of the tra n s fo rm e rs and diodes and low s tray capacitances a re required . Because of the forw ard voltage of the diodes, th is modu­la to r is not suited to low voltages but ra th e r to low cu rren ts from high im pedances. Schw artz13 gives a d rift figure of 10“7 am ps over a tem p era tu re range of 20°C to 50 °C, w hilst Keonjian and Schm idt3'1 have detected cu rren ts as low as 10 _1^ am ps at room tem p eratu re . D uring the switching period, spikes w ill be coupled to the output c ircu it by dynamic m ism atch of the diode capacitances. I t is not considered tha t th is m odulator is satisfacto ry for the signal levels involved p r io r to am plification of low- level tran sd u ce r output voltages. D u al-em itte r tra n s is to rs o r F. E. T. 's a re m ore suitable for u se afte r such am plification. A ring-m odulato r can be useful for t r a n s ­ducers whose output is cu rren t ra th e r than voltage, and very fast chopping ra te s and good frequency response (MHz) a re possib le.

2. 2 Sinusoidal C a rr ie r

With some solid s ta te m odulators, the in te re lec trode capacitances cause spikes to be coupled into the signal c ircu it during the r is e and fall of a sq u a re - wave gating signaL Because of non -linear capacitances, differing c ircu it re s is tan ce during tu rn -o ff and tum -on , and asym m etrica l gating signals, the charge tra n s fe rre d

10 AMPLITUDE MODULATION WITH TRANSFORMER COUPLING

to the signal circuit during turn-off does not normally cancel that transferred during tum-on. As a result, a net charge is injected into the signal circuit during every chopping cycle and an offset voltage develops, with resultant drift voltages as the rate of charge injection varies. It can be shown that the offset voltage and resultant drift can be reduced by using a sine-wave in place of a square-wave carrier2** . jn other circuits such as the vibrating capacitor modulator, a sinusoidal carrier is preferred because it is easier to obtain than a square-wave carrier.

When recovering the input signal from the modulated carrier, phase sensitive demodulation must be used if feedback is to be applied. The demodulation is usually followed by a low-pass filter and in this case the useful band-width extends to about one-tenth of the carrie r frequency. It has been shown that the use of a suit­able demodulator could increase the useful band width, perhaps to greater than the ca rrie r frequency® but the circuits required a re too complex to be practicable.

2. 2.1 Vibrating Capacitor Modulator

Refs. 11, 15, 35. (See Figure 6)

^coupling

Output

variable

Figure 6: VIBRATING CAPACITOR MODULATOR

This type of modulator is particularly suited to high impedance circuits and the measurement of minute currents, and is based on the principle that if a capacitor having a d. c. voltage across it has its capacitance varied, there will be a change of voltage across the capacitor and a current through it. The magni­tudes of this current and voltage depend on the source impedance, signal voltage, capacitance change and rate of capacitance change. For the application being considered, the capacitance can be varied in an approximately sinusoidal manner by a vibrator, at a frequency of up to several kHz, giving a useful band width of 100 Hz or so.

This type of circuit generally requires feedback to linearize the tran s­fer function against changes in vibration amplitude and stray capacities. The driving signal for the vibrator need not be isolated because the capacitor plate can be insulat­ed from the vibrator.

AMPLITUDE MODULATION WITH TRANSFORMER COUPLING 11

The d rift behaviour of the vibrating capacito r m odulator, when used fo r m easuring sm all voltages from re la tive ly low source im pedances, is re la tively poor, un less a g rea t deal of ca re is taken in the design and p repara tion of the capacit­o r p la tes. V ariations in contact potential of the capacito r p lates cause voltage fluctuations and drift. Day to day d rif ts of 10“15 am ps with high im pedance sources and 1 mV with low impedance sources have been achieved25.

2. 2. 2 Field Effect T ra n s is to r M odulator

The F. E. T. may be used as a m odulator for a sinusoidal c a r r ie r as w ell as for a square-w ave c a r r ie r . The F. E. T. ac ts as a variab le re s is tan ce whose value depends on the applied ga te-so u rce voltage. By varying the g a te -so u rce voltage in a sinusoidal m anner about a b iasing voltage, an approxim ately sinusoidal c a r r ie r w ill be generated . Such a m odulator has been used at 1 kHz in a d. c. am plifier, to produce a c ircu it having a d rif t ra te of 1 fiV /°C with a 30 kß input im pedance22 .

2. 2. 3 V ariable Capacitance Diode Bridge

Refs. 4 ,13 , 36

In a sem iconductor diode, the junction capacitance

C = ( 1 )TTv d + V

w here V = applied voltage in re v e rse d irec tion

Vd = diffusion potential (about half a volt in silicon diodes)

n = 2 for step p - n junction

= 3 for graded p - n junction

thus, the capacitance of the diode w ill be a function of the applied b ias (forw ard or rev erse ).

R eferring to F igure 7, if E Sig# = 0 and the bridge is balanced, no m odulation signal w ill appear a t the output. When a sm all signal voltage is applied, the b ridge becom es unbalanced and a c a r r ie r frequency signal appears a t the output, with magnitude and phase determ ined by the magnitude and po larity of the input signal. The varia tion of diode capacitance for a 5 mV applied signal is approxim ate­ly 0. 2% and so, for accura te reading of m illivolt signal levels, a ll components in the bridge would have to be stab le to about 0. 002%. Schwartz*2 rep o rts that AVd =- 1 m V /°C and = 650 p p m /°C and a lso that 5/li V /°C d rif t figures have been

achieved in a laborato ry environm ent with a 1 Mß source impedance.

12 AMPLITUDE MODULATION WITH TRANSFORMER COUPLING

^coupling "^conjugate

— j

\ Output

” W W

\ r AAl / C a r r ie r R \ /-\ *^R frequency

v\ w *~7/~ drive \ / _______________

Figure 7. VARIABLE CAPACITANCE DIODE BRIDGE .

4Two com m ercial am plifiers based on this bridge, the Philb rick P2A

and AEG rep o rt d rifts of 50 p V /°C and 30 p V / ° C respectively . Philbrick rep o rt a stab ility of - 100 p V p e r day.

Pow er gain is achieved during the modulation but null-balance feed­back is generally requ ired to lin ea rize the tra n s fe r ch arac te ris tic . This m odulator is suitable for d irec t application to the tran sd u ce r outputs if the stab ility is accep t­able. The c a r r ie r frequency can be made of the o rd er of MHz and a frequency response of tens of kHz is read ily achieved. The input re s is tan ce is very high but the input capacitance can be sev e ra l hundred pF.

3. OTHER FORMS OF MODULATION WITH TRANSFORMER ISOLATION

An alte rnative to varia tion of the amplitude of the c a r r ie r w aveform is varia tion of its frequency. This method of modulation is inherently less sensitive to noise added to the modulated c a r r ie r since such noise changes the amplitude of th is w aveform but has little effect on its frequency (or ra te of zero crossings) until the re la tive magnitude of the noise is large. For both a sinusoidal and a square-w ave c a r r ie r , the modulation and demodulation c ircu its a re usually m ore complex than those requ ired for am plitude modulation. The inherent insensitiv ity to changes of gain in the coupling c ircu its and to additive noise may offset the increased complexity. The useful band-width with frequency modulation is norm ally le ss than one-tenth of the c a r r ie r frequency.

The frequency m odulated c a r r ie r can be iso lated by passing i t through a tra n sfo rm e r in the m anner shown in F igure 1. By using a Schm itt tr ig g e r to detect the z e ro -c ro ss in g of the m odulated c a r r ie r and averaging the output of a m onostable m ultiv ib rato r which fire s at every +ve going (say) zero crossing , the demodulation

OTHER FORMS OF MODULATION WITH TRANSFORMER ISOLATION 13

tra n s fe r c h a rac te ris tic (frequency deviation v ersus output voltage) can be made quite linear. If the modulation tra n s fe r c h a rac te ris tic (input voltage versus frequency deviation) can a lso be made linear, it may not be n ecessa ry to apply negative feed­back, thus sim plifying the c ircu its .

The CMRR achieved with a given tra n s fo rm e r w ill be b e tte r for a f r e ­quency m odulated c a r r ie r than for an am plitude m odulated c a r r ie r . This is because the s tra y signal coupled into the tra n sfo rm e r secondary by the common-mode voltage (between the p rim ary and ground), has little effect on the frequency of the modulated c a r r ie r .

The rem ark s made below with re sp ec t to a frequency modulated square wave apply in general to pulse width, pulse frequency, pulse position etc. modulation.

3 .1 Pulse o r Square Wave C a rrie r

Ref. 37

With pulsed c a r r ie r s , such form s of modulation as pulse code m odula­tion (PCM), pulse position modulation (PPM) and pulse frequency modulation (PFM) can be employed. B ecause of the complex c irc u its requ ired for coding and decoding PCM, it w ill not be considered fu rther.

A possib le approach is to frequency modulate a tra in of pulses o r square waves. K. M uirhead of the D epartm ent of Engineering P hysics uses such a technique in h is handling of se ism ic data. A c ircu it which can be used in the isolation of tran sd u ce r signal voltages from th e ir associated com m on-m ode voltages is shown in F igure 8. The operation of th is c ircu it is as follows: The am plified tran sd u ce rsignal voltage is converted to a p roportional constant cu rren t which is used to charge the frequency determ ining capacito r in an astab le m ultiv ib rato r. The deviation in the m u ltiv ib ra to r’s repetition frequency is a lin ea r function of the deviation in charg ­ing c u rre n t and thus of the am plified signal voltage. K. M uirhead has verified that th is tra n s fe r function (between signal voltage and frequency deviation) is both stab le and lin ea r and that even fo r an accuracy of 0. 5% of full sca le , feedback is not necessary . The modulated signal is coupled to ground via an iso lating tra n sfo rm e r in much the sam e m anner as with am plitude m odulated signals, an im portant d iffe r­ence being that frequency modulated signals a re le ss affected by im pressed noise and induced com m on-m ode voltages, the design of the isolating tra n sfo rm e r thus being sim plified. The signal from the secondary of the tran sfo rm er operates a Schmitt tr ig g e r which d rives a m onostable m ultiv ib rato r. The output pulse -width of the m onostable is chosen to give a unity m ark -sp ace ra tio when the input signal to the modulator is zero . The output stage develops a voltage p roportional in magnitude and sign to the deviation of the m ark -sp ace ra tio from unity and thus proportional to the frequency deviation.

Without excessive cost o r c ircu it com plexity, th is output voltage can

14 OTHER FORMS OF MODULATION WITH TRANSFORMER ISOLATION

Functional Arrangement

V+ (from PS)• - - - - T- - - - - - - - - - -I

> c< *>

1 . _ _ Y

V- r 1r

........i

/ - C)/ / ' _ _ __ ------------ —

iFrequency modulated square wave to isolation transform er prim ary

1 n_____ !

I = a + b x E , c siga, b = constants

Astable MultivibratorModulator

Output

AdderF. M. Signal from isolation transform er secondary R„ = R,

Demodulator

Figure 8. FREQUENCY MODULATED ISOLATION AMPLIFIER

OTHER FORMS OF MODULATION WITH TRANSFORMER ISOLATION 15

be m ade to rep re sen t the signal voltage to w ithin b e tte r than 1% of full scale . The attainable frequency response is determ ined by the m odulator’s m ean frequency and no difficulty should be experienced in obtaining a useful bandwidth of sev e ra l kHz.The CMRR w ill be adequate if ca re is taken with the shielding of the floating am plifier and input leads and balancing of com m on-m ode input im pedances. The input re fe rre d d rift depends mainly on the fj. A709C am plifie r and as is shown in Appendix 10. 2, this can be reduced to a reasonable figure.

3. 2 Sinusoidal C a rr ie r

Frequency modulation of a sinusoidal c a r r ie r is norm ally achieved by varia tion of a reactive elem ent in the frequency determ ining network of an oscilla to r. Phase modulation may be thought of as m odulation of frequency by the derivative of the modulating waveform. In the case of a sim ple L - C tuned c ircu it, the change in frequency is inversely proportional to the square root of the change in e ither of the elem ents and thus is reasonably linear only for sm all changes in the elem ents. A fu rth er difficulty is that the tra n s fe r c h a rac te ris tic of e lec tron ica lly variab le r e ­active elem ents, such as a reactance tube o r voltage variab le capacito r, may be both non-linear and subject to changes in c h a ra c te r is tic s . For the above reasons it is often n ecessa ry to apply overa ll negative feedback. The demodulation of the m odulat­ed c a r r ie r can be achieved in the sam e m anner as was outlined in section 3 .1 above, and th is p rocess w ill be both stable and lin ea r. In the case of phase modulation, it may be n ecessa ry to m aintain a re fe ren ce signal and unless th is re fe ren ce is av a il­able to both the m odulator and dem odulator, troub le with d rif t may be experienced.

Because of the need fo r overa ll negative feedback if reasonable accuracy is demanded, it is considered tha t a sinusoidal c a r r ie r is in fe rio r to a square-w ave or pulse c a r r ie r in the p re sen t application and it w ill not be examined further.

4. ISOLATION WITH A DIFFERENTIAL AMPLIFIER

Refs. 2 ,3 ,1 0 ,3 8 ,3 9 ,4 0 ,4 1 ,4 2 (See F igure 9)

E lectronic c irc u it techniques which amplify d ifferen tia l signals with inherently high re jection ra tio s for com m on-m ode signals, a re available. Rejection of common-mode voltages in favour of d ifferen tia l voltages can exceed 120 dB at very low frequencies, but th is ra tio usually falls off fa irly quickly for high source im ped­ances and as frequencies in c rease . Two possib le approaches to re jec tion of the common-mode a re to apply the tra n sd u c e r’s output to a d ifferen tia l am plifier - (a) with and (bj without p r io r am plification by a floating am plifier. A ttenuation w ill be requ ired to bring the com m on-m ode voltage into the allowable range for the d iffe r­ential am plifier, because the com m on-m ode voltage is assum ed to be too la rg e for d irec t application.

16 ISOLATION WITH A DIFFERENTIAL AMPLIFIER

Jsig A

'em A

tran sd u ce r------W - -------

P rec is io nA ttenuatorP a ir

j

1

I-AOutput

S

r< < , D ifferential^ A m plifier

C ircuit for High Common-mode Voltages

« Output f

C ircuit for Low Common-mode Voltages

F igure 9. COMMON-MODE ISOLATION WITH A D IFFER- ENTIAL AMPLIFIER.

D ifferentialEcm A A m plifier

4 .1 Common-Mode R ejection Without P r io r A m plification

(See F igure 9)

If we consider a la rge com m on-m ode voltage (say 1, 000 V) and a sm all signal (say 5 mV) i t may be seen that a CMRR of 2 x 107 : 1 w ill be required for 1% e r ro r . This figure is not read ily achieved by conventional d ifferential input c ircu its . A part from th is objection, if only one side of the tran sd u cer is connected to the common-mode voltage, any output re s is ta n c e in the tran sd u cer adds to one a rm of the attenuator p a ir and so causes uneven division of the common-mode voltage. This w ill cause the e r ro r in division of the com m on-m ode voltage to appear as a d ifferen tia l input to the d ifferen tia l am plifier and it w ill be am plified. F or a 5 mV signal on 1, 000 volts of common mode, the division ra tio of the two attenuators will

ISOLATION WITH A DIFFERENTIAL AMPLIFIER 17

have to agree to w ithin 0. 000005% (not a p rac tica l proposition), if the e r ro r due to d ifferen tia l am plification of common-mode is to be le ss than 1%. For lower com m on­mode voltages and h igher, low impedance, signal levels this c ircu it could becom e p rac ticab le . The frequency response w ill be good and the input impedance w ill depend on the a ttenuator. The d rift s tab ility w ill depend on the division ra tio and stab ility of the a ttenuato r and on the d rift c h a rac te ris tic s of the operational am plifier. F or large attenuator division ra tio s the d ifferen tia l input signal w ill be reduced and the effective am plifie r d rif t (re fe rred to the tran sd u ce r output) w ill be equal to the am plifier d rift m ultiplied by the a ttenuato r ra tio .

4. 2 Common Mode R ejection with P r io r A m plification

(See F igure 10)

As mentioned above, a signal of 5 mV and a com m on-m ode voltage of 1, 000 V req u ire s an agreem ent between the two a ttenuator ra tio s of 0. 000005%. A p rac ticab le figure is . 01% 43 . This im plies tha t the signal voltage has to be am pli­fied by floating am p lifie rs to a level of - 5 volts, i. e. 10 volts d ifferential. The use of an inphase and antiphase signal voltage from equal (low impedance) am plifier out­puts at the top of the a ttenuato rs reduces the problem of inaccura te division of the com m on-m ode voltage caused by unequal com m on-m ode source im pedances. With the h igher d ifferen tia l signal voltage p resen ted to the d ifferen tia l am plifier, a CMRR of 80dB is requ ired for 1% e r ro r and th is is read ily achieved*. The c ircu it outlined w ill give quite sa tisfac to ry perform ance if a finite d. c. im pedance to ground (due to the a ttenuator pair) is acceptable for the tran sd u ce r being considered. The c ircu it uses four am p lifie rs , th ree of these a re floating (see Appendix 10.1) and one has a high CMRR and (preferably) a la rge com m on-m ode range. Fortunately, the F airch ild ^A709C operational amplifier^® is sa tisfac to ry in a ll four positions - these am plifiers a re available o ff-the-she lf a t a reasonable p rice , so that the overa ll cost need not be excessive.

5. ISOLATION BY OPTICAL COUPLING

Light sources and de tec to rs a re generally not linear in operation and fo r th is reaso n the c ircu its to be considered below w ill make use of the light coupling fo r conveying frequency and not am plitude inform ation or as an on-off indicator fo r pa ra lle l d ig ital inform ation.

5 .1 Photon-Coupled Isolation With a Frequency Modulated C a rrie r

Light-coupled iso la to rs with excellent e lec tro s ta tic shielding, wide band-width and high breakdown voltages a re now available. ^ At p resen t these devices a re ra th e r expensive but p rice s can be expected to fall as they becom e m ore common. There is a fa irly la rge loss of signal during isolation but with frequency modulation this loss can easily be made up by am plification.

18 ISOLATION BY OPTICAL COUPLING

Floating A m plifier

P rec is io nA ttenuatorP a ir

D ifferentialA m plifier

Output

C ircuit fo r high level tran sd u ce r outputs

' s i g 0tEcm 0

^ +A > -...f ..-1 V

FloatingA m plifiers

\ s

-A ~1

P rec is io nA ttenuatorP a ir

i? D ifferen tia l \ A m plifier-A 0>----- •

»Output

aC ircu it fo r low level tran sd u ce r outputs.

F igure 10. ISOLATION BY A DIFFERENTIAL AM PLIFIER AFTER PRIOR AMPLIFICATION.

ISOLATION BY OPTICAL COUPLING 19

An obvious application is to replace the isolation transform er of Figure 1 by such an isolator and a signal-level restoring amplifier. The convention­al isolation transform er is likely to give quite adequate performance in this situation, however, and it is unlikely that the additional cost of the photon-coupled isolator and amplifier could be justified in normal applications.

5. 2 Digital Light Coupling

Light coupling AnalogueOutput

8 light 8 light detectors

8 bit (. 4%) Analogue to digital converter

Buffer Amplifiers and D-A converter

Figure 11. ISOLATION OF COMMON-MODE BY DIGITAL LIGHT COUPLING.

Referring to Figure 11, the transducer output can be amplified and sampled at short intervals by a floating analogue to digital (A - D) converter whose parallel binary outputs can be coupled to a digital to analogue (D - A) converter at ground potential via neon-lamp-photo-diode pairs or the photon coupled isolators mentioned in Section 5.1. It would of course be possible to use a serial rather than parallel transmission of information, thus reducing the number of optical circuits to one. The D-A converter can reconvert the sampled values in sympathy with the sampling times and so yield an approximation to the transducer outputs. A-D and D-A converters can function at tens of thousands of sam ples/sec. and the frequency response could extend to several kHz. The input impedance and drift stability depend mainly on the amplifier preceding the A-D converter. The CMRR will be determined primarily by stray currents in the floating circuits prior to digitization. The light coupling itself can have virtually infinite rejection.

Unfortunately, the above system is complex and will probably be too expensive to be considered except for very high common-mode voltages such as are associated with power transm ission lines.

£RS P H Y S .S > )

L i b r a r y

20 ISOLATION BY SAMPLING AND STORING

6. ISOLATION BY SAMPLING AND STORING

If the tran sd u ce r output is am plified, sampled, and stored in a floating digital o r analogue sto re , the s to re can be disconnected and retu rned to ground poten­tia l whereupon it can be read out. An analogue sto re could take the form of a sam ple and hold operational am plifier c irc u it which would have to be connected to the tr a n s ­ducer and back to ground by m echanical sw itches because of the high common-mode voltage. The m echanical sw itches would lim it the speed of operation to a few tens of s a m p le s /s e c .

A digital sam pler and s to re would typically consist of an A-D convert­e r feeding m agnetic co res. These co res need not be re tu rned to ground potential to be read because the ’’d rive" and "sense" lines can be insulated from the floating c ircu its . The sam pling speed in th is case could be considerably g rea te r - up to sev era l tens of thousands of sa m p le s /se c .

The input im pedance, d rift s tab ility and common-mode rejection will m ainly depend on the floating p re -am p lifie r.

7. ISOLATION BY COUPLED FLUX

In con trast to the use of the ra te of change of flux in a tran sfo rm er core as the method of isolation, it is possib le to use the magnitude and direction of the flux itse lf. In p rac tice , it is somewhat m ore difficult to m easure the magnitude of the flux than it is to m easure its ra te of change, and the non-linear nature of the B - H c h a rac te ris tic s of the m agnetic core m eans that some form of null feedback is requ ired . In th is case, however, the provision of such feedback is quite sim ple in as much as a ll that is requ ired is an additional winding on the m agnetic core whose m. m. f. is equal to and opposes the m. m. f. of the tran sd u ce r winding. Two methods of using coupled flux as a m eans of iso lation a re d iscussed below.

7 .1 P u lse Relaxation A m plifier

The c ircu it of F igure 12(i) u ses a magnetic co re with a B - H ch arac t­e r is tic s im ila r to that depicted in F igure 12 (ii). Taking the sim ple case of no applied a. c. b ias , a tra in of alternating pu lses is applied to the secondary c ircu it. The m agnitude and duration of these pulses is such tha t if the co re was satu rated in the -B d irec tio n p r io r to the application of a pulse which tends to d rive the flux in the +B direction , then the co re w ill ju s t be sa tu ra ted in the +B d irection at the end of this pulse. D uring a pulse, until the co re is sa tu rated , little cu rren t flows in the load. R eferring to F igure 12(ii), if the flux p r io r to the application of such a pulse was not at (e) (where the previous pulse would have left it) but due to the action of a control winding m. m. f. was a t some other point, say point (c), the core would satu ra te e a r lie r in the pulse and the pulse voltage would appear ac ro ss the load ra th e r than

ISOLATION B" COUPLED FLUX 21Pulse Supply

Low-pass filterOutput

A. C. bias____ >

Feedback

Figure 12 (i)

+B

-B . !sat

Figure 12 (ii)

average output current l d *

control m. m.

without a. c. bias

with a. c. bias

a. c. bias

Figure 12 (iii)FIGURE 12. PULSE-RELAXATION AMPLIFIER

22 ISOLATION BY COUPLED FLUX

the secondary winding for a greater proportion of the pulse period. If the control m. m. f. remained constant, the next pulse (in the opposite direction) would have to drive the flux from point (d) to saturation in the -B direction, that is, over the full range of flux density and as a result the current flowing in the load would be sm aller than that which flowed during the previous pulse, thus giving rise to an average com­ponent of current in the load which varies with the control winding m. m. f. If the control winding m. m. f. is small and lies in the range bb’ it will not affect the average output current, as shown in Figure 12(iii). The addition of an a. c. bias (synchronized with the pulse supply so that its peak and thepube coincide) with a peak to peak amplitude of aaf overcomes the dead zone and produces a transfer character­istic sim ilar to that shown in Figure 12 (iii). It is doubtful if this characteristic would be sufficiently stable to avoid the use of feedback. The provision of negative feedback is a simple matter because it can be applied via a feedback winding which can be at ground potential. Morgan and McFerran^5 claim exceptionally good drift stability for this type of amplifier giving a figure of 10“^ watts from -70°C to 140°C with a 50ß input resistance contrasted with about 10“® watt for a conventional mag­netic amplifier.

Inasmuch as the control m. m. f. depends on the current flowing in the control winding (which will have a relatively long time constant L/R), the frequency response of such an amplifier will be limited by the changing input impedance and frequency of the pulsed supply to some tens of Hz.

This circuit will be useful for transducers whose output impedance and voltage is low and whose frequency spectrum is limited, e. g. thermocouples.

Morgan and McFerran give a circuit for a two core amplifier which reduces the loading on the input circuit and prevents the pulses being fed back to the transducer^5 .

7. 2 Second Harmonic Type Magnetic Modulator

Refs. 11,46,47 (See Figure 13)

The B - H loop of a magnetic m aterial is usually symmetrical with respect to the origin B = 0, H = 0 and although an a. c. voltage applied through a resisto r to a coil surrounding a core of such material may be distorted by the B - H characteristic, this distortion normally contains only a small percentage of even harmonics as the positive and negative waveforms are sim ilar. If the resultant a. c. magnetisation is displaced with respect to the origin by a d. c. bias, the distortion is not symmetrical, and even harmonics are present - the phase and magnitude of the even harmonics depend on the polarity and magnitude of the d. c. bias. These even harmonic components (typicaHy the 2nd) may be detected and used as an indication of the d. c. bias. As the relationship between the magnitudes is non-linear, feedback to give a null balance is required. Two matched cores are used and opposed to prevent the fundamental voltage from being fed back to the signal source. For high

ISOLATION BY COUPLED FLUX 23

Low Second Harmonic Content Oscillator, frequency f.

FrequencyDoubler

Tuned to 2f

Outputf + 2f

Feedback

T , T - matched pair

Figure 13. SECOND HARMONIC TYPE MAGNETIC MODULATOR

sensitivity, a very large number of turns on the signal winding is needed and, as a result, the frequency response is limited to only a few Hz by the large inductance present in the signal path. Schwartz quotes sensitivity as being 10"® amps with lkß input re s is tan ce^ . This type of circuit is satisfactory for transducers with a low impedance and predominantly d. c. output spectrum. A stability of 3 x 1 0 watts drift over several hours is given by Noble and Baxandall^.

8. OTHER METHODS OF ISOLATION

8.1 "Overlapped CircuitM.

The "overlapped” circuit is, in effect, two half-wave chopper- stabilized amplifiers connected in parallel, being so arranged (by overlapping) that no information present in the input signal is lost (see Figure 14). The demodulation technique shown maintains the d. c. level of the output signal by setting the output to zero when it is known that the input for that path is zero, i. e . , when it has been chopped. This operation will be referred to as "resetting". During the interval be­tween successive resettings, the output voltage will "droop" slightly because of the limited low frequency response of amplifiers A1 and A2 and the coupling circuits. This response must be sufficient to limit the droop to an acceptable figure (if necessary, feedback will of course improve the droop).

By using a m ark-space ratio somewhat greater than unity in each of the overlapped signal circuits, the input signal wiH always be present in an amplified form from one or other of the output circuits, and by using diodes (if the signal

24 OTHER METHODS OF ISOLATION

II a >

"1-4

4

Output

1J Diodes may replace switches

if input polarity is fixed___——- Input signal

Ecm

Output signal at point a

Output signal at point b

Combined output at point c

Figure 14. "OVERLAPPED" CIRCUIT AND TYPICAL WAVEFORMS

polarity is known beforehand) or switching (with overlap) one output and then the other to the output terminal* the upper frequency response of the complete circuit will be limited only by the amplifiers A1 and a 2 and the coupling circuits. As the speed of the operation of the modulating device is not a limitation to the upper frequency

OTHER METHODS OF ISOLATION 25

response, photo-resistors, which otherwise are very good modulating elements, can be used as the input choppers and a chopper stabilized amplifier of good performance can be constructed.

Conventional solid state switches (such as F. E. T. ’s or dual-emitter transistors) can be used in the output "resetting” and combining operations. Overall feedback could be used to improve the performance, if desired.

To modify the overlapped circuit so that it is suitable for use as an isolation amplifier, isolation may be achieved by the insertion of capacitors or wide - band transform ers after the first set of switch contacts; see Figure 14.

The capacitors and transform ers reject the d. c. component of the common-mode voltage, but the capacitors will couple a. c. components to ground and to offset this, a third capacitor C3 is connected to the positive input of the differential amplifiers.

The variations in the common-mode must not be larger than the common-mode range of the differential amplifiers or must be so slow that the capac­itors attenuate these variations to a reasonable level.

The general performance will be sim ilar to the amplifiers of section 2.1 but the band-width will be greater.

8. 2 Isolation by Electromagnetic Coupling

It would be reasonable to expect this method of isolation to be the least sensitive of aH methods to the common-mode voltage. Electromagnetic coupling includes optical coupling which has already been discussed. A technique which could be used is shown in Figure 15. A method of modulation and demodulation not sensitive to changes in carrier amplitude should be chosen. The use of a transform er coupled floating power supply such as that shown in Appendix 10.1 would degrade the CMRR of the circuit of Figure 15, and an independent power supply e. g. a battery, would be necessary if the potentially very large CMRR of this circuit is to be realized.

It is not expected that it will often be necessary to go to the lengths described above to obtain satisfactory isolation and this type of isolation will not be further discussed.

8. 3 Acoustical Coupling

Acoustical coupling could in some circumstanaes be a satisfactory method of isolation. Pulse width or pulse frequency modulation would overcome variations in the gain from the acoustical driver to the acoustical receiver. A large power loss is associated with acoustical coupling and to reduce this loss and avoid cross talk with other amplifiers and the environment, it may be necessary to use

26 OTHER METHODS OF ISOLATION

Ec m OV '

Figure 15. ISOLATION BY ELECTROMAGNETIC COUPLING

’’sound pipes" to connect the d riv e r and re ce iv e r. The frequency response a tta in ­able w ill be lim ited by the availability of suitable d r iv e rs and rece iv e rs and the increased attenuation a t high acoustical frequencies.

8. 4 M echanical Coupling

It is possib le to achieve e le c tr ic a l iso lation by making the ro tational or linear position (or velocity) of an insu lating rod depend on the output of the t r a n s ­ducer. The position o r velocity of the rod would be converted into e lec trica l te rm s by a potentiom eter o r coil in a m agnetic field. T here a re few possib le applications w here th is method would be p rac ticab le and i t w ill not be exam ined further.

9. CONCLUSIONS

P ossib le c ircu its for iso la tion am plifie rs have been examined. I t is not possib le to say tha t any one c irc u it is the b e s t fo r aH applications but it appears that the c ircu it utilizing frequency m odulated square waves examined in section 3.1 w ill be suitable in many cases.

In the iso lation of low -level signals having a re s tr ic te d frequency spectrum , the chopper-type am p lifie rs a re useful. The v ib ra ting -capacito r type of am plifier appears to have the b e st perfo rm ance if sm all cu rren ts from high im ped­ances a re to be m easured.

If the signal levels a re high, the tran sd u ce r output (including the com m on-m ode voltage) can be attenuated by a p a ir of voltage d iv iders and applied to a conventional d ifferen tia l am plifier.

CONCLUSIONS 27

The variab le-capacitance diode bridge type of m odulator req u ires feedback but gives a good overa ll perfo rm ance for m edium signal levels and w ill be useful when a wide bandwidth is requ ired .

A common problem with iso lation am plifiers requ iring feedback to linearize and im prove th e ir perform ance, is tha t in applying a d. c. feedback voltage to the input, the feedback voltage m ust be iso la ted from the grounded output am plifier. This often im plies tra n s fo rm e r coupling and the overa ll linearity of the am plifier w ill then be determ ined by the linearity of the m odulator, tra n s fo rm e r and dem odulator in the feedback c ircu it. For th is reason , c ircu its which do not req u ire feedback a re to be p re fe rred .

A wide range of c ircu its has been presen ted and from th is it should be possib le to choose an isolation c ircu it to su it m ost applications.

10. APPENDICES

10.1 Floating A m plifier (See F igure 16).

The m odulation by the low -level tran sd u ce r outputs of a c a r r ie r wave­form is sim plified if the outputs a re f ir s t am plified. This am plification can be achieved by an am plifie r which is iso la ted from ground and has its own floating power supply. A m plifiers of the F airch ild gA709C type consume a typical 80 mW from - 15 volt supplies. T heir low pow er requ irem en t w ill be increased , by the need to have zener diode stab ilized supplies, to about 300 mW. This power can be easily supplied by a tra n s fo rm e r wound on a f e r r i te toroid , operating from a square-w ave drive.

F airch ild 2N3638A tra n s is to rs make good, cheap zener diodes with about 6. 8 volts a t 20 mA. This is achieved by operating the tra n s is to rs with r e v e r s ­ed polarity , the b a se -e m itte r junction being re v e rse b iased and acting as a zener, while the collector«4)ase junction is forw ard b iased and tem p era tu re com pensates the zener. Four of th ese tra n s is to rs a re used in each of the floating power supplies.

The F airch ild piATOSC is not usually suitable as an operational am pli­fie r when operating with an input re s is ta n c e of g rea te r than 20 kß o r so. Using the methods outlined in Appendix 10. 2, the /iA709C can be made to operate sa tisfac to rily with a 100 kß input impedance. The am plifier is quite sensitive to changes in supply voltage and for th is reason, the zen er diode stab ilized supply is required . The floating am plifier and power supply should be enclosed in and connected to a shield which is connected to the com m on-m ode potential, to prevent capacitive and re s is tiv e common-mode c u rre n ts flowing in the signal leads.

28 APPENDICES

~ i

2 x 2N3638A

2 x 2N3638A

+ 14 volts common- 14 volts

Shield

15 kHz square ---- wave

Fairchild /iA709C

- AEsig

Figure 16. FLOATING AMPLIFIER

10.2 Temperature-Dependent Drift Reduction in the Fairchild uA709C Operational Amplifier.

Refs. 38, 39 (See Figure 17 )2

41 42 There are four factors contributing to tem perature dependent drift ’

(i) The temperature coefficient of the em itter-base voltage of Q1 and Q2 is about -2. 5 mV/°C. By close electrical and therm al matching of the transistors, the temperature coefficient of the difference between their em itter-base voltages (typically 1. 0 mV) is held to a level of about - 3 \iV /°C .

(ii ) The forward common-emitter current gains of Q1 and Q2 are dependent on both the temperature and collector currents and will normally be matched to within a few percent. The tem perature sensitivity of current gain is approximately 1% per degree centigrade and the

APPENDICES 29

V+

Differential output to next stage

* I = constant (deliberately varied with temperature)

FAIRCHILD |iA709C INPUT STAGE

gain increases by about 50% for each decade of increase in collector current (at microamp current levels).

(iii) The collector-base leakage currents in Q1 and Q2 (caused by intrinsicand minority carriers) are not usually matched to better than about 20%. These leakage currents approximately double for every 8 C temperature rise and accordingly reduce the bias currents required, but not usually by an equal amount.

(iv) In the fiA709C, the emitter current of the differential input pair isdeliberately varied with tem perature so that the t r ans-conductance( = ß ) and therefore the voltage gain, remains substantially

R.inconstant. If the current gains of Q1 and Q2 are not perfectly matched, this variation in emitter current will cause a variation in the difference between the bias currents of Q1 and Q2.

The first factor produces a drift in the input "offset voltage" i. e. the voltage which must be applied between the input terminals to obtain zero output voltage. The other three factors combine to cause a drift in the input "offset current", i. e . , the difference in the currents into the two input term inals with zero output voltage.

When R^ is less than lOkfi or so, the drift in offset voltage is the predominant factor in overall drift whereas for R^ larger than 10 kß the drift due to

30 APPENDICES

offset cu rren t becom es dominant. The norm al recom m endation is that the total value, R of the equalising re s is to r plus the equivalent res is tan ce of the offset voltage adjustm ent network, should be equal to . Adopting this recom m endation, the equivalent input d rift, with R^ = 100 kfl is typically 50 \xW/°C, i. e. , an o rder higher than the d rift due to the em itte r-b ase d ifferential. Fortunately, the ra tio of the input cu rren ts into Q1 and 02 tends to rem ain fa irly constant and by choosing the ra tio of R^ to R equal to this ra tio , the change in the equivalent voltage offsets can be reduced (see Figure 18). In a typical case, the ra tio of input cu rren ts was 1. 21at 0°C, 1. 22 at 10°C and 1. 25 at 20°, 30°, 40°, 50° and 60°C. By choosing the

as 1. 23, for th is case it was possib le to balance the equivalentra tio of Rn to R_1 eq

Esig

forw ard. - v \a ~

O

feedback------W —|llA709C

k 'Offsetcu rren tadjustm ent

Offset voltage adjustm ent

out

^ fo rw ard X ^feedback

^fo rw ard + ^feedback

Figure 18. METHOD OF COMPENSATION

drift in the input offset voltage caused by the offset cu rren t to within 5fiV/°C over the range from 0°C to 60°C. By a suitable com bination of offset voltage adjustm ent and offset cu rren t adjustm ent, the d rift due to the change in offset cu rren t can be made to oppose the d rift due to the change in offset voltage and so give an even lower overall drift. This technique is p a rticu la rly applicable to the case where operation over only a sm all tem pera tu re range (say - 8°C) is d esired and in such a situa tion an equivalent input d rift of 1(liV /°C average has been achieved with R forw ard =100 kfl and R feedback = 1 Mft. In th is p a rticu la r case, slow random fluctuations of the output, equivalent to about - 2 0 jlV peak at the input w ere observed. For the impedance levels and tem p era tu re range quoted, the tem p era tu re dependent d rift had been reduced to the point w here it was of secondary im portance.

By using the method outlined, the /xA709C may be used in high im ped­ance c ircu its which would norm ally preclude its use. Balancing the d rift caused by offset cu rren t against that caused by offset voltage reduces the overall d rif t in low

APPENDICES 31

impedance c ircu its .

It has been shown that operation of the input d ifferentia l p a ir at equal e m itte r-b a se voltage (and perhaps with unequal co llecto r cu rren ts), reduces the d rift of the em itte r-b a se d ifferentia l with tem peratu re . With the /xA.709C, this technique may be useful when the m ain component of d rift is the change in the em itte r- base d ifferential. Hoffait and Thornton rep o rt at least a five-fold reduction in the tem p era tu re coefficient of d rift^2 . The use of th is method m eans that th e re will be an offset voltage in the output cf the /liA709C, approxim ately equal to the gain as a feedback am plifier m ultiplied by the input offset voltage. This can be allowed for in the next am plifier stage.

10. 3 Iso lated D rive C ircuit for M etal-Oxide Sem iconductor T ran s is to rs (MOST’S).

Many applications exist, such as m ultip lexers, w here MOST’s can be used as s e r ie s sw itches. To reduce the settling tim e requ ired for the tu rn -on and tu rn -o ff tran s ien t, a switching voltage waveform with a fast r is e tim e (say < l^iS) is required . The switch may have to rem ain "on” for long periods, while the input which has been connected is m onitored. The switching w aveform w ill generally have to be iso lated from ground and tra n sfo rm e r coupling can be a suitable solution, apart from the fact th a t it is difficult to design tra n s fo rm e rs with a sufficiently wide f r e ­quency response. A half- o r full -wave re c tif ie r driven from a fa irly sim ple t r a n s ­fo rm er can be used, but the tu rn -o ff tim e w ill be slow com pared to the tu rn-on tim e unless a d ischarging c ircu it is incorporated .

The c ircu it shown in F igure 19 sim plifies the design of the tran sfo rm er.

D rain

Source

Figure 19. ISOLATED DRIVE CIRCUIT

32 APPENDICES

All that is requ ired is that a positive going, fast r is e - tim e pulse of sufficient am pli­tude and duration to charge capacito r C (through Diode D^ ) to the requ ired tu rn-on voltage, be produced at the tra n s fo rm e r secondary. A negative-going pulse d is ­charges the capacitor through tra n s is to r Q operating in the com m on-base mode. The gate leakage cu rren t for the MOST is very sm all and may be neglected. The charge on the capacitor leaks away mainly as a re su lt of the diode D^’s rev e rse leakage cu rre n t and the tra n s is to r ’s co llec to r-b ase leakage cu rren t. These leakage cu rren ts tend to rem ain constant with applied voltage and so the capacito r d ischarges linearly . Taking typical values of C = 1, 000 pF, I re v e rse = InA» Ick 0 = 2nA,

dV I 3 x 10~9----- = — = ----------- = 3 volts p e r seconddt C 10“9

By overdriving the MOST during tu rn-on , the voltage can easily be m aintained at a sufficient level for a second o r m ore. Diode D2 and re s is to r R. may be necessa ry to prevent re v e rse breakdown of the e m itte r-b a se junction in Q. R esis to r may be used to reduce the peak cu rren t through diode D2 and into the em itte r of Q.

The isolating tra n sfo rm e r need be no m ore com plicated than a fe rr i te toroid having a few dozen p rim ary and secondary tu rns. The secondary r is e tim e is lim ited by the product of the leakage inductance and the capacitance being discharged.

By driving tra n s fo rm e r T with either positive pulses or a triangu lar w aveform having a fest positive-going r is e - tim e and a slow fa ll-tim e (so that t r a n ­s is to r Q does not conduct), the MOST may be kept turned on indefinitely. Fortunately, the tran s ien t induced in the signal c ircu it each tim e the voltage is re s to red is very sm all. This is because the step change of voltage a c ro ss Cgd is sm all and it charges through the d ra in -so u rce re s is tan ce of the F. E. T. which, being turned "on" is low. As an example, if Cgd = 2 pf and the gate voltage is re s to red in 1/lxS a fte r it hasfallen 1 volt, with R = 1, 000 ßon

—12 6 “"6I tran s ien t = C dV = 2 x 1 0 x l O = 2 x 1 0 Amps,

dt

V tran s ien t = I x R = 2mV for I jlxS in the signal c ircu it. This tra n s ien t can be reduced even furfiier by re s to rin g the gate voltage m ore often, e. g. , a fte r it has fallen 0 .1 volts.

REFERENCES 33

11. REFERENCES

1. MEYER-BROTZ, G and A. KLEY: "The Design of D ifferential D. C.A m plifiers with High Common-Mode Rejection. " E lectronic E ngineer­ing, February , 1966, pp 77-81.

2. MIDDLEBROOK, R. D. : D ifferential A m plifiers (John Wiley and Sons, Inc.New York, 1963.

3. SLAUGHTER, D. W. : "The Em itter-C oupled D ifferential A m plifier",T ran s . IRE, M arch 1956. pp. 51-53.

4. PHILBRICK RESEARCHES IN C .: "SP2A O perational A m plifier", DataSheet.

5. CLEVITE CORPORATION: "B rush High Voltage D. C. A m plifier13-421580", Data Sheet 4215-80A.

6. DYNAMICS INSTRUMENT CORPORATION: "No. 6050 Wideband D ifferent­ia l D. C. A m plifier", E lec tro n ics , A pril 27, 1962, p. 104.

7. BUCKER FIELD, P. S. T .: "The ’P a ra lle l T ’ D. C. A m plifier", P roc. IEE,Pt. IH, 1952, p. 99.

8. MACLEOD, I. D. G. : "Frequency Response of C a rr ie r Type D. C.A m plifiers", Submitted for Publication.

9. AIRPAX INC. : "GP-1 M echanical Chopper", Data Sheet.

10. SCHAFER, C. R. : "D. C. A m plifier Using A ir Coupled Chopper".E lec tron ics, M arch 1950, pp. 104-105.

11. KANDIAH, K. and D. E. BROWN: "High Gain D. C. A m plifiers", P roc.IE E , Vol. 99, P t. 2, 1952, pp. 77-81.

12. WALSTON, J. A. and J. R. MILLER: Chopper A m plifiers, Chapter 10 in"T ran s is to r C ircuit D esign", (McGraw Hill, 1963).

13. SCHWARTZ, S. (Editor): D. C. A m plifiers: P a r t 2 in Selected Sem i-Conductor C ircu its Handbook (John Wiley and Sons, 1961).

14. ANONYMOUS: "Solid State Choppers for T elem etry A pplications",Product P ro file , Solid State Design, May 1965, pp. 17-18.

15. FAIRCHILD: "3N88 NPN High Speed Choppers", Data Sheet.

R I

P ub lica tio n s by D epartm en t of E ng ineering P h y sics

No. A uthor T itleF ir s t

Pub lished R e -issu e d

E P -R R 1 H ibbard, L. U. C em enting R o to rs fo r the C a n b e rra H om opolar G en era to r

May, 1959 A pril, 1967

E P -R R 2 C arden , P . O. L im ita tions of R ate of R ise of P u lse C u rre n t Im posed by Skin E ffect in R o to rs

Sept., 1962 A pril, 1967

E P -R R .3 M arshall, R. A. The D esign of B ru sh es fo r the C a n b e rra H om opolar G en era to r

Jan ., 1964 A pril, 1967

E P -R R 4 M arshall, R. A. The E lec tro ly tic V ariab le R esis ta n ce T e s t Load/Sw itch fo r the C an b e rra H om opolar G en era to r

May, 1964 A pril, 1967

E P -R R 5 Inall, E . K. The M ark II Coupling and R o to r C en tering R e g is te rs fo r the C a n b e rra Hom opo- l a r G en era to r

Oct. , 1964 A pril, 1967

E P -R R 6 Inall, E . K. A Review of the S pecifica­tions and D esign of the M ark II Oil L ubricated T h ru s t and C en tering B earin g s of the C an b e rra H om opolar G en era to r

N o v .,1964 A pril, 1967

E P -R R 7 Inall, E . K. P ro v in g T es ts on the C a n b e rra H om opolar G en­e r a to r w ith the Two R oto rs Connected in S e rie s

F e b . ,1966 A pril, 1967

E P -R R 8 B rady, T.W. N otes on Speed B alance C on tro ls on the C an b erra H om opolar G en era to r

M ar. ,1966 A pril, 1967

E P -R R 9 Inall, E . K. T e s ts on the C an b e rra H om opolar G en era to r A rran g ed to Supply the 5 M egaw att M agnet

May, 1966 A pril, 1967

Publications by D epartm ent of Engineering P hysics (C ont.) R2

No. AuthorF ir s t

T itle Published R e-issued

EP-RR 10 Brady, T.W . A Study of the P erfo rm ance June, 1966 A pril, 1967of the 1000 kW M otor Gen­e ra to r Set Supplying the C anberra Homopolar Gen­e ra to r Field

EP-RR 11 Macleod, I.D.G. Instrum entation and Control O ct., 1966 A pril, 1967of the C anberra Homopolar G enerator by On-Line Com­puter

EP-R R 12 Carden, P .O . Mechanical S tre sses in an J a n . , 1967Infinitely Long Homogeneous B itte r Solenoid with Finite External Field

EP-R R 13 Macleod, I.D.G. A Survey of Isolation A m pli- Feb. , 1967fie r C ircuits

EP-R R 14 Inall, E .K . The M ark III Coupling fo r Feb. , 1967the R otors of the C anberra Homopolar G enerator

EP-RR 15 Bydder, E. L.Liley, B .S.

On the Integration of M ar. ,1967"B oltzm ann-L ike”Collision In tegrals

EP-RR 16 Vance, C. F. Simple T h yristo r C ircu its Mar. ,1967to P u lse -F ire Ignitrons fo r Capacitor D ischarge

EP-RR 17 Bydder, E. L. On the Evaluation of E lastic Sept. ,1967and Inelastic Collision F r e ­quencies for H ydrogenic-Like P lasm as

EP-R R 18 Stebbens, A.Ward, H.

The Design of B rushes fo r M ar. ,1964 S e p t., 1967the Homopolar G enerato r at The A ustralian National University

Copies of this and other Publications (see list inside) of the Department of Engineering Physics may be obtained from:

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