A 10.2 Mbps Pulse Harmonic Modulation Based Transceiver for Implantable Medical Devices

1296 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 46, NO. 6, JUNE 2011

A 10.2 Mbps Pulse Harmonic Modulation BasedTransceiver for Implantable Medical Devices

Farzad Inanlou, Student Member, IEEE, Mehdi Kiani, Student Member, IEEE, andMaysam Ghovanloo, Senior Member, IEEE

Abstract—A low power wireless transceiver has been presentedfor near-field data transmission across inductive telemetry links,which operates based on pulse harmonic modulation (PHM).This PHM transceiver uses on-off keying (OOK) of a pattern ofpulses to suppress inter-symbol interference (ISI), and its charac-teristics are suitable for low-power high-bandwidth telemetry inimplantable neuroprosthetic devices. To transmit each bit across apair of high-Q LC-tank circuits, the PHM transmitter generates astring of narrow pulses with specific amplitudes and timing. Eachpulse generates a decaying oscillation at the harmonic frequencythat the receiver LC-tank is tuned at, which is then superimposedwith other oscillations across the receiver at the same frequency,to minimize the ISI. This allows for reaching high data rateswithout reducing the inductive link quality factor (to extend itsbandwidth), which significantly improves the range and selectivityof the link. The PHM receiver architecture is based on non-co-herent energy detection with programmable bandwidth andadjustable gain. The PHM transceiver was fabricated in a 0.5- mstandard CMOS process, occupying 1.8 mm . The transceiverachieved a measured 10.2 Mbps data rate with a bit error rate(BER) of at 1 cm distance using planar implant sized(1 cm ) figure-8 coils. The PHM transmitter power consumptionwas 345 pJ/bit and 8.85 pJ/bit at 1 cm and zero link distances,respectively. The receiver dissipates 3 mW at 3.3 V supply voltage.

Index Terms—CMOS, implantable medical devices, impulseradio ultra wideband, inductive coupling, low power, near-field,neuroprostheses, pulse harmonic modulation, telemetry, trans-ceivers.

I. INTRODUCTION

T RANSCUTANEOUS data telemetry is one of the mostimportant functions in a group of implantable medical

devices (IMDs), known as neuroprostheses, which substitutesensory or motor modalities that are lost due to an injury or adisease [1]. Well-known examples are the cochlear implantsand visual prostheses, which need to transfer a large volumeof data from external artificial sensors to the IMD, or invasive

Manuscript received July 20, 2010; revised February 13, 2011; acceptedMarch 03, 2011. Date of publication May 10, 2011; date of current versionMay 25, 2011. This paper was approved by Associate Editor Ranjit Gharpurey.This work was supported in part by the National Institutes of Health, NIBIB,grant 1R21EB009437-01A1, and the National Science Foundation under awardECCS-824199.F. Inanlou is with the School of Electrical and Computer Engineering,

Georgia Institute of Technology, Atlanta, GA 30308 USA (e-mail:[email protected]).M. Kiani and M. Ghovanloo are with the GT-Bionics Lab, School of Elec-

trical and Computer Engineering, Georgia Institute of Technology, Atlanta, GA30308 USA (e-mail: [email protected]).Color versions of one or more of the figures in this paper are available online

at http://ieeexplore.ieee.org.Digital Object Identifier 10.1109/JSSC.2011.2134570

brain–computer interfaces (iBCI), which collect a massiveamount of data from the central neural system (CNS) andtransfer the data across the skin to the outside of the bodyto control the patient’s environment or prosthetic limbs aftersignal processing [2]–[7].The challenges involved in designing transcutaneous data

links relate to the extremely limited space and power availableto the IMD for establishing a wideband and robust connection.Because of the significant electromagnetic field absorptionin the tissue, which exponentially increases with the carrierfrequency, high bandwidth must be achieved at the lowestpossible carrier frequencies [8], [9]. This requirement rulesout the majority of commercially available wideband wirelessprotocols, such as Bluetooth or WiFi, which operate well in theair at 2.4 GHz. On the other hand, there are specific standards,such as Medical Implant Communication Service (MICS),operating in the 402 405 MHz band, which can only offer alimited bandwidth (300 kHz) [10]. Therefore, there is a needfor low power, robust, and wideband wireless links with novelmodulation techniques that are specifically tailored towardshigh performance IMD applications.Near-field inductive coupling is the most common method

that has been utilized for establishing wideband data telemetrylinks with IMDs. In these links, robustness can be measuredby bit error rate (BER) in the presence of various sourcesof external interference, supply ripple, load changes, digitalswitching noise, and coupling variations due to mechanicalvibrations and coils misalignments [9]. It should be noted thatsimilar applications of the near-field data transmission canbe found in radio-frequency identification (RFID), contact-less smartcards, and high throughput wireless sensors, wherebatteries are avoided due to extreme size, cost, and lifetimeconstraints [11], [12].The majority of modulation techniques that have been used

in near-field inductive links for IMD applications modify a si-nusoidal carrier signal based on the data to be transferred acrossthe link. Amplitude shift-keying (ASK), frequency shift keying(FSK), binary/quadrature phase shift keying (BPSK/QPSK),and load shift keying (LSK), leading to ASK, are examples ofsuch methods [9], [13]–[17]. The use of carrier signals for datatransmission was attractive in the early IMDs because the samelow frequency data carrier could be used for transferring powerto the IMD. However, the power carrier has to be separatedfrom data carrier in the modern high performance IMDs, inwhich much wider bandwidth is required for data while thepower carrier frequency cannot be increased due to excessive

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INANLOU et al.: A 10.2 Mbps PULSE HARMONIC MODULATION BASED TRANSCEIVER FOR IMPLANTABLE MEDICAL DEVICES 1297

loss in the tissue at higher carrier frequencies [18]. On theother hand, the use of high frequency data carriers for robustwideband communication with IMDs require complex andpower consuming frequency stabilization RF circuits, such asphase-locked loops (PLL), which is not desired.This has motivated us to devise a new data transmission

technique for near-field wideband applications, called pulseharmonic modulation (PHM), which has similarities to theimpulse radio ultra-wideband (IR-UWB) in the far-fielddomain [19], [20]. It should be noted that there are otherpulse-based near-field data transmission methods, developedfor chip-to-chip communication and body area networks [21],[22]. However, they require an inductive link with a low qualityfactor, , to achieve wide bandwidth, which is not suitable forthe IMD applications, where higher transmission distance andbetter selectivity are needed.The PHM transmitter generates a string of narrow pulses with

specific amplitudes and timing to transmit each bit across a pairof high-Q LC-tank circuits. Each pulse generates a decaying os-cillatory response at the harmonic frequency that the receiverLC-tank is tuned at, which adds to the oscillations at the samefrequency that are resulted from subsequent pulses within thatbit period to minimize the inter-symbol interference (ISI) at thereceiver input. This allows reaching high data rates without re-ducing the inductive link quality factor by adding dissipativeelements to extend its bandwidth. This method also improvesthe signal-to-noise ratio (SNR) at the receiver, leading to im-provements in the BER, transmission range, selectivity, and ro-bustness against misalignments of the link [19].We recently described the PHM concept and its theoretical

analysis in [19] along with measurements from a discreteproof-of-concept prototype. In this paper we present a fullyintegrated CMOS transceiver based on the PHM technique,which block diagram is shown in Fig. 1. On the transmitter side(Tx), is charged up to a voltage set by a digital-to-analogconverter (DAC) and is charged up to . is then dis-charged into the primary coil followed by via an LC-Drivercircuit according to a specific timing that is dictated by anFPGA that accepts the serial data bit stream to be transmitted(Tx-Data).The sharp current impulses passing through dueto discharges, couple onto the secondary tank,and initiate/terminate an oscillation at the resonancefrequency, , across the receiver (Rx) input.Inside Rx, which operates based on a non-coherent energy

detection (ncED) scheme, the received signal is amplified,squared, and low-pass filtered, and finally, a comparator re-covers the serial data bit stream. A brief overview of inductivelinks for data telemetry as it applies to the PHM is includedin Section II, followed by detailed PHM transceiver circuit

Fig. 1. Block diagram of the pulse harmonic modulation based transceiver.represents the parasitic capacitance of and switches at the output of the LCDriver.

descriptions in Section III. Section IV presents the transceivercharacterization and measured results, followed by discussionsin Section V. Finally, the concluding remarks are included inSection VI.

II. PULSE HARMONIC MODULATION FOR NEAR-FIELDINDUCTIVE DATA TELEMETRY

A. Inductive Link Impulse Response

Fig. 2 shows a lumped equivalent circuit model of an induc-tive link. The primary ( transmitter) and secondary (receiver) sides are composed of LC-tank circuits and their asso-ciated parasitic components ( and ). tank is tuned at, while the tank, depending on the data rate and trans-

mission range requirements, can be either tuned to or left atits self resonance frequency (SRF), in which case in Figs. 1and 2 simply represents the parasitic capacitance of . Sinceand are loosely coupled (small ) and the current in

tank is very small, unlike inductive power transmission links[23], we can safely neglect the effect of loading on theTx circuitry to simplify our equations. Hence, the inductive linktransfer function in the S-domain can be described as in equa-tion (1), shown at the bottom of the page, where , , , and

are the Tx output voltage, Rx input voltage, current passingthrough , and induced voltage across , respectively. Otherparameters are lumped circuit elements in Fig. 2. is com-posed of two second-order systems, one originating from theprimary and the other from the secondary LC-tanks, each ofwhich can be expressed as

(2)

(1)


Fig. 2. Lumped model for near-field data telemetry through coupled coils.

where is the damping ratio, is the natural frequency, andis the natural damping frequency of the

system. From (1) and (2), these second-order system parame-ters can be expressed in terms of the lumped circuit elementsin Fig. 2,

(3)

(4)

(5)

(6)

Assuming both second order systems are under damped, i.e.,and , which is often the case for the LC-tanks used indata telemetry links, (1) can be rearranged as

(7)

Now we can break up into the sum of its first order com-ponents

(8)where

(9)

(10)

and apply the inverse Laplace transform to find the impulse re-sponse for the inductive link

(11)

(12)

High-Q is desired for the inductive data transmission linksused in IMD applications to improve transmission range and

robustness of the link against interference without increasingthe transmitted power. and can be calculated from (5)and (6) by substituting them in , which indicates thatshould be small and thus, . As a result, in (12)

can be simplified into two exponentially decaying oscillations,one with a long time-constant ofon the secondary and the other with a short time-constant of

on the primary. The sum of these twoterms will result in an oscillation across in Fig. 2, whichbuilds up rapidly but decays slowly [19]. The envelope ofcan be expressed as

(13)

B. Pulse Harmonic Modulation

Using the inductive link impulse response from Section II-A,one can calculate for any arbitrary input waveform by con-volving it with . However, if we only apply narrow pulsesacross , with pulse width and amplitudeto reduce Tx power consumption, then we can expectto appear across [19]. On the other hand, in high-speedpulse-based data transmission, it is important for the receivedsignal (i.e., the impulse response), resulted from transmitting asingle bit, to rise sharply (small ) to enhance detection and de-cline rapidly (small ) to minimize ISI with the following bit,and reduce BER. Both of these requirements suggest thatand should be lowered to improve bandwidth, which is themethod that has been adopted in previous carrierless near-fielddata links [21], [22]. However, this is not a suitable choice forrobust transcutaneous communications.Considering that has a sharp rise time and a slow decay,

in PHM we generate a rapid oscillatory response across byapplying a first pulse, called the initiation pulse, across at theonset of every digital bit ‘1’ . Then we transmit a secondpulse (or pattern of pulses), called the suppression pulse(s), be-fore the end of the bit period, , to create a second oscillation.We choose the amplitude, , and delay, , of the suppressionpulse such that the second oscillation has the same amplitudeas the first one by the time that it appears across , but 180out of phase. The result is the cancellation of both oscillatoryresponses after the arrival of the suppression pulse, leading toa rapid decline in the ISI, which is the amplitude of the oscilla-tion across by the time that a new bit is due to begin. In thisimplementation, for a digital bit ‘0’, we simply do not transmitany pulses [19].In order to minimize BER, we should first allow the oscilla-

tion resulted from the initiation pulse to build up. According to(13), this can take . Hence, to reduce the buildup time,according to (5), and should be as small as possible.Infact the parasitic capacitance of is often sufficient to form. Then we should apply the suppression pulse at an odd mul-

tiple of a half cycle to ensure that the two responses aregoing to be out of phase

(14)

where is the number of cycles that is allowed to havebefore it is suppressed.


Fig. 3. (a) Schematic diagram of the LC-Driver block in Fig. 1(b) LC-Drivertiming diagram ( is the signal controlling the MUX in the PHM Tx in Fig. 1,which connects to either or the comparator output).

For a perfect cancellation, when choosing the suppressionpulse amplitude, , the decay of from the initiation pulseat should be considered. The amplitude of can befound from its envelope in (13). Since , the first term,which represents the buildup period, can be neglected and theratio between the suppression and initiation pulses can be foundfrom

(15)

Using (2)–(15), and can be easily calculated from the in-ductive link lumped circuit values and other parameters, suchas the desired data rate and pulse width , todesign a PHM-based near-field data telemetry link.In practice, perfect cancellation between initiation and sup-

pression pulses is not possible due to timing jitter, parasitic com-ponents, and process variations [19]. Nonetheless, as shown inSection IV, PHM is an effective method for reducing the ISI andachieving wide bandwidth at lowBERwithout lowering the linkquality factor or consuming too much power.

III. PULSE HARMONIC MODULATION BASED TRANSCEIVER

A. Transmitter Design

The PHM transmitter, which is enclosed in the upper dashedbox in Fig. 1, operates by charging a pair of capacitors, and, up to and , respectively, and discharging

them into to generate sharp initiation and suppression pulses.is provided by a 5-bit DAC, which controls the voltage

at the lower terminal of . The DAC was designed based onfolded resistor string architecture using 50 k resistors [24].Fig. 3(a) shows a more detailed schematic of the LC-Driverblock in Fig. 1. This circuit, which is controlled by an off-chipFPGA through and signals, can operate either in single-ended or differential modes, which switching diagram can beseen in Fig. 3(b).In the single-ended mode, ‘0’ and the right node ofis always connected to by . In order to generate the

initiation pulse, in Fig. 1 is set to ‘0’, causing the multiplexerin Fig. 1 (MUX) to connect to , to turn on andcharge to full scale, i.e., . During this charging period

‘0’ and gate is open. Therefore, no current passesthrough . At the onset of a bit ‘1’ period, toggles to ‘1’for a very short time period, (only a few ns in this design),during which the inductor current, , starts increasing at a rateproportional to . This current variation in isresponsible for inducing a voltage across , which is depictedas in Fig. 2. Of course there will alsobe voltage drops across and , depending on their sizes,which need to be considered in a more detailed analysis. To endthe initiation pulse, is toggled back to ‘0’, at which timeshorts the left node of to and provides a path for toreturn back to zero without causing undesired oscillations.After terminating the initiation pulse, the FPGA applies the

PHM time delay in (14), , and at the same time prepares thetransmitter for generating the suppression pulse by setting‘0’, similar to the initiation pulse. The difference, however, isthat this time, ‘1’ , causing the MUX to connect tothe output of a comparator, which compares and .The comparator output is initially high because

. Therefore, is turned on and charges up to thepoint where . At this time, the comparator outputgoes low and turns off to maintain the voltage acrossat . The rest of the steps for generating the sup-pression pulse is similar to the initiation pulse, and includestoggling ‘1’ for to connect in parallel withand induce across . Ob-viously the 5-bit DAC input, S[1:4], can be adjusted such that

, which was calculated in (15).In the differential mode, the initiation pulse is generated by

the right side of the LC-Driver in Fig. 3(a) by connecting the leftnode of to ( ‘0’) and charging to full scale, i.e.,, through . At the onset of a bit ‘1’ period, toggles

to ‘1’ for , during which increases towards right. Thenboth and are lowered, during , to connect both endsof to and reduce its current. At the same time, ischarged to and the suppression pulse is generatedby the left side of the LC-Driver in the exact same way as in thesingle-ended mode. It should be noted that in this mode, sincethe suppression pulse is already out-of-phase with respect to theinitiation pulse ( flowing in opposite directions), unlike (14),should be an even multiple of a half cycle, or simply

(16)

of and should be small to charge andrapidly and meet the timing requirements of the desired datarate. In other words, and for and

should be large. On the other hand, driving larger devicesresults in higher power consumption per bit. In this design wechose as a compromise between andpower consumption requirements. Similar constraints should beconsidered in the design of , , , and , which aremuch larger devices responsible for driving , and as suchshould be designed considering the chip area constraints also. Inthis design, and


were selected based on specifications and post-layout sim-ulation results. It should also be noted that the parasitic capac-itances added by these switches to and terminals aredwarfed by the parasitic capacitance of and values ofand , which are off-chip components.

B. Receiver Design

The PHM signal at the Rx input, , is similar to the low-fre-quency equivalents of the IR-UWB on/off keying (OOK) sig-nals, in which the presence or absence of a number of oscilla-tions within a certain band represent bits ‘1’ or ‘0’, respectively[25], [26]. Among different architectures for such receivers,ncED is popular due to its low complexity [27]–[29]. This archi-tecture does not require high power consuming blocks such aslocal oscillator (LO) or PLL for down-conversion either, whichmakes it suitable for low power applications. A recent com-parison in terms of energy per bit and data rate between co-herent and non-coherent UWB and narrowband receivers in [30]showed an order of magnitude lower power consumption innon-coherent receivers for data rates up to 10 Mbps. Moreover,applying this method to high-Q inductive links, which have highselectivity, can diminish some of the drawbacks of the ncED ar-chitecture, such as susceptibility to noise and interference.The block diagram of our ncED-based PHM Rx is shown in

the lower dashed box in Fig. 1. The PHM signal across isamplified and squared before passing through a Chebyshev LPFfor its energy to be compared with a reference value, which isdetermined by the noise floor and level of interference in thesystem. All blocks are designed fully differential to have betterimmunity against common mode interference, supply ripple,and noise.1) LNA: The LNA block, shown in Fig. 4(a), which has

common-source cascode architecture with untuned load, pro-vides 27 dB gain at the input of the PHMRx. In order to achievedata rates above 10 Mbps, should be tuned at a high fre-quency to provide enough oscillations per bit at the Rx input.We chose MHz for resonance frequency and100 MHz for the LNA bandwidth. Resistive load was used toavoid off-chip inductors and capacitors, which are needed oth-erwise due to the low frequency of the RF signal. The high-resonance of the -tank at the LNA input eliminates theneed for any additional bandpass filtering. The LNA gain canbe reduced to 13.5 dB by closing in Fig. 4(a) when theinput RF signal is large due to strong coupling between and. In this case the LNA power consumption is also reduced in

half by cutting the tail current to save power.2) Passive Self-Mixer: The amplified RF signal is applied

to a passive self-mixer, shown in Fig. 4(b), to be converted to abaseband signal proportional to its energy. The RF and LO portsof the mixer are tied together to implement a squaring function.Transistors in this circuit act as switches with the gate-sourcevoltage close to the threshold voltage to provide the maximumpossible gain.3) Third-Order Chebyshev Low-Pass Filter: The squared

signal at the output of the mixer passes through a third-orderChebyshev LPF with 0.5 dB ripple in the pass band,which is shown in Fig. 5(a) [31]. The normalized filter transfer

Fig. 4. Receiver front-end circuits. (a) Adjustable gain LNA. (b) Passive self-mixer circuit.

Fig. 5. (a) Block diagram of the third-order Chebyshev low-pass filter.(b) Schematic diagram of the fully differential OTA with CMFB.

function, which has been realized by a bi-quad stage followinga first-order low-pass stage, can be defined by

(17)

where and determine the DC gain and the 3 dB cut-offfrequency dB rad/s . is the capacitance andis the transconductance of each operational transconductanceamplifier (OTA) stage.

dB is adjustable by 3 bits, a [0:2], which change the valueof in (17). This method was preferred over changing tokeep the low frequency gain of the LPF constant regardless of

dB. TheOTAwas implemented by a fully differential ampli-fier, shown in Fig. 5(b). Cascode stages and source degeneration


TABLE ITHIRD-ORDER CHEBYSHEV LOW-PASS FILTER SPECIFICATIONS

*From post-layout simulations.

Fig. 6. Schematic diagram of the variable gain amplifier (VGA).

resistor were employed to increase the output impedance andlinearity of the OTA, respectively. In this circuit, if transconduc-tance of the input transistors, and , is sufficiently large,i.e., , then , independent of the tailcurrents and input voltage swing. A common-mode feedback(CMFB) circuit was also added to set the commonmode voltageat the OTA outputs to a predetermined value, . is used inthe CMFB circuit to increase the stability of the CMFB loop byreducing its gain. Table I shows the specifications of the Cheby-shev LPF.4) Variable Gain Amplifier (VGA): In order to improve the

sensitivity of the PHM Rx for different coupling distances, theRx gain can be adjusted both in the RF front-end (1-bit coarsecontrol) and baseband (3-bits fine control) stages by the LNAandVGA, respectively. A fully differential open-loopVGApro-viding 10 to 30 dB gain in 8 steps (G[0:2]) was implementedusing the circuit shown in Fig. 6. In order to reduce power con-sumption, the VGA tail current was also changed with the gaincontrol bits to obtain the necessary gain-bandwidth product.5) Comparator: Fig. 7 shows the schematic of the fully dif-

ferential high speed comparator, which has positive feedback atthe input. This comparator has an adjustable reference voltage,

, that sets the threshold at which the comparator switches.The voltage difference at the VGA output is compared toto generate the received Rx-Data bit stream. The effects of thecommonmode voltage variations at the comparator input are re-duced because of the fully differential design of both VGA andcomparator.

Fig. 7. Fully differential comparator circuit.

Fig. 8. PHM transceiver chip microphotograph.

Fig. 9. PHM transceiver measurement setup. Inset: Inductive telemetry linkmade of a pair of planar figure-8 coils [32] ( and in Fig. 1) on FR4 PCB,on which the transceiver chips have been directly wire-bonded.

IV. MEASUREMENT RESULTS

The PHM transceiver prototype, shown in Fig. 8, was fabri-cated in a 0.5- m 3M2P standard CMOS process, occupying0.61 mm of chip area. Two chips were used in our experi-mental setup, shown in Fig. 9. Each chip was glued and wire-bonded directly on a 2-layer custom designed PCB (1 oz copperon 1.5-mm-thick FR4) and covered with epoxy. Each PCB in-cluded a 2-turn planar figure-8 coil (see Fig. 9 inset), whichspecifications are summarized in Table II [32]. The coils wereheld in parallel and perfectly aligned using only non-conductingmaterials, such as PVC and Plexiglas. At cm between thetwo coils and perfect alignment, they had a simulated value of

[32]. In addition, based on


TABLE IIFIGURE-8 COILS SPECIFICATIONS

*From HFSS finite element analysis simulations.

at MHz, the bandwidth of the inductive link can becalculated as MHz. In order tominimize the effect ofinterconnects and SMA connectors on the coils’ magnetic fieldwe added 3.4 cm and 4.25 cm clearance on the PCB between theTx and Rx coils and their associated PHM transceivers, respec-tively. In practice, the data coil and transceiver microassemblywill be packed inside an IMD with shorter interconnects andsmaller parasitic effects, while the tissue volume conductor canadd to the coils’ parasitic components and reduce their SRF [33].Evaluation of the PHM transceiver performance in a more real-istic setup, inside tissue volume conductor, is among our futureresearch directions.Generation of the random serial data bit stream to be trans-

mitted, Tx-Data, and BER analysis of the recovered data,Rx-Data, in Fig. 1 were conducted by a pair of TektronixGB1400 (GigaBERT), shown in Fig. 9. Once the two Giga-BERTs are synchronized via a separate hardwired clock signal,Tx-clk in Fig. 1, they can measure the wireless link BER in realtime. Instead of the receiver block in the PHM transceiver chip,we initially used a custom-designed discrete RF receiver [19],in which we had easier access to the internal nodes, to observeand evaluate the effects of PHM on the received signal. Wetried to avoid probing the inductive link directly, which couldchange its characteristics.In order to achieve Mbps, which was our target

for this PHM transceiver prototype, the oscillation acrossmust be initiated and suppressed within ns. If we ded-icate complete cycles for the oscillation to build up afterthe initiation pulse, and three more cycles for the oscillation todie out after the suppression pulse, before the next bit arrives,then needs to be 60 MHz. It should also be pointed out thatin a second order system with , such asin Fig. 2

(18)

where is the peak magnitude of the transfer function,i.e., the second term in the denominator of (1). Therefore, in-creasing by reducing while maintaining constant canlead to higher and higher gain for the inductive link, bothof which can lead to better SNR at the receiver input. This is amajor advantage of the PHM over previous wideband carrier-less data transmission methods, which can only operate whenthe link Q-factor is low [19]. High gain and Q-factor are key ininductive data links used in IMDs because the coils distance, ,is in the cm range, orders of magnitude larger than applicationssuch as chip-to-chip communication [21], and there might be astrong interference from a nearby inductive power carrier [18].For the above reasons, we chose MHz. Using the

coils specified in Table II and , the main PHM param-

Fig. 10. Transmitter waveforms at the LC-Driver input and receiver waveformsat the LNA and LPF outputs of the custom designed receiver, (a) with and (b)without PHM, when MHz, Mbps, , ns,

, ns, and mm. Elimination of the suppression pulse in(b) renders the received data completely undetectable due to strong ISI.

eters were found to be ns and based on(14) and (15), respectively. In Table II, it can be seen that notuning capacitor has been used for the primary coil to re-duce Tx power consumption. We also noticed that at this ,since s, the performance of the PHM trans-ceiver is not very sensitive to , and the wireless link also worksfine with . Therefore, the PHM Tx was set to operatein the single-ended mode in order to further reduce power con-sumption. Fig. 10(a) and (b) from top to bottom show one of theLC-Driver control signals, , which was applied to the PHMTx as shown in Fig. 1, the received RF signal at the output of theLNA, which is proportional to at the input of the receiver,and the LPF output, which is proportional to the energy con-tent of the received pulse around , with and without PHM, re-spectively. It can be seen that the elimination of the suppressionpulse in Fig. 10(b) renders the received data completely unde-tectable at Mbps, while Fig. 10(a) clearly showsthe effectiveness of the PHM in reducing the ISI. Considering

and the model shown in Fig. 2, the channelloss in these measurements was dB.Fig. 11 shows a snapshot of the key measured waveforms

in the PHM transceiver setup of Fig. 9. From top, the trans-


Fig. 11. PHM transceiver waveforms from top: Transmitted serial data bitstream at 10.2 Mbps, one of the LC-Driver control signals, receiver VGAoutput, and recovered data bit stream, (a) with and (b) without PHM.It canbe seen that while everything between these two measurements are the same,without PHM the Rx Data in (b) is completely erroneous.

mitted serial data bit stream at Mbps, of theLC-Driver, receiver VGA output, and recovered serial data bitstream are shown with the figure-8 coils aligned at mm.In Fig. 11(b), because of the strong ISI in the absence of the sup-pression pulse, the receiver has failed to recover the serial databit stream at this data rate. On the other hand, utilizing the PHMtechnique in Fig. 11(a) has resulted in correct detection of theserial data bit stream in an otherwise similar condition.To better characterize the performance of the PHM-based

transceiver, we measured the BER at various data rates, whilechanging from 5 to 10 mm. The results, shown in Fig. 12, in-dicate that if the acceptable BER limit is considered , thenthe best data rate that can be achieved with the current PHMtransceiver at mm is 10.2 Mbps. The PHM Tx supplyvoltage, , in this case was at 3.3 V, consuming 3.52 mW,which is the equivalent of 345 pJ/bit. The PHM Rx power con-sumption in these conditions was 3 mW. Consuming the sameamount of power at mm resulted in Mbpsat BER . It should be noted that the receiver bandwidthlimitation is the main reason for high BER at 11.4 Mbps and12 Mbps. The cutoff frequency of the Chebyshev LPF in the Rxshould be increased in order to achieve higher data rates. Alsothe signal to interference ratio decreases due to the ISI when thePHM pulses are too close to each other at high data rates, which

Fig. 12. BER versus coils distance, , at different data rates, when PHM-Txsupply voltage is constant at .

TABLE IIIPHM-BASED TRANSCEIVER SPECIFICATIONS

leads to more BER degradation. Without PHM, on the otherhand, the data rate could only reach 1.5 Mbps at mmwith BER . Table III summarizes the key specificationsof the PHM transceiver ASIC.Even though the present transceiver has not been optimized

for minimum power consumption in this large feature sized andrelatively high threshold voltage process, we tried to lower thePHM Tx power by reducing its from 3.3 V down to 2.4 V,while measuring the BER at mm. Fig. 13 depicts theresults of this experiment, which demonstrates the strong rela-tionship between BER, Tx power, and Rx sensitivity.In order to accurately measure the minimum required energy

to transfer high speed data and compare the PHM transceiverperformance with other recently published near-field datatelemetry links, we reduced down to 0.1 mm by separatingthe two coils with a sheet of paper and lowering downto 1.2 V. Since the FPGA was still operating at 3.3 V, thePHM-Tx control signals in Fig. 1 were also divided down to1.2 V. Finally, was reduced from 7 ns to 3 ns using theFPGA gate delays [34]. The average energy consumption inthis case was measured by differentially recording the voltageacross a resistor in series with the PHM-Tx supply pin, and inte-grating it over a certain period of time as random data was beingtransferred across the link at 10.2 Mbps. The results showedthat for a BER the PHM-Tx power consumption wasonly 8.85 pJ/bit, which can be further reduced by migrating thePHM transceiver to smaller feature sized processes.


TABLE IVBENCHMARKING OF RECENT INDUCTIVE DATA TELEMETRY LINKS

*Only the links with potential IMD application are listed.**Calculated from component values given in the paper.Load shift keying consumes low power for data transfer at the cost of reducing the deliveredpower to the IMD by up to 50%.Biphase modulation.

Fig. 13. BER versus transmitter power consumption at mm andMbps, when of the PHM-Tx has been reduced from 3.3 V to 2.4 V.

V. DISCUSSION

It can be concluded from the measurement results using thecoils in Table II that matching the primary and secondary reso-nance frequencies, while desired, is not a precondition to reli-able data recovery with the PHM technique. It was observed inmeasurements that tuning the primary coil at MHz in-creases the received PHM signal amplitude ( in Fig. 2) at thecost of higher power consumption in Tx. To better understandthe effects of mismatch between the primary and secondary res-onance frequencies on the ISI and received signal amplitude,which are important in the PHM performance, we swept inFig. 1 in a simulation while keeping all other parameters con-stant according to Tables II and III. The result in Fig. 14 showsthat sweeping in a wide range does not have a significanteffect on the ISI.The key to successful ISI cancellation in the PHM is proper

adjustment of and to a lesser extent on the Tx based on theresonance frequency and time constant of the LC-tank on the Rxaccording to (14)–(16). Therefore, whether PHM is being usedfor uplink (i.e., from inside to outside of the body) or downlink(from outside into the body), it is always possible to accuratelyadjust either (e.g., by using an array of tuning capacitors)

Fig. 14. Simulated Rx amplitude ( in Fig. 2) for both data and ISI versusprimary parasitic capacitance, i.e., in Fig. 1, while keeping all other param-eters constant according to Tables II and III.

or (e.g., by a programming delay), respectively, whichever isaccessible from outside of the body, to create optimal conditionsfor the PHM operation. If the same pair of LC tanks is to beused for both transmitting and receiving data in a half-duplexfashion, a closed loop scheme can be employed to adjust andbased on the ISI or BER, in the downlink or uplink modes,

respectively, both from outside of the body.Finally, it should be noted that for a complete stand-alone re-

ceiver, a clock and data recovery (CDR) scheme is required.In high performance IMDs, such as cochlear implants and vi-sual prostheses, the power needed for the operation of the im-planted device is transmitted via a separate inductive link, re-sponsible for wireless power transfer [23], [32]. In our CDRscheme, we plan to utilize the power carrier, which frequencyis adjusted by a crystal oscillator, to generate synchronized Txand Rx clocks on both sides of the skin barrier. Oversamplingor simple encoding schemes, such as Manchester, are amongother possible alternatives to recover synchronized clock fromthe data bit stream. Adding CDR to the PHM transceiver is thesubject of our ongoing research.

VI. CONCLUSION

We have presented the integrated circuit implementation ofa low power transceiver based on PHM technique in a stan-dard CMOS process. The PHM technique, which was intro-


duced for the first time in [19] with a discrete prototype, hasenabled us to achieve a high data rate (10.2 Mbps) despite uti-lizing a narrowband (1.4 MHz) high- inductive link, whichis needed in IMD applications to extend the transmission range, increase the SNR, improve the link selectivity, and mini-

mize the ISI at the receiver input. The PHM-Tx block is com-posed of an LC-Driver circuit to generate sharp initiation andsuppression pulses with adjustable amplitudes and timing. ThePHM-Rx block is a highly configurable ncED receiver with lowpower consumption. It can be seen in Table IV that the PHMtransceiver can achieve a data rate of 10.2 Mbps with a BER of

across a 10 mm inductive link, set up between 1 and2.25 cm figure-8 coils. This performance has been achievedwith an unprecedented of 541.9 and 48 at low frequencyand 67.5MHz, respectively. The differential design of the trans-ceiver building blocks and utilization of high- inductors, madepossible by the PHM, are expected to immune this data link, to ahigh extent, against external interference, particularly from thestrong power carrier in a multiband wireless link for IMD ap-plications [18].

ACKNOWLEDGMENT

The authors would like to thank U. M. Jow from theGT-Bionics Lab for design and fabrication of the data telemetrycoils.

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Farzad Inanlou (S’06) received the B.S. degreefrom Ferdowsi University, Mashhad, Iran, in 2006,and the M.S. degree from Boston University, Boston,MA, in 2008, both in electrical engineering.During summer 2008, he was with Boston Sci-

entific Neuromodulation, Valencia, CA, where heworked on the development of implantable spinalcord stimulation (SCS) devices for the treatmentof chronic pain. Since 2008, he has been at theGeorgia Institute of Technology, Atlanta, GA,where he is working toward the Ph.D. degree. His

research interest is the design of Analog/RF integrated circuits for biomedicalapplications.Mr. Inanlou is the recipient of the 2010 EAPSI Fellowship from the National

Science Foundation, and the Georgia Institute of Technology President’s Fel-lowship.

Mehdi Kiani (S’09) received the B.S. degree fromShiraz University, Shiraz, Iran, and the M.S. degreefrom Sharif University of Technology, Tehran, Iran,in 2005 and 2008, respectively.He joined GT-Bionics Lab at the Georgia Insti-

tute of Technology, Atlanta, GA, in 2009 where heis working towards the Ph.D. degree.

Maysam Ghovanloo (S’00–M’04–SM’10) wasborn in 1973 in Tehran, Iran. He received the B.S.degree in electrical engineering from the Universityof Tehran, Tehran, Iran, in 1994, the M.S. degreein biomedical engineering from the AmirkabirUniversity of Technology, Tehran, Iran, in 1997, andthe M.S. and Ph.D. degrees in electrical engineeringfrom the University of Michigan, Ann Arbor, in2003 and 2004, respectively.From 2004 to 2007, he was an Assistant Professor

in the Department of Electrical and Computer Engi-neering, North Carolina (NC) State University, Raleigh. In June 2007, he joinedthe faculty of Georgia Institute of Technology, Atlanta, where he is currently anAssistant Professor and the Founding Director of the GT-Bionics Laboratory inthe School of Electrical and Computer Engineering. He has authored or coau-thored more than 70 peer-reviewed conference and journal publications.Dr. Ghovanloo is an Associate Editor of the IEEE TRANSACTIONS ON

CIRCUITS AND SYSTEMS II, IEEE TRANSACTIONS ON BIOMEDICAL CIRCUITSAND SYSTEMS, and a member of the Imagers, MEMS, Medical, and Displays(IMMD) subcommittee at the International Solid-State Circuits Conference(ISSCC). He is the 2010 recipient of a CAREER award from the NationalScience Foundation. He has also received awards in the 40th and 41st DesignAutomation Conference (DAC)/ISSCC Student Design Contest in 2003 and2004, respectively. He has organized several special sessions and was a memberof Technical Review Committees for major conferences in the areas of circuits,systems, sensors, and biomedical engineering. He is a member of the Tau BetaPi, AAAS, Sigma Xi, and the IEEE Solid-State Circuits Society, IEEE Circuitsand Systems Society, and IEEE Engineering in Medicine and Biology Society.

A 10.2 Mbps Pulse Harmonic Modulation Based Transceiver for Implantable Medical Devices

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