4 Detailed Work Packages - Ofcom · The majority of S-band radars are in the frequency range 2.7 to...

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Study into Spectrally Efficient Radar Systems in the L and S BandsOfcom Spectral Efficiency Scheme 2004 - 2005 (SES-2004-2) 13

4 Detailed Work Packages

4.1 A review of the spectrum utilised by radar in the L & S bands and recommendations on which (if any) could be moved to lightly-used higher frequency bands

4.1.1 Spectrum review The subject of spectrum usage in the radar bands has been covered in several preceding reports; in particular [1], [2] and [3], which were all produced for Ofcom. There is probably little which is new that can be added to previous work, and this section provides a summary of the important points discussed in those reports.

4.1.1.1 L-Band - 1215 to 1365 MHz The frequency band 1215 to 1350 MHz is used by the Directorate of Airspace Policy (DAP) for a network of primary radars, subject to coordination with MOD. In practice the civilian radars are limited to frequencies above 1243 MHz, whilst the band 1350 to 1365 MHz may also be used by DAP, subject to coordination with DTI. There are 34 L-band frequencies assigned to UK civil ATC radar, and 4 of these frequencies have more than one radar assigned to them. The bandwidths of these assignments (at the -20dB points) vary between 3.9 MHz and 16.7 MHz. Most radars require more than one frequency in order to enable frequency diversity or multi-pulse operation.

These radars are used for the control of en-route air traffic into, over and out of the UK. They typically operate out to 200nm range, and usually they also have associated secondary surveillance (SSR) radars. There are 9 en-route civilian L-band radars in the UK, together with one other L-band equipment (at Heathrow) configured for TMA operations, and another which is used only for training. There are also 3 MOD L-band radars. The radars located in England tend to be situated towards the eastern side of the country. (Just outside the band under consideration, secondary radar uses the frequencies 1030 and 1090 MHz.)

This band is also used for position determination using the GPS L2 frequency (1227 MHz), for space to earth communications by earth exploration satellites, by amateurs (using wide-band TV repeaters), and by meteorological wind profiling systems. The European Galileo satellite position-location system will also use frequencies in L-band.

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Study into Spectrally Efficient Radar Systems in the L and S Bands 14 Ofcom Spectral Efficiency Scheme 2004 - 2005 (SES-2004-2)

4.1.1.2 S-Band – 2700 to 3100 MHz The band 2700 to 3100 MHz is widely used by Directorate of Airspace Policy (DAP) & MOD for S-band primary surveillance radar. The lower part of the frequency band (2.7-2.9 GHz) is used for aeronautical radio-navigation, and the band 2.9-3.1 GHz is generally used for maritime radio-navigation. Other users of S-band may include aircraft telemetry, command and control of UAV’s, and electronic newsgathering and outside broadcasting.

There are 55 S-band frequencies assigned to UK civil ATC radar, of which 12 are used at more than one radar. Typically the bandwidths of these assignments (at the -20dB points) vary between about 2.4 and 10 MHz.

Most civil regional airports in UK are equipped with S-band radars, which are generally used for monitoring Terminal Manoeuvring Area (TMA) operations, i.e. airport approach, take-off and landing; usually they give coverage out to a range of 60nm. (Some airfields also operate X-band radars in addition to the S-band equipment, mainly for controlling vehicle movement around the airfield; typically the X-band radars have a maximum range of 30nm.) The majority of S-band radars are in the frequency range 2.7 to 2.9 GHz. From information available in the public domain there are about 120 S-band radars operating in UK, of which 84 are military. Some of the military radars could be used for purposes other than air traffic control (details are not readily available), but the civilian systems are all used for ATC.

Of the civil systems, 19 are pulse compression systems and 15 are single plain pulse systems. The number of frequency allocations required for each radar varies according to main and standby requirements. At the -20dB bandwidth points the civil radars use a total of 460 MHz bandwidth in a total occupied spectrum of 375MHz. The limits on out-of-band radiation are defined by the -40dB bandwidths - these are much greater than the operationally required bandwidths. At these levels there is a very high degree of frequency overlap between different radars, but it is possible to operate in this manner due to the use of PRFD.

4.1.1.3 Spectrum planning considerations Many of the civil radar systems are located in southern or eastern England, with a high concentration in the Greater London area (2 at Heathrow, 2 at Gatwick, others at Farnborough, Luton and Stansted). More than one-third of the total number of radars (civil and military combined – 43 radars) are located in southern England. In the London area for example, the average distance between neighbouring radars is only a few tens of kilometres, whereas in the Northern and Western parts of the country the separations are often measured in hundreds of kilometres. This uneven spatial distribution of radars means that in the South-East it is a challenging task to find adequate frequency allocations within the available spectrum, whilst in the North and West it is a much easier task.

When the possibilities for improving spectrum utilisation are investigated, it is of interest to consider whether both co-channel and adjacent channel interference may need to be taken into account. Conditions in the region of greatest radar channel demand (i.e. the London area and South-East England) may be usefully illustrated by the National Grid square TQ (100km x 100km). Within this square there are 10 radars, and for these radars the typical co-channel frequency re-use distance varies between about 70 km and 120 km, depending on the degree of terrain screening available.


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If there are 10 radars located within a 100km x 100km square the mean distance between neighbouring radars is approximately 30km. Compared with the co-channel situations (70-120km separation), neighbouring radars at a separation of 30km would experience interference levels approximately 40-80dB stronger (very much dependent on the terrain), which suggests that for these closely-sited radars adjacent channel protection criteria are likely to be an important consideration.

The problems of radar frequency assignment are complicated by the fact that it has been difficult to agree what interference thresholds should be used for planning & coordination of radar systems. Furthermore there is no ‘standard’ radar that could be used as a benchmark in such planning calculations - each radar tends to have its own particular design features, and requires individual treatment.

It may be noted that radar systems experience other unwanted signals (which may be considered as forms of ‘interference’) at present - e.g. rain, weather, birds, ground clutter, other radars. Furthermore in other radar bands there are already precedents of radar systems being engineered to coordinate with low-power devices operating in the same band. For example in UK licence-exempt devices (802.11h) have a carrier-sense capability to avoid interference to existing radars at 5.8GHz. The use of carrier-sense multiple access (CSMA) by these devices is required by the UK regulator in order to permit safe coordination with the radars.

When the possible implications are considered (in terms of spectrum utilisation) of the measures discussed later in this study, it is important to note that modifying / improving just a few isolated radars will bring relatively little benefit. The radar transmitted spectrum extends a long way in frequency from the nominal centre frequency, and the far sidelobes of neighbouring radars will often have significant regions of frequency overlap. If just a few radars are improved, then although this may make some additional spectrum available to other systems in a few particular locations, most regions will still receive significant levels of low-level out-of-band radiation from other (unimproved) radars. Isolated pockets of spectrum opportunity will be difficult to use, and are unlikely to be attractive to potential users.

The major benefits of improving radar spectral efficiency will come from widespread deployment of the technique (whichever approach is chosen), followed by re-planning of the radar frequency allocations to take advantage of the spectrum savings. For example, most of the techniques to be discussed in the rest of the report, may be regarded as allowing a reduction of the channel allocation required by each radar to some proportion, say P%, of the channel width currently occupied. (Of course, the value of P varies between the different techniques, as do the costs and the timescales for implementing them.) If all of the ATC radars were to be improved using the chosen technique, then it would be possible to implement a revised frequency allocation plan in which (to a first approximation) all the channels were simply reduced in width to P% of their current values, thus freeing up a block of spectrum for use by other systems. Such a block of contiguous free spectrum will be much more useful to other systems than isolated pockets of frequencies perched upon the sidelobes of high-power radar transmissions.


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4.1.2 Radar system aspects at higher frequencies In order to assess any impact of moving current L and S band civil radar functions to higher frequency band, the existing radar requirements must first be considered.

4.1.2.1 Existing radar requirements The performance requirements for ATC radars are detailed in the CAP 670 Air Traffic Services Safety Requirements document [4].

For this the Operational Requirement (OR) is 60 nautical miles and the SRA is 40 nautical miles. Thus the CAP 670 requirement is for the radar system to detect a 1m2 target at 60 nm with 80% single look probability of detection at 10-6 false alarm rate, and to achieve 90% single look probability of detection at 40 nm. 60nm = 111.12km and 40nm = 74.08km.

Section 10 of CAP 670 defines radar accuracy requirements:

10.1.1 When used for surveillance radar approach (SRA) purpose (i.e. primary only), the accuracy shall be better than 1 degree of bearing and 55 Metres, + 5% of target range (365).

10.1.2 When used for radar separation, the intended minimum separation standard shall be justified (366).

Section 11 of CAP 670 defines the radar resolution requirement:

For 3 NM separation the equipment shall resolve two targets at 1 NM separation and for 5NM separation the equipment shall resolve two targets at 3NM, both to a probability of 95% or greater throughout the required azimuth and range as defined in the OR (456).

Section 12 of CAP 670 defines the radar coverage requirement:

12.1.2 The radars shall have a theoretical coverage, in the areas of the OR, which corresponds to 80% detection of the returns from a 1m2 target. This increases to 90% for areas providing SRA procedures. For primary targets this theoretical cover shall assume Swerling case 1 targets (458).


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4.1.2.1.1 Primary radar bands CAP 670 also classifies the primary radar bands and their radar services:

Table 4.1-1

Frequency Wavelength Frequency band CAP 670 Radar Service 590 MHz to 598 MHz 50cm medium/long range radar services

1215 MHz to 1365 MHz 23cm L medium/long range radar services

2700 MHz to 3100 MHz 10cm S short/medium range radar services

9000 MHz to 9200 MHz and 9300 MHz to 9500 MHz

3cm X short range radar services

15.4GHz to 15.7GHz 2cm Ka GMR, very short range radar services

34.5GHz to 35.5GHz 1cm Ku ASMI, very short range radar services

4.1.2.2 Frequency band comparisons In order to evaluate L and S Band radar performance and enable comparisons with other higher frequency radar bands on a ‘generic’ basis an Excel spreadsheet was developed.

4.1.2.2.1 Basis for spreadsheet calculations A common form of the ‘radar range equation’ using the signal to noise ratio can be expressed as:

LNFBTkR

FFGGPNS

ns

rtrtt43

222

)4( π

λσ= Equation 4.1-1

Where: is the transmit power, and are the antenna transmit and receive gains, tP tG rGσ is the target radar cross section, λ is the wavelength, and are the pattern propagation factors for transmit and receive, which account for the antenna patterns and non-free space wave propagation.

tF rF

R is the range to the target, k is Boltzmann’s constant, is the system noise temperature, is the noise bandwidth, usually matched to the transmitted pulse, and is the sum of system losses.

sT nBL

The basic spreadsheet uses a simplified, frequency independent, version of the radar range equation for the civil ATC radar calculations. This estimates the signal to noise ratio of the return from a target at a given range according to:

sns

rtt

LNFBTkRAGP

NS

42)4( πσ

= Equation 4.1-2


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Where the antenna effective aperture is:

πλ

4

2r

rG

A = Equation 4.1-3

To estimate the transmitted RF power required to achieve a specified signal to noise ratio, the above equations can be expressed as:

( )rt

snst AG

LNFBTkRSNRP

σπ 424.

= Equation 4.1-4

using the same parameters as before, with as the antenna effective receive area, and with a 50% antenna efficiency assumed.

rAR is fixed at 111km and σ is 1 square metre for

the CAP670 requirement.

Specific radar systems have different architectures, which are not straightforward to represent for generalised comparisons. So for this comparison as many radar parameters as possible were kept constant for the different radar frequency bands (although this has required some assumptions to be adopted), and we have assumed a common system architecture. These assumptions and other factors are reflected in the discussion of the results below.

A simple rectangular antenna was assumed, with width and height based on typical narrow beamwidths (1.25°), except where specific values were identified (L-band and S-band radars). Narrower beamwidths, and thus increased antenna gain, are one approach to increasing detection range for a given transmit power. However narrower beamwidths require more time to provide surveillance of the same volume of coverage, and CAP 670 also places requirements on update rates for different radar roles. These would not normally be achieved with significantly narrower beamwidths without associated changes in radar system architectures, which would have additional cost implications and so are not considered further.

Atmospheric and weather effects on radar systems’ performance are well established, and are supported by a large body of published work. In general, atmospheric attenuation of radio waves in longer range radar systems is not normally an issue below about 1GHz, it is relatively modest at L-band and S-band, and it starts to become significant at C-band and above. At some specific frequencies the atmospheric attenuation becomes more severe due to molecular resonance, ~22GHz for water and 60GHz for oxygen. At 60GHz this effect is very severe and makes anything other than short range radar impractical. This feature can be used to provide short range covert communications, and systems where frequency reuse after a few kilometres of separation can be utilised, such as vehicle telematics.

Weather attenuation due to rainfall, snowfall, etc., can also significantly affect radar performance. Long range radar uses the lower frequency bands, such as L-band and S-band, which are less affected by weather than the higher frequencies. The weather effects occur for only a small percentage of the time, but when they do, the effects can be significant. The effects with increasing rainfall rates are broadly linear with increasing rainfall and increasing frequency, with some limiting at the higher frequencies, above X-band. The


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rain attenuation can be significant with modest rainfall rates, where the region affected by rainfall extends over large distances. The spreadsheet includes atmospheric attenuation and some limited rainfall losses. A 20km range extent of 3mm/hr rainfall (moderate rain) was assumed for all radar frequency bands. Table 4.1-2 shows the atmospheric attenuation coefficients for the clear atmosphere and for 3mm/hr and 10mm/hr rainfall for comparison, with figures calculated using ITU-R methods [6] and [7]. These are very similar to the values shown in Barton [8].

Table 4.1-2 : Attenuation coefficients (2-way) for radar bands

Radar Band L S C X Ku K Ka V W

Frequency (GHz) 1.3 3 5.5 10 15 22 35 60 95

Atmospheric attenuation coefficient (dB/km)

0.012 0.014 0.017 0.027 0.056 0.36 0.19 30.0 0.78

Rainfall attenuation coefficient, 3mm/hr, (dB/km)

0.0003 0.002 0.01 0.07 0.23 0.54 1.4 3.2 4.7

Rainfall attenuation coefficient, 10mm/hr, (dB/km)

0.001 0.007 0.04 0.33 0.90 1.9 4.3 8.6 11.6

Weather patterns in the UK indicate that rainfall rates, durations and occurrence vary across the region [9]. The effects of rainfall on radar performance should be considered for each radar frequency band, as rainfall rates of 10mm/hr are not uncommon, and 100mm/hr has been recorded occasionally [10].

Backscatter from rain drops also affects the radar in the form of clutter, which can also reduce detection performance. In general the higher the operating frequency, the greater the effect on the radar.

4.1.2.2.2 Clutter variation with higher frequencies Land, sea and weather clutter characteristics are complex functions of operating frequency, polarisation, grazing angle, clutter type and other parameters. At present there are no universally accepted models of clutter, although there are various models in general use [8] and [11]. In characterising clutter the following aspects must be considered:

Radar cross-section

Temporal statistics and correlation

Spatial statistics and correlation

Spatial extent

The radar cross-section defines the level of the clutter signal as seen at the radar. In all cases, this is a function of radio frequency, polarisation, clutter type and grazing angle. In many cases, the variation due to frequency is largely due to propagation effects, particularly at low grazing angles. In general there is a linear relationship between clutter RCS and wavelength. At low grazing angles (below about 1°) propagation effects, such as multipath


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and screening, make this relationship non-linear. However, as stated above, there is no accepted model for this behaviour. In fact, it will be highly scenario specific.

Skolnik [12] provides a simple model for rain clutter, which shows that rain reflectivity varies as the fourth power of radio frequency:

126.14 107 −⋅⋅⋅= rRFGHzη Equation 4.1-5

where η is the rain reflectivity in m2/m3

RF is the radio frequency in GHz

r is the rainfall rate in mm/hr

The total radar cross-section from rain clutter is the reflectivity (see above) multiplied by the volume of the radar resolution cell which encompasses the clutter.

The implication of the equation above is that radar sensitivity to wanted targets in the presence of rain is reduced as frequency increases. Sophisticated radar processing is usually performed to counter clutter, but ultimately detection thresholds are increased and sensitivity reduces.

There has been ongoing research into the effect of frequency on clutter statistics, e.g. K-distribution statistics for modelling temporal and spatial statistics of sea (and land) clutter reflectivity.

In general surface clutter spectral spread increases with increasing radio frequency, indicating that detection of small low-speed, targets will be compromised at high frequencies. Rain clutter spectral spread is due to a combination of factors, including wind shear, differential drop fall rate etc., but will generally increase with higher frequencies also.

4.1.2.2.3 Other RF losses The spreadsheet uses estimated system noise figures and other losses based on general radar engineering experience, with proportionately higher losses at higher frequencies.

In summary the basic CAP 670 requirement for an ATC radar system is the capability to detect a 1 m2 target at 111km range, with an 80% single look probability of detection and a false alarm rate of 10-6. This probability of detection requires a signal to noise ratio of about 18 dB, assuming a Swerling 1 type target. The spreadsheet calculations estimate the output power required by the radar to achieve this under typical noise limited operating conditions, and provide additional calculations on field and signal strengths to inform other work packages. The spreadsheet data has used several assumptions, and thus the results are designed to provide comparative information rather than precise values for a given frequency band.

The results in the spreadsheet clearly indicate that for a given range, the higher frequency radar bands will suffer greater atmospheric path losses. Higher frequencies also suffer increased system losses. In part all these losses could be compensated with increased output power. This in turn requires more prime power, and higher frequencies are typically


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less efficient at converting prime power into transmitted power, so this aspect of the operating cost will be greater. However, increasing the transmitted power also increases the reflected energy from ground and weather clutter at the radar receiver, and thus may increase the dynamic range requirements and demand more complex processing techniques to remove or cancel.

4.1.2.2.4 Receiver dynamic range Receiver dynamic range is primarily driven by detection criteria determined from the radar cross section of wanted targets at long range and ground clutter in the main beam at short range.

The dynamic range of a receiver can be defined as the larger of two ratios: the ratio of the received signal power that will saturate the receiver to the minimum detectable signal [16]. In ideal situations, the minimum detectable signal is determined by the mean-square noise level. However in cluttered situations the minimum detectable signal is often determined by the clutter levels. Both ground clutter returns and weather clutter increase with increasing frequency, so raising the minimum detectable signal, i.e. reducing detection performance. Thus moving ATC radars to higher frequency bands will require more demanding receivers in terms of their dynamic range. In digital signal processing radar, the total system dynamic range is typically determined by the number of A/D bits used to sample the receiver outputs. In older systems this was usually 8-bits, with technology improvements, more modern radar are beginning to utilise higher performance A/D converters using 12 or 14 bits.

4.1.2.3 Lower frequencies This section of the study is tasked with examining issues of moving radars to higher frequencies. Historically however, radar has also operated at lower frequencies and 590MHz to 598MHz is still allocated in the UK to the aeronautical radio-navigation service on a primary basis. This is also known as the 50cm band, and is now confined to TV Channel 36. However new assignments are subject to co-ordination with a number of neighbouring countries: Belgium, Denmark, Germany, Ireland, Luxembourg, Morocco, Norway, Spain and the Netherlands. Based on the assessment in section 4.1.1 it seems unlikely that the 8MHz spectrum available at 590MHz would suffice for more than a few radars from the 132 ATC radars in current UK use.

However, it is of interest to note that the planned phasing-out of analogue TV in favour of digital television will produce a ‘digital dividend’ of spare channels to be released for other purposes. The six channels to be released (in the UK at least) are Nos. 33, 34, 35, and 37, 39 and 40 - i.e. channels which lie either side of the Channel 36 radar allocation. This raises the interesting possibility of wider use of the 50cm band for ATC coverage at some future date.

Lower frequencies are much less affected by atmosphere and weather losses, but require larger antennas to provide the 1.25° or 1.4° azimuth beamwidth. A 1.4° azimuth beamwidth antenna at 600MHz would be ~25m wide. This antenna would therefore be considerably larger, heavier and create more windage, requiring more energy to rotate, than current L and S-band ATC radars.


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High power from the radar could affect nearby TV receivers, which are probably relatively wideband and cover Channel 36. In order to ameliorate this, the TV receivers might need limiters to be fitted. Nevertheless, the basic problem remains that the limited amount of spectrum available at lower frequencies would only be able to support a small fraction of the required airspace coverage. Thus radar solutions using these frequencies are unlikely to be either economic or feasible under these conditions.

4.1.2.4 Radar band results The radar detection calculations are based on several assumptions, and so the results depend upon what parameters have been assumed to be constant between the different frequency ATC radars. The basic CAP 670 detection requirement can be accomplished at different frequencies by making various choices of peak power and pulse length, see Figure 4.1-1. Mean power will be significantly less than the peak power shown, depending upon the duty ratio, which is typically a few percent, but will still require the generation of many kilowatts of RF power. Again it should be stressed that these results should be used to illustrate trends rather than to provide exact values.

CAP670 RF Output Power

1.E+01

1.E+02

1.E+03

1.E+04

1.E+05

1.E+06

1.E+07

1.E+08

1.E+09

0 10 20 30 40 50 60 70 80 90 10

Frequency GHz

Peak

RF

Out

put P

ower

kW

0

No Atmosphere Clear Air Attenuation 3mm/hr rain

Figure 4.1-1 : Estimated required peak RF output power

In general the higher the operating frequency the higher the required transmitted power. Also, due to lower power conversion efficiencies at the higher frequencies, the prime input power, and thus operating cost, would be significantly higher at the higher frequencies. This efficiency difference should become less pronounced when newer semiconductor technologies, such as Gallium Nitride (GaN), become available.

The results confirm what is already well known, that the lower-bands (L and S) require less output power to achieve the CAP 670 detection criteria. This appears to be primarily due to the much larger receive antenna area for a given angular resolution, about 1.25° or 1.4°. This in turn is dictated by the accuracy resolution and update rate requirements.


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It is also apparent that the power requirements increase disproportionately above about 10GHz: this is due to the atmospheric and rainfall attenuation losses. Thus detailed analysis will be required for radar systems intending to use this frequency range, where these ‘environment’ losses can be extremely severe.

4.1.2.5 Safety aspects The RF radiation hazard (RADHAZ) of microwaves to humans is a well known safety concern. Current microwave radiation safety levels are recommended, with power densities of 50W m-2 for occupational users, and with 10W m-2 for the general public, averaged over an interval of a few minutes 1, see ICNIRP guidelines [13]. Most modern airports have an ATC radar, often within the airport boundary, and so general public safety levels apply. ATC radars are also located in rural areas, where again general public safety levels apply. In practical situations there is likely to be an inherent exclusion zone of several hundred meters around the ATC radars when they are operating, due to their location.

In view of the significant powers required at some frequencies a check was made that the estimated radar radiation levels would be satisfactory; both main beam and sidelobe power levels were examined for all frequency bands.

4.1.2.5.1 ATC main beam power density The main beam RF power level is calculated as power density:

24 rGPS tt

π=

Equation 4.1-6

, while the electric field strength, E , due to the radar pulses, is expressed as:

SE 377= Equation 4.1-7

Estimated peak field strengths at 1km range from the radar were calculated for all radar bands, see Figure 4.1-2; the results include atmospheric effects (the calculated value for 60GHz is 2.6 E+87 Vm-1 and is not shown).

1 Power densities are to be averaged over any 20cm2 of exposed area and any )68( 05.1f minute

period, where is in GHz, to compensate for progressively shorter penetration depth as the frequency increases.

f


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Peak Field Strength at 1km

1.E+01

1.E+02

1.E+03

1.E+04

1.E+05

1.E+06

1.E+07

1.E+08

0 10 20 30 40 50 60 70 80 90 10

Frequency GHz

Fiel

d st

reng

th v

olts

/met

re

0

Figure 4.1-2 : Estimated Peak field strength at 1km

Mean power levels will depend on the radar system peak output power, duty ratio (1% assumed), beamwidth and dwell time, which is linked to antenna rotation rate (10 rpm assumed).

The spreadsheet was used to estimate the conditions where the 10W m-2 average power level for the general public would be approached. For the radar systems considered here, including atmospheric losses, this average power limit would be exceeded by a 22GHz (K-band) radar at ~1km range, by a 15GHz (Ku-band) radar at ~500m range and by a 5GHz (C-band) radar at >100m range. Estimated mean power levels at 100m range are shown in Figure 4.1-3. A safety zone around a radar of 100+m radius seems relatively practical for the general public, and so mean power levels are unlikely to pose problems for systems up to C-band. This calculation should be undertaken for the actual radar system being considered. Increasing the duty ratio would increase the average power a little, but in most cases the radar beamshape would provide an element of safety, as only the lower portion of the main beam would approach the general public at ground level. Again safety zones need to be carefully considered for all radar frequencies, and particularly the higher frequency systems, where larger safety zones might not be so straightforward to provide at locations such as airports. Whilst safety aspects for those inside airport buildings can be solved by suitable building design and materials, there may be many people involved in working around civil airliners immediately outside airport buildings.

4.1.2.5.2 ATC radar sidelobe power The RF power from the ATC radar sidelobes must be regarded as a continuous RF hazard while the ATC radar is rotating (operating), even though the main beam is surveilling successive portions of the airspace. The actual radar sidelobe levels will be dependent on the antenna design. ATC radars tend to use front fed reflector antennas with a tapered aperture illumination. This would typically provide a cosec2 amplitude pattern in elevation and low azimuth sidelobe levels.


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In order to examine the sidelobe power aspect, it is suggested that average power densities of -25dB (compared to the main beam) be used for calculation purposes. This is a pessimistic assumption, as the rear hemisphere of the ATC radar antenna (opposite to the transmit direction) is likely to have much lower sidelobe power, and so this would reduce the mean power when averaged over a full rotation. Assuming a 100m safety zone range, the sidelobe power calculation only approaches the general public safety level at 10GHz and higher frequencies, see Figure 4.1-3.

Mean Power levels at 100m

1.E-01

1.E+00

1.E+01

1.E+02

1.E+03

1.E+04

1.E+05

1.E+06

0 10 20 30 40 50 60 70 80 90 10

Frequency GHz

Mea

n Po

wer

Wm

-2

0

Mean Beam Sidelobes

Figure 4.1-3 : Estimated mean power levels at 100m

4.1.2.5.3 Civil aircraft Civil aircraft flying in the vicinity of modern ATC radars are likely to be illuminated by the radar main beam and so will be exposed to relatively high power RF radiation, but this is likely to be at ranges of several hundreds of metres or greater, and not the 100m used in the calculations above. ATC radars are continuously rotating so the mean power levels are unlikely to pose problems, and any civil aircraft is very unlikely to be near continuously illuminated.

High intensity radiated field (HIRF) testing is one form of radiated susceptibility testing required for the certification of critical function and essential function 2 avionic systems [14] on civil aircraft. The aim of this HIRF testing is to simulate the high peak and mean RF field environments typical of radar installations. For certification the avionic system must not sustain damage or exhibit degraded performance when subject to HIRF levels while

2 Critical Function: A function whose failure would prevent the continued safe flight and landing of the aircraft. Essential Function: A function whose failure would reduce the capability of the aircraft or the ability of the crew to cope with adverse operating conditions.


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operating. Different HIRF levels apply to different frequency bands. For L-band and S-band, the HIRF certification levels are 2,000Vm-1 to 3,000Vm-1 peak and 200Vm-1 mean. Similar levels apply to higher frequency bands up to 18GHz, and above this reduce to 600Vm-1 peak and 200Vm-1 mean. No HIRF values are listed for frequencies above 40GHz, but the values for 40GHz are assumed to apply to all higher frequencies in the following graphs.

Table 4.1-3 : HIRF Certification Field Strengths

Frequency, GHz Peak Field Strength, vm-1

Mean Field Strength, vm-1

1 – 2 2000 200

2 – 6 3000 200

6 – 8 1000 200

8 -12 3000 300

12 – 18 2000 200

18 – 40 600 200

These HIRF certification levels have been added to Figure 4.1-2 for comparison, see Figure 4.1-4.

Peak Field Strength at 1km

1.E+01

1.E+02

1.E+03

1.E+04

1.E+05

1.E+06

1.E+07

0 10 20 30 40 50 60 70 80 90

Frequency GHz

Fiel

d st

reng

th v

olts

/met

re

100

ATC Radar peak HIRF mean HIRF

Figure 4.1-4 : Estimated Peak Field strength at 1km

4.1.2.6 HIRF comparisons The results indicate that the existing L-band and S-band radars are unlikely to create field strengths at ranges of 1km from the radar that exceed the HIRF mean levels, and they are well within the HIRF peak field strengths, and so provide a safety factor. However with the higher frequency radar systems, above about 14GHz, the peak field strengths that they produce at 1km range begin to exceed the HIRF peak values. These HIRF values are used


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to certify modern aircraft electronics. Thus, detailed analysis would be required for specific radar systems intending to operate at these higher frequencies, to ensure that the field strength they produce would not exceed the HIRF values, with a suitable safety factor. Electronics systems in older aircraft using UK airspace may not have been tested to the current HIRF limits, or even for lower field strengths of 300Vm-1 peak.

This comparison of the required radiated field-strength with the HIRF limits was repeated for 500m and 100m ranges, and the results are shown in Figure 4.1-5 and Figure 4.1-6.

Peak Field Strength at 500m

1.E+01

1.E+02

1.E+03

1.E+04

1.E+05

1.E+06

1.E+07

0 10 20 30 40 50 60 70 80 90

Frequency GHz

Fiel

d st

reng

th v

olts

/met

re

100


Figure 4.1-5 : Estimated Peak Field strength at 500m


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Peak Field Strength at 100m

1.E+01

1.E+02

1.E+03

1.E+04

1.E+05

1.E+06

1.E+07

1.E+08

0 10 20 30 40 50 60 70 80 90

Frequency GHz

Fiel

d st

reng

th v

olts

/met

re

100


Figure 4.1-6 : Estimated Peak Field strength at 100m

This suggests that at 100m only the L-band and S-band ATC radars produce peak field strengths that are less than the HIRF field strengths.

Clearly there are various possible approaches to reducing peak radiated power, and thus field strength. For example the pulse duration could be increased, however, at the higher frequencies above C-band, RF power devices tend to be vacuum tube or magnetron based and can support only low duty (~1% - 5%) and thus short pulses at the required PRF. Alternatively multiple beam systems could be used to extend the dwell time and thus reduce peak radiated power. However this would involve multiple receiver channels and a more complex radar system architecture that is different from the generic approach considered above. These alternative approaches may then require further resources to restore ATC radar performance.

4.1.3 Summary Many of the civil radar systems are located in Southern or Eastern England, with a high concentration in the Greater London area. In the London area for example, the average distance between neighbouring radars is only a few tens of kilometres, whereas in the Northern and Western parts of the country the separations are often measured in hundreds of kilometres. This uneven spatial distribution of radars means that in the South-East it is a challenging task to find adequate frequency allocations within the available spectrum, whilst in the North and West it is a much easier task. Both co- and adjacent-channel coordination criteria may be important when frequency planning for ATC radar.

The major benefits of improving radar spectral efficiency will come from widespread deployment of the technique (whichever approach is chosen), followed by re-planning of the radar frequency allocations to take advantage of the spectrum savings. Such re-planning could be used to re-position the radars into a smaller total bandwidth, thus freeing up a


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block of spectrum. Such a block of contiguous free spectrum will be much more useful to other systems than isolated pockets of frequencies perched upon the side-lobes of high-power radar transmissions.

The possibilities for moving ATC radars to higher frequency bands (where the demand for spectrum may perhaps be less intense) have been examined by studying the requirements for radar detection performance. The Air Traffic Services (ATS) Safety Requirements specify the ranges at which targets of a certain size must be detectable and distinguishable from neighbouring targets. These requirements in turn place constraints on the radar system parameters (for example antenna beamwidth, and transmitted waveform), and if a system is to be designed to operate at higher frequency bands, then this carries implications for the transmitter power that must be radiated.

On both device power and safety considerations it appears that it would be quite impractical to attempt to re-locate ATC radars to frequency bands above about 10 GHz. At higher frequencies the transmitter powers required greatly exceed the capabilities of power devices that are available now, or are likely to be available in the foreseeable future. Furthermore, even if such devices were available, then the radiated fields would be so strong that at short ranges (e.g. at 1km from the radar) there would be severe hazards to both personnel and to the electronics on board aircraft.

It is thus concluded that, of the currently designated radar bands, the only frequency bands (on technical grounds) to which ATC radars could possibly be moved would be C-band (5 GHz) and X-band (10 GHz). However, these bands are themselves already relatively congested, and so may not offer significant opportunities for re-location of L- and S-band radars. The detailed questions of how much spare capacity there may be available in these bands, the costs and disruption involved in moving bands, and the regulatory implications of doing so, are all topics, which must be addressed outside this study.


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4.2 Assessment of the economies of introducing new spectrally efficient radar technologies The objective of this work package is to examine the economics of introducing new radar technologies for the purposes of improving the spectrum efficiency of radar systems.

The economics of introducing selected candidate technologies from those discussed in the rest of this study are assessed in the context of the provision of Air Traffic Control radar services in both the L and S bands.

As previously discussed, there are three alternative approaches to potentially allowing other uses of the radar bands:

1. Relocate radar services to higher bands

2. Improve the spectral efficiency of radar systems and re-plan allocations to take advantage of this – thus freeing up contiguous spectrum in other parts of the band

3. Develop technology to allow radar and other services to share the bands without causing unacceptable interference to each other

The technological approaches assessed, which fall into one or more of these categories, include:

New pulsed radar designed for a new operating frequency band

Modification/upgrade of existing (S-band) TWT radar to reduce spectrum requirements

High power transmitter / drive upgrade to allow more spectrally efficient waveforms

Ultra narrow band radar

Before evaluating the economic costs of these technological approaches, it is useful to consider the economic benefits to the user of new spectrally efficient radar systems. Typically the economic benefit of improving spectrum efficiency is based on the amount of spectrum liberated for other uses and the economic value of that spectrum, which the user may realise through spectrum trading or reduced license payments. The following sub-section thus outlines various methods of assessing the economic value of spectrum.

4.2.1 Alternative spectrum management and valuation methodologies The increasing demand for radio spectrum in the UK has meant that it is very important to manage spectrum allocation and pricing.

The Wireless Telephony (WT) Act 1998 requires the Secretary of state, in setting spectrum license fees to have regard in particular to various spectrum management factors:

Efficient spectrum use and management

Economic benefits


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Development of innovative services; and

Competition

This legislation makes sure that spectrum pricing is not used as a form of taxation by the government; it promotes efficient use of the radio spectrum to be transferred to and used by the user who values it most highly.

It promotes competition by increasing the availability of spectrum for use in the most valuable service and it facilitates economically valuable innovation as new users enter the market to offer new services

Some of the spectrum does not have any likely spectrum value and there are international constraints on a lot of bands

As a whole the value of radio to the economy exceeds £24 billion per annum (in 2002)[17], about 2.5% of the UK GDP (Gross Domestic Product).

4.2.1.1 Spectrum trading Ofcom is currently introducing limited spectrum trading, this permits a more liberal use of spectrum by allowing change of ownership and use where bidders can sell and buy spectrum at anytime. Ofcom forecasts that 72% of spectrum will be traded in this way by 2010.

Ofcom proposes gradual implementation of spectrum trading which began in 2004 and will continue until expected full implementation in 2007. There is also an agreement for Aviation and Maritime comminations to use trading by 2007. For radar trading is proposed to be introduced between 2007 and 2009

4.2.1.2 Literature survey A literature survey was carried out in order to understand the current thoughts and techniques used to price spectrum and to also explore differences in regions across the globe. The following lists some sources of information.

Institute of Electrical Engineers (IEE)

Ofcom

Various Radiocommunication Agencies

CEPT (European Conference of Postal and Telecommunications Administrations)

Warwick University

Other Economics websites


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This survey involved examining currently used and proposed techniques and methods for calculating spectrum pricing. Intellectual reports, such as the Cave report [18] along with responses from various parties like C4 and ITV etc. to those reports and the conclusions they drew from them were investigated.

A number of techniques were examined:

Auctions

AIP (Administered Incentive Pricing)

Smith-Nera (1996)

Diamond Mirrlees (1971)

4.2.1.3 Auctions Auctions have been used in the UK and abroad to determine the value and allocation of Wireless Telegraphy licenses. The concept is that operators who value the spectrum most will pay more and use it more effectively and innovatively.

4.2.1.3.1 Examples 3G: March 2000: April 2000 23 Billion

Wireless Broad band1: October 2000

Wireless Broad band2: June 2003 6.6 Million

4.2.1.3.2 Advantages Auctions are thought to be economically efficient. This is because the very nature of the auction means those operators who value it most (and can get most value from it) will pay only what it is worth to them. This does depend on robust business modelling by the operators.

Auctions are open to anyone, which allows new entrants into the spectrum market, adding new ideas and approaches to the market place.

4.2.1.3.3 Disadvantages Auctions are not suitable for high-volume, low value licenses, as the preparation for an auction can be extensive. A lot of time is used to advertise and gain awareness of the auction and this is thought not to be an effective use of an auction for low revenue licenses. Furthermore to make an auction accessible, memorandums have to be issued with information about the auction. A website may be created for information distribution. Bidders may also be prepared for the auction by means of training and/or mock auctions.

Auctions tend to increase the costs to operators who will need to recover the cost from users. Higher cost services will reduce the number of users, and as a result could reduce


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tax revenues for the government. An example UMTS is an emerging technology and service, the success of which is not totally assured. Auctions do not ensure the presence of the most competent operators, but merely the operators with the largest disposable cash.

Auctions introduce an additional and unquantified uncertainty for operators and service providers.

The uncertainties caused by auctions could reduce the incentive of investors to invest in risky new technologies, which may not get the spectrum needed to assure a reasonable prospect for successful deployment. This could in turn lead to suppression of innovation.

The government has indicated that auctions will be used selectively in the UK for new national or regional services, where there are more applicants for the spectrum than can be accommodated.

4.2.1.4 Administered Incentive Pricing (AIP) AIP is an important mechanism for promoting efficient spectrum management. AIP does this by showing to the user the opportunity cost of using the spectrum. It is a spectrum structure that is based on value compared with alternatives. AIP was pioneered by the then RA (Radio-communications Agency) and introduced into the Wireless Telegraphy act in 1998.

AIP is used to influence the choices made by the users, to make existing users reflect on their spectrum needs and release any surplus spectrum. Likewise communications users may re-evaluate their medium options and decide that a hardwired rather than radio medium would be preferable. AIP allows new entrants to have a greater chance of gaining access to the spectrum if their use has a potentially higher value.

AIP is a spectrum structure that is based on value compared with alternatives. The price should influence the usage of the spectrum and is targeted at economic efficiency. In the absence of market-determined prices for spectrum, AIP are a surrogate for missing markets. AIP helps to encourage efficient use of the spectrum by providing a financial incentive for the Wireless Telegraphy act licence holders to utilise their spectrum allocation fully.

4.2.1.4.1 Applied AIP AIP has been applied to the following services:

Fixed Wireless Services - Point-to-Point Microwave Links

Programme Making and Special Events

Business Radio

Television and Radio Broadcast:

2G mobile telecommunications


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4.2.1.4.2 AIP Mechanisms AIP is aimed at economic efficiency. Economic Efficiency has three aspects to it. They are the following:

Allocative Efficiency : An allocation of inputs that maximises the value of goods and services produced such that no other allocation can increase the well being of one economic agent without harming that of another.

Productive Efficiency : an allocation of inputs in the production of goods and services that produces a given level of output at the lowest possible cost.

Dynamic Efficiency : inputs are allocated to the production of goods and services over time so that productive and allocative efficiency are maintained in response to changes in technology or consumer preferences.

The Cave Report [18] Para. 7, recommends that productive efficiency be used, indeed Diamond-Mirrlees [19] states that AIP strictly aims at productive efficiency.

4.2.1.4.3 Smith- NERA method This method was created by Smith-NERA in 1996 and is used by Ofcom for calculating opportunity costs of the spectrum, making use of ‘the least-cost alternative’, it should result in Productive Efficiency. The method calculates the willingness to pay for a marginal unit of spectrum. In the case of a mobile phone operator, a marginal unit of spectrum may be 2 x 2.4MHz. The least cost alternative for a mobile phone operator is the lowest cost way to meet current customer demand at the current quality level by adding and subtracting marginal units of spectrum from the operators’ current spectrum holding. This cost represents the marginal value of the spectrum to the operator.

Example method: For this example there are three spectrum uses, and three different frequency bands. The three uses are named U1, U2, U3 and the three frequency bands are F1, F2, F3. Management has allocated Use U1 to band F1, U2 to F2 and U3 to F3.

Assuming that U1 and U2 are equivalent to cellular and fixed link and that band F3’s frequency is below 2GHz and F4 above 2 GHz.

Suppose company X in market M which uses spectrum S and another input, I (maybe service content/quality) produces output, O (Service). The firm’s output can be expressed as the following function:

OXM = f (IXM, SX

M)

Company X wants to maximise profit and minimise costs. If unit of spectrum, ∆S is added to or taken away from SX

M, (which is the companies whole owned spectrum) a compensating change in the input IXM (service content/quality) could be made in order that there is no change at the output (service), OX

M. This shows the rate of substitution between ∆S and ∆I (Change in spectrum and service content/quality). For a change in spectrum size of ∆S = 1


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there would be a change in service content/quality, ∆I implied, where ∆I is multiplied by it’s price (company investment input). This gives a money representation of the rate of substitution. By using this method in other spectrum sectors comparisons can be made between different spectrum sectors using this common money number.

Table 4.2-1 represents a hypothetical Smith-Nera least cost alternative method. The values in the cells are calculated as in the proceeding paragraph. So, 500 in use U1, frequency F1 is the value, expressed in money terms using the least-cost-alternative input, of the marginal unit of spectrum. For example a unit of spectrum may be worth four base stations which each have a price of 125. Values in the other cells are also a marginal unit of spectrum. If U1 was a mobile phone network, F2 in U1 may have certain characteristics that mean that data rate or call quality are degraded, causing it to have lower marginal value. It can be seen that this table does not satisfy productive efficiency, as the unit values are not equal across the spectrum.

Table 4.2-1 : Productive inefficiency

Frequency Bands

Uses F1 F2 F3

U1 500 375 0

U2 175 300 150

U3 50 50 75

It can be seen from this table that market values are not even across the spectrum. It can be seen that use, U1 has a high marginal spectrum value in Band F1. Frequency band F2 is a bad substitution for F1 in band U1. For use U2, band F2 has the highest marginal value for U2. Because U1 has a higher marginal spectrum value, it would be beneficial to release some of band F2’s spectrum and hand it over to use U1. In this way the marginal unit of frequency band, F2 applied to U1 could produce the same output in use U1, whilst freeing up resources to compensate U2.

It can also be seen that spectrum in band F3 used for U3 could be reallocated to use U1.

Table 4.2-2 shows the new marginal values after reallocation of spectrum.

Table 4.2-2 : Marginal values after reallocation

Frequency Bands

Uses F1 F2 F3

U1 435 320 0

U2 160 320 125

U3 65 60 125

4.2.1.4.4 AIP and productive efficiency: Consider Frequency band F1. Suppose there are four users U1, to U4 occupying frequency band F1, with marginal values (MV) for frequency band F1 as follows:

MVU1 = 600, MVU2 = 550, MVU3 = 500, MVU4 = 450


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With AIP = 500 for example, user U4’s market value is less so can afford to release some spectrum, which would then be offered to U1 and U2 as their market values are greater and so to use the extra spectrum would be beneficial – over time adjustments will occur such that MVUA = 500 for all users UA

4.2.1.5 Potential users / Markets of S-band spectrum The S-band (2700 – 2900MHz) is primarily allocated to aeronautical radionavigation (radar) services and is subject to international agreements on its’ use. These agreements limit this band to aeronautical radionavigation use, currently no other uses are allowed. This means that at the moment opportunity costs for this band are zero.

It is not in the scope of this report to calculate spectrum value in terms of £/MHz for S-band spectrum released through radar spectrum efficiency improvements. However, should future developments introduce more flexibility into the allocation of spectrum in the S-band, it is pertinent to consider the most important alternative applications this spectrum could be used for, thereby throwing light on the economic and commercial pressures.

4.2.1.5.1 3rd generation radiomobile communication services (3G / UMTS / IMT2000) The radiocommunications industry is hoping for a decision on an extension band for 3G operations by January 2008. Currently the 2600MHz band is being considered by CEPT’s division, the ECC* (Electronic Communications Committee). The 2500 to 2690MHz band is also wanted to secure the long term success of the terrestrial UMTS and IMT-2000 technologies.

According the UMTS forum (www.umts-forum.org/) as of January 2006 there were 50 Million 3G/UMTS subscribers worldwide and this will continue to rise through the migration of 2G and 2.5G users to 3G. This will mean there will be a need for more spectrum for the increased capacity. The work for the global harmonisation of the extension band started in 1997.

Should further expansion be needed for terrestrial UMTS/IMT-2000 and possibly satellite versions too, then part of the S-Band (2700 – 2900MHz) could be considered. World harmonisation is very important for this technology to ensure manufacturer costs are lowered and interoperability for handsets worldwide.

4.2.2 The predicted costs of putting new technologies into a band Since radar often need to meet Safety of Life requirements, the process of putting new radar technology into service can be both expensive and time consuming. This section outlines the typical stages of organisation and development required to replace or upgrade ATC TMA radar. As a manufacturer of radar systems, BAE Systems Insyte is also able to give an indication of price & timescales for the development and production of some of the new technologies for improving radar spectral efficiency detailed elsewhere in this report.

Assuming an ‘Off the shelf’ solution, the process of replacing or upgrading a radar can be roughly divided into three stages which cover approximately 20 months from start to the radar entering service and require at least 5 people from the customer / user community.


http://www.umts-forum.org/

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Specification & Procurement Installation Acceptance, Hand over & Qualification

9 - 12 months 4 - 6 weeks 11 - 12 months

However, if significant technology development is required, as would most likely be the case for spectrally efficient radar technology, the timescales and costs would increase:

Specification & Procurement Development, production

and installation Acceptance, Hand over &

Qualification

12 - 15 months 18 – 60 months 11 - 12 months

These different stages are outlined below:

4.2.2.1 Specification and procurement Once a need has been identified for new or upgraded radar; the customer (in this case either NATS or a regional airport) need to draw up detailed specifications and begin communication with potential bidders. The bid phase needs to be managed from the customer side, while the subsequent contracting and procurement phase can be managed in cooperation with the successful bidder. This whole process typically takes 9 - 15 months depending on the scope and complexity of the equipment (‘off the shelf’ or new technology)

4.2.2.2 Development, production and installation All of the technological solutions for improving radar spectral efficiency detailed elsewhere in this report involve the development of new radar technology. As a manufacturer of radar systems, BAE Systems Insyte is able to give an indication of price & timescales for the development and production of these technologies. It is understood, however, that in the context of this report, these prices are not capable of acceptance by any potential customers; who would need to define specific requirements and undertake a formal bid process.

These prices are based on historical bids, current bids and expert knowledge. The production prices are based on one off production.

The technological options selected for this costing process are:

1. Receiver/signal processor modification (see section 4.3.4.1.1)

2. Transmitter based modification (see section 4.3.4.2)

3. New design ATC radar using conventional technology (see section 4.1)

4. New design ATC radar using continuous wave (CW) technology (see sections 4.6, 4.7, 4.8)

The prices presented are an indication of typical development and production prices that could be expected from a European supplier of surveillance radar.


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Table 4.2-3 : Typical development and production prices of spectrally efficient radar technology

Solution Description Development Production 1 Receiver/Signal Processor Modification £ 2.80 M £ 0.72 M

2 Transmitter Modification £ 2.30 M £ 1.30 M

3 New Design ATC Radar £ 14.00 M £ 2.50 M

4 New design (CW) ATC radar - -

Due to the immaturity of CW technology, solution 4, Insyte are unable to advise meaningful development and production prices and have therefore been unable to price against this solution.

The sections below set out the typical costs and timescales associated with the implementation of the various potential spectrum efficiency solutions discussed above. The information given here is advisory and represents the sort of figures that might be expected from European suppliers in the surveillance radar business if requirements for such solutions were placed on the market. The figures have been arrived at on the assumption that the equipments involved would be purchased on a “one-off” or small quantity basis, and may be considerably bettered if purchasing in large quantities. Indeed, it should be noted that in the case of solutions requiring large development programmes, suppliers may not be prepared to undertake such development without a commitment to substantial production runs over which the development costs could be amortised.

4.2.2.2.1 Receiver / Signal Processor based modification This solution requires the design and development of a set of hardware and software modules to form a revised receiver chain that can optimise performance over a reduced transmitted bandwidth as described in section 4.3.4.1.1. It is likely that variant(s) of this design would need to be established to cope with the differences in form and interfaces should it be necessary to fit this modification to different models of radar. The figures in this section apply to the development of a receiver / signal processor modification for a single model of radar only.

In estimating an industry typical development cost for such a receiver / signal processor upgrade package we have considered that the manufacturer would need to address:

Requirements analysis and management, sub-system specification.

Subsystem design- Rx, timing & control, SPU, BITE, Tx interface.

Prototype build and test of Rx, SPU, control elements.

RMA analysis, spares ranging, technical documentation.

Sub-system and modified system level design proving.

Performance modelling and analysis.

Table 4.2-4 : Costs and development times for Receiver and SPU modifications


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Development timescale: 18 months

Development cost: £2.8M

Unit Production Cost (UPC): £720k each

Procurement lead time (of developed product): 12 months

Setting to Work timescale: 6 weeks

4.2.2.2.2 Transmitter based modification This solution requires the design and development of a linear amplifier that will allow the adoption of complex spectrally efficient drive waveforms (see section 4.3.4.2). The transmitter would be required to be designed to be a functional replacement for those currently used in a specific radar – but be capable of being driven by a complex rather than rectangular pulse. It is likely that variant(s) of this design would need to be established to cope with the differences in form and interfaces should it be necessary to fit this modification to different models of radar. The figures in this section apply to the development of a replacement transmitter for a single model of radar only.

In estimating an industry typical development cost for such transmitter we have considered that the manufacturer would need to address:

Requirements analysis and management, sub-system specification.

High power sub-system design-Tx, modulator, rf components.

Low power subsystem design- timing & control, I/F to waveform generation, BITE.

Prototype build and test of Tx, control elements.


Sub-system design proving.

Performance modelling.


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Table 4.2-5 : Costs and development times for transmitter based upgrade


Development cost: £2.3M

Unit Production Cost (UPC) £1.3M each (assuming dual channel equipment)


Setting to Work timescale: 4 weeks

4.2.2.2.3 New design of ATC radar (‘conventional’ technology) This solution requires the design and development of a new generation of ATC radar, operating in a ‘new’ or previously restricted part of an ‘old’ band (see section 4.1). Whilst the design would be based on conventional (ie. microwave, pulse) techniques it would be optimised for spectral efficiency.

In estimating an industry typical development cost for such a new radar system we have considered that the manufacturer would need to address:

Requirements analysis and management, system specification, system level design.

High power sub-system design-Tx, antenna, modulator, rf components, tuning gear.

Low power subsystem design- Rx, waveform generation, timing & control, I/F to SPU, BITE, data extraction.

Prototype build and test of antenna.

Prototype build and test of Tx, Rx, control elements.


Sub-system and system level design proving.

Performance modelling and trials analysis.

Table 4.2-6 : Costs and development times to design new ATC pulsed radar using conventional technology in an alternative band


Development cost: £14 M

Capital cost: £2.5M each (assuming dual channel equipment)


Setting to Work timescale: 6 weeks (excluding civil works)


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4.2.2.2.4 New design of ATC radar (‘CW’ technology) This solution requires the design and development of a new generation of ATC radar, based on the novel technique of continuous wave (CW) transmission (see sections 4.6, 4.7, 4.8).

It is not considered useful at this time, given the current state of the art, to attempt to put any costs to the development or procurement of an ATC radar based on CW technology. Whilst theoretically possible, the concept has yet to enter any technology demonstration phase. Any costs suggested for such a system, should it ever reach commercial viability, would therefore be purely speculative. It may be noted however that for sensor systems of this complexity, it generally takes at least ten years to go from proof of concept to the availability of a saleable product with a proven level of performance.

4.2.2.3 Acceptance, Hand over & Qualification In addition to the development and capital procurement of the solution equipment outlined above, each will require a degree of qualification and acceptance by the operating unit and by NATS/CAA. Where the solution is a modification that does not substantially alter the fundamental design and performance of an existing installation, the costs may not be especially significant, but where the radar is deemed to have been substantially changed or the radar is a wholly new installation (particularly of a new type) the qualification and safety approval process can be protracted.

Typically, for a new radar system, this process would take around 12 months assuming no significant problems. In terms of man power the acceptance process would require engineering support from both the customer and supplier.

4.2.3 Economic and programmatic considerations The main customers for radiolocation services in the UK are National Air Traffic Services (NATS), regional airports and the UK Ministry of Defence (MoD). Between them they operate in excess of 100 S-band pulsed radar of various configurations.

NATS are currently engaged in a wide reaching radar replacement programme as they replace their older systems with the current state-of-the-art Raytheon ASR-10SS solid state radar, these new radar will have a nominal lifetime of around 15 – 20 years unless life extension upgrades are procured. The radar deployed at regional airports are of various ages and configuration, each airport, however, manages its own radar service support with new procurements or life extension upgrades.

Meanwhile, the UK MoD has a large number of radar approaching the end of their useful life over the next 3 – 7 years. In order to maintain military ATC capability beyond this timescale new radar or life extension upgrades will need to be procured.

Figure 4.2-1 summarises these timescales along with indications of the likely opportunities for insertion of new spectrally efficient radar technologies.


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Linear SS Tx

available

Possible availability of CW, Passive or

Sharing radar

NATS buying class C Tx, ASR 10-SS

Mid-life upgrade opportunity

Phased replacement / life ext of ASR 10-SS

Civil ATC

0 – 5 years 5 – 10 years 10 – 15 years 15+ years

Military ATC

Replace / life ext Military ATC required

Figure 4.2-1 : Timescales of likely radar procurement and technology availability in the UK

Considering first, the UK MoD timescales. It is clear that the alternative potential spectrally efficient technologies such as Superresolution reduced bandwidth, CW or Statistically shared systems will not be near to maturity for ATC radar application within the next 7 years and would not be considered as part of military ATC capability provision at that time. However, new high power linear S-band pulsed radar transmitters could be developed and deployed (as part of a life extension upgrade to an existing radar or in a new radar design) within this timescale.

4.2.4 Conclusions If improved spectral efficiency is introduced as a requirement of MoD radar, then linear S-band pulsed radar technology would provide the best and lowest risk option at that time; alternatives to this strategy for the MoD would be:

Procurement of completely new radar systems operating in a different band (as described in section 4.1 and costed in section 4.2.2). This would be more expensive and entail more risk for both the MoD and manufacturer.

Life extension upgrade to current TWT Watchman radar including reduction of transmitted bandwidth from 2.5MHz to 1.0MHz (as described in section 4.3.4.1.1 and costed in 4.2.2.2.1). However, this solution would not result in such good spectral efficiency improvements as the linear transmitter option.

Considering

the current programme of NATS radar procurement of Raytheon ASR-10SS class C transmitter based systems, which provide close to the best spectral efficiency of current radar technologies,


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the nominal 15 – 20 year lifetime of these radar and

the considerable expense of radar replacement before the end-of-life.

It is clear that the first opportunity for deployment of alternative spectrally efficient radar technologies such as CW, Passive or Statistically shared systems is unlikely to before 15 – 20 years hence.

There are, however, likely to be a number of opportunities for early, mid or late life upgrades. The timescales for these are again consistent with the development and potential availability of linear S-band transmitter systems, which would provide considerable spectral efficiency improvements and are technologically and economically low risk compared to the alternative options.

The significant spectral efficiency improvements available through the adoption of linear solid state transmitter systems, their relatively low technological risk, the predicted development timescales and the expected costs mean that should spectral efficiency become requirements of ATC radar, then this technology is by far the best candidate for procurement into both Civil and Military ATC capability.


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4.3 A study into the practical problems of generating high power coded waveforms using travelling wave tube or alternative devices The objective of this work-package is to study the practical problems of generating high power coded waveforms using travelling wave tube or alternative devices. For the purposes of this study, the coded waveforms examined here will be selected with a view to improving the spectral efficiency of radar systems while at the same time maintaining operationally important radar performance requirements such as detection capability, range resolution and minimum range.

There are two types of coded waveforms that need to be considered as part of this study:

those which can be used to generate a radar pulse with the minimum RF bandwidth, subject to the minimum performance requirements

those which allow radars to operate with a waveform that minimises the interference to radars of a like type, thus improving spectral efficiency by reducing the frequency reuse distance. (This allows a network of radars to operate with fewer allocated frequencies)

The application of these waveforms will differ between radar technologies and specific applications. In the examples studied here the focus has been on S-band Terminal Manoeuvring Area (TMA) ATC radar.

In particular the objective is to study how coded waveforms, proposed for spectrally efficient radar systems in previous studies (for example [1] and [2]) may be implemented in practical radar transmitters. These studies have examined the spectral properties of a variety of waveform designs on the basis that the proposed waveforms will be transmitted into space without corruption and thus maintain their spectral properties. This may not be the case for practical radar transmitters.

In order for spectrally efficient coded waveforms to be effective; radar transmitters need to be capable of generating the waveforms with the following attributes :

an acceptable level of distortion;

an acceptable peak and mean power level;

an acceptable efficiency in terms of prime power usage;

an acceptable level of spurious or noise emissions;

an acceptable cost to the operator.

In practical radar systems the high powers required to maintain detection of small targets at long ranges means that only a limited variety of transmitter devices are suitable for radar applications. These radar transmitter devices have certain performance limitations that affect their ability to transmit uncorrupted coded waveforms and thus allow improved radar system spectral efficiency.


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A variety of radar transmitter devices are first reviewed in this work-package for their applicability to coded waveform transmission, new techniques and transmitter technologies are then explored for application to upgraded or new radar systems.

4.3.1 Review of existing transmitter techniques and their applicability to coded waveforms

4.3.1.1 Current radar systems Many current civil radar systems use either magnetron or travelling wave tube transmitter devices. These radar systems use rectangular or near-rectangular trapezoidal pulses because these are the most efficient way of delivering maximum energy on target during the pulse period. Using rectangular pulses allows high-power transmitters to be run into saturation without fear of distorting the pulse whilst providing the maximum power efficiency. These amplifiers run in saturation to allow wideband operation with minimal output power variation across the frequency band. Running the amplifier in saturation also reduces pulse droop, which is important to maintain sufficient energy delivery to targets at long ranges. In more modern civil radar, solid-state devices are being used; these are also run in saturation, which allows multiple amplifiers to be easily matched for summation of signals in amplitude and/or phase.

In order to achieve the required range resolution, a radar waveform must have sufficient bandwidth, as determined by the relationship:

rcBW

∆=

21

Equation 4.3-1

Where : BW = 3dB signal bandwidth (Hz)

c = speed of light (~3x108 m/s)

∆r = range resolution (m)

This ‘Resolution Bandwidth’ can be achieved in different ways:

Using short, unmodulated, pulses of length τ, where τ = 1/BW. In order to meet detection requirements sufficient energy must also be delivered to the targets so peak power is typically large.

Using longer, modulated, pulses where the bandwidth of the modulation is the same as the resolution bandwidth. Longer pulses allow proportionately lower peak powers for the required energy on target. However, such long pulses cause eclipsing at short ranges, i.e. while the long pulse is still being transmitted, the radar receiver must be turned off so cannot detect targets in the eclipsed region. Most long pulse radar therefore also utilise a short pulse regime to overcome this problem. Modulations used for radar applications have typically been ‘Chirp’ waveforms utilising Linear swept Frequency Modulation (LFM) or Non-Linear swept Frequency Modulation (NLFM), see section 4.3.3.1. Pulse compression filters in the radar receiver effectively trade the bandwidth of incoming pulses for time/range resolution.


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The most practical way to implement a coded pulse for use with existing transmitter systems is to generate the pulse at low power and amplify it up to a high power using the radar transmitter. Modern digital signal generation techniques can be employed to generate high quality coded waveforms. Therefore replacement of the pulse generation circuitry within the radar transmitter with modern digital pulse code waveform generation circuitry could result in more spectrally efficient radar. This section will identify the feasibility of this technique. Alternatively if amplifying a coded waveform is not feasible due to the nature of the radar transmitter then it may be possible to modify the radar pulse by some other means during the pulse generation of the radar transmitter.

This review will briefly describe existing radar transmitter techniques. The different radar transmitter techniques will then be reduced to just those used on radar within the UK. These will be described in more detail with a view to improving spectrum efficiency.

4.3.1.2 Radar transmitter requirements Radars must meet several system requirements. The most important requirement is the ability to detect targets out to a specified maximum range. The job of the transmitter is to provide sufficient RF energy to detect the target. For a radar to detect targets at a given range, R, then it must overcome a range loss proportional to 1/R4. As described above, the radar transmitter could thus provide short pulses with high peak powers or longer pulses with lower peak powers to achieve the same probability of detection.

Another consideration is the pulse repetition frequency (PRF) and it’s reciprocal, the pulse repetition interval (PRI). In order to improve SNR and overcome target fading variability, multiple pulses must be integrated together by the radar signal processor, thus the radar needs to illuminate the target several times on each scan. To avoid ambiguous target returns, the PRI must be longer than the round trip time for a pulse reflected off a target at the specified maximum range. This limits the number of pulses illuminating a target within the time for the radar beam to scan past the target. Taken together the PRF and pulse length determine the duty cycle the radar transmitter must maintain according to:

duty = PRF.τ Equation 4.3-2

Depending on the transmitter technology, there is typically an upper limit on duty:

TWT ~1% → 5%

Solid state ~10% In addition, for solid state transmitters there is typically a maximum pulse length which can be sustained. For current solid state technology this is around 200µs.

As indicated in 4.3.1.1, high power transmitter devices, suitable for radar typically operate in class C, which means that the RF amplifier devices are run into saturation. Under these conditions, it is relatively straightforward to apply phase modulation, however, in order to maximise power efficiency and reduce noise output the amplifiers must be turned on very quickly, which constrains the rise and fall times of the RF pulses. These rise and fall times are very much controlled by the transmitter technology used and will be discussed in the following report sections.


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Many radar applications (in particular ATC) require some form of Doppler processing in order to discriminate moving targets from stationary or slow moving clutter. Such Moving Target Indication (MTI) and Moving Target Discrimination (MTD) systems thus require good pulse-to-pulse frequency stability. Power oscillator based radars generally have poor frequency stability, however, driven amplifiers tend to perform much better due to the user of synthesised source or crystal oscillator based frequency generators in the low power stages. The Doppler processing integrity remains dependant on the quality of the amplifying stages.

Due to the safety and mission criticality of radar systems, radar transmitters are generally designed for high efficiency, reliability and maintainability. A radar transmitter is thus always compromise of all of the technical and mission critical requirements outlined above for any given application and budget.

4.3.1.3 Existing radar transmitters There are several types of transmitters used in civil radars in the UK. These fall into two main categories, power oscillators or power amplifiers. Power oscillators are typically self-excited oscillators such as the magnetron. Power amplifiers can be split further into driven tube amplifiers and solid-state amplifiers. Driven tube amplifiers include crossed-field amplifiers (CFAs), klystrons and travelling-wave tubes (TWTs). Solid-state amplifiers are typically transistor amplifiers based on bipolar technology and more recently FET and LDMOS technology. The high power microwave tubes can be of two designs. One type has orthogonal magnetic and electric fields such as the magnetron and CFA. The other type are linear beam devices, which have a continuous electron beam traversing an interaction region for example the klystron and TWT.

A radar transmitter is more than just the RF power source, it also includes all the ancillary equipment required to support it. It includes the low-level signal generation and amplifiers. Pulse waveforms require pulse modulators to control the high power tubes (this is not required for transistors). There also needs to be power supplies, cooling, safety interlocks and monitoring devices.

Radar transmitters are typically operated saturated, i.e. completely on or off with no intermediate power levels. This maximises efficiency, however in the case of coded waveforms it may be desirable to have amplitude-tapered, or shaped pulses. These require amplifiers that operate in a linear region on the rising and falling edges of each transmitted pulse. These linear transmitters have traditionally been much less efficient than saturated radar transmitters.

4.3.1.4 Transmitter power oscillators

4.3.1.4.1 Magnetrons The magnetron is a high power microwave oscillator that has been used in radar transmitters for over 60 years. It is a crossed field device such that it employs magnetic and electric fields that are perpendicular to each other. Magnetrons are simple, compact, low cost devices that produce high peak powers with high efficiency. They can produce pulses


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of fixed duration and frequency, but with a random starting phase and little control of their rising and falling edges.

+

-

-

-

-

+

+

+

Cathode

Electron Cloud

RF output

AnodeVanes

Figure 4.3-1 : Diagram showing the construction of a conventional magnetron

A cross-section of a conventional magnetron structure is shown in Figure 4.3-1. The magnetron converts the potential energy of an electron cloud near a cathode into RF energy in a series of cavity resonators defined by the radial vanes of the anode. The size of these cavities along with their inductive and capacitive properties determines the resonant frequency of the magnetron.

The heated cathode in the centre of the magnetron provides a source of electrons. Electrons are accelerated from the cathode towards the microwave resonant cavities in the anode. The presence of the magnetic field produces a force on each electron causing the electrons to spiral away from the cathode forming a cloud of electrons around the anode. At the anode the cloud of electrons falls under the influence of the RF field at the vane tips. The RF field modulates the electrons (this effect is known as velocity modulation) automatically forming a collection of electron spokes. On each subsequent half cycle of RF oscillation, this spoke pattern will rotate to maintain its presence in the opposing field. The resonant cavity vanes are designed such that the RF signal will be at opposing potential at each vane tip. This gives maximum power transfer of energy to the RF field and is called the mode. Inserting inductive pins into the inductive region at the back of each resonant cavity performs tuning.

The anode structure is designed to support the mode by electrically connecting alternative vanes together, commonly known as strapping. A magnetrons anode structure dimensions are governed by its frequency. This and the separation distance needed between the vane tips to prevent arching determine the maximum number of vanes possible and the overall power of the magnetron.

A coaxial magnetron differs from the conventional type by the addition of a high Q resonant stabilising cavity around the anode structure. RF energy is coupled through slots from alternative resonant cavities in the anode structure into the stabilizing cavity. RF energy is then coupled from the stabilizing cavity to the waveguide output.


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Most of the energy in a coaxial magnetron is stored in the stabilizing cavity; hence there is less energy at the resonator vane tips reducing the tendency to arcing. This allows magnetrons to be built with more resonators giving lower cathode emission densities, longer life and higher reliability. The frequency of the coaxial magnetron can be tuned by moving a tuning piston into the stabilizing cavity. The high Q resonant cavity around a coaxial magnetron greatly increases the frequency stability against input power and load variation, whilst providing a large amount of harmonic suppression. Unfortunately the cavity reduces starting time, hence the tube is more susceptible to time jitter effects. Coaxial magnetrons result in a larger and heavier magnetron for a given RF output.

Magnetrons must be operated in the correct mode, ensured by setting the correct rate of rise (RRV) of the cathode voltage. If this rise is too fast then the magnetron will not start to oscillate or oscillate in the wrong modes. If the rise time is too slow lower current modes may be excited or poor starting jitter may be produced. If the rate of fall of the modulating pulse on the cathode is too slow then magnetrons produce a noise output following the RF pulse.

Using magnetron-based transmitters with coded pulses is difficult because the shaped pulses cannot be generated at low power and then amplified by the transmitter. The magnetron itself would have to generate the shaped pulse. This is difficult to achieve. Magnetrons require special modulator circuits to make sure the correct rate of rise of voltage appears on their cathode, if this is wrong then the magnetron may not fire into the correct state causing spurious oscillation and missed pulses.

Generally magnetrons designs have already maximised their spectral efficiency by careful design of the modulator circuitry to give the correct RRV for oscillation and minimum spurious output. The addition of a high Q stabilising cavity in a coaxial magnetron ensures the best spectrum efficiency available from a magnetron. The output frequency spectrum of a conventional magnetron is illustrated in Figure 4.3-2. This figure shows the wide spread of the fundamental frequency. Also shown is the noise generated by the magnetron at higher frequencies. Figure 4.3-3, shows the spectrum spread from a coaxial magnetron, the output frequencies and powers are the same from both magnetrons. Clearly the spectrum spread from the coaxial magnetron is much narrower than the conventional magnetron. There are also no large spurious emissions generated at high frequencies.

It is clear that magnetrons are not suitable for nearly all forms of coded waveform; however, with careful design the spectral efficiency has been improved, particularly with the introduction of coaxial magnetrons. In older radar systems it may be more cost effective to consider upgrade to coaxial magnetrons as a means of improving spectral efficiency rather than a move to TWT or Solid State.


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Figure 4.3-2 : Spectrum Signature of Conventional Magnetron (200MHz/DIV, 10dB/DIV,Centre Frequency 2.84GHz)

Figure 4.3-3 : Spectrum Signature of Coaxial Magnetron (200MHz/DIV, 10dB/DIV, Centre Frequency 2.8GHz)


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4.3.1.5 Driven tube transmitter power amplifiers

4.3.1.5.1 Crossed-field amplifiers The crossed-field amplifier (CFA) is a microwave amplifier utilising perpendicular magnetic and electric fields. As in a magnetron the RF-DC interaction region is a region of crossed magnetic and electric field hence its name. They are characterised as non-linear saturated amplifiers that are smaller in size, have higher efficiency (40-60 percent) and use lower voltages than linear beam tubes. CFAs are moderately wideband devices (10-20 percent) and are capable of extremely high peak powers with good low noise performance. They do suffer low gain so are usually found used with a medium power TWT as a driver amplifier. CFAs are commonly used in phased arrays because of their good phase coherency and linearity, but they are also utilized in Doppler systems, frequency-agile radars and pulse compression systems.

CFAs fall under two categories: injected-beam where the electrons are injected into the interaction region by an electron gun and distributed-emission where the electrons are emitted by a cathode. The injected-beam CFA is generally not suitable for high powers so will not be considered further. CFAs can be operated CW, but pulsed operation is the most common for radar applications. CFAs can be constructed in either a linear or a circular format, these are illustrated below in Figure 4.3-4.

Input Output

Slow wave circuit

Matchingstructures

Cathode(a)

Magneticfield

Slow wave circuit

Cathode

Input Output

Matchingstructures

(b)

Magneticfield

Figure 4.3-4 : Simple representation of a CFA, (a) linear format and (b) circular format.

CFAs can be very difficult devices to drive requiring high power isolators both at their inputs and outputs. The drive level must be correct otherwise the CFA will operate in undesirable modes when the anode voltage is applied. They are not linear amplifiers but more like phase locked oscillators. Their output power is determined by the dc power being supplied to the tube. They will remain stable over a particular frequency range for given input and output conditions. For any given input drive level there is a fixed range in output power the tube will operate over, determined by the cathode current, the onset of a competing oscillation or a limitation in the gain of the main amplifying mode.

CFAs could be used with coded pulse schemes. The pulses would be generated at a lower power and used to drive the CFA. A CFA normally remains open circuit even when it’s dc power supply is applied to the tube and only starts conducting when the RF signal is applied


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and secondary emission occurs. The amplification continues until the RF pulse is removed. When the RF pulse is removed the accumulation of space charge particles can cause spurious oscillations similar to noise on the fall of each pulse. This is not a problem in non re-entrant CFAs.

CFAs are not common devices. There are not currently any radar using them within the UK. CFAs are now old technology that has been superseded. They do not offer the value of magnetrons or the performance of TWTs or solid state, so it is unlikely they will ever be used in radars again within the UK, CFAs thus will not be considered any further.

4.3.1.5.2 Klystron A diagram of the klystron is shown below in Figure 4.3-5. The klystron consists of a heated cathode that emits a beam of electrons shaped by the electron gun. The electron gun consists of the cathode, modulating cathode, anode and the collector that removes the electrons. The electrons emitted from the cathode converge into a high-density beam of electrons. Around the beam are multiple RF resonant cavities at anode potential. Surrounding the Klystron is a magnet (either a solenoid or permanent magnet) providing an axial magnetic field to confine the electrons to a long thin beam.

ModulatingAnode

Cathode

RF Cavities

Collector

Drift Space

Interaction Gaps

Anode

RF in RF out

Electron beam

Figure 4.3-5 : Diagram of the principle parts of a three-cavity klystron amplifier.

The RF signal is applied across the interaction gap of the first cavity. Electrons that arrive at the interaction gap experience acceleration or deceleration depending on whether the RF signal is at a maximum (peak of sinusoid) or minimum (trough of sinusoid). This effect is called velocity modulation and causes bunching of the electron beam in the drift space. Further interaction gaps cause amplification of this effect. The interaction gap of the output cavity is placed at a point of maximum bunching so power can be extracted from the density-modulated beam. The collector is usually held at a negative potential to reduce the kinetic energy of the electron beam so less heat is dissipated thus increasing efficiency.

Klystrons usually have 3 to 4 stages with about 15 to 20 dB gain per stage; so gains of 50 dB are typical, however gains up to 70 dB can also be expected. Peak powers in excess of 1 MW are possible with bandwidths between 1 to 8 percent of the carrier frequency and efficiencies of 40 to 60 percent. Klystrons are required to be carefully tuned to prevent them producing significant harmonics or oscillating.


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Klystrons could also be used to amplify coded pulses for a radar transmitter, however Klystrons are also class C devices operating heavily into saturation. Therefore the coded pulses would get distorted as they pass through the saturated stage of amplification. Like the CFA, Klystrons are not used for ATC radar within the UK, however they are used on some UK ships. Klystrons will not be considered further for the purpose of this study because they are not used over the heavily populated areas of the UK where there may be sufficient benefit from the sale of spectrum to the communications industry.

4.3.1.5.3 Travelling Wave Tube Travelling wave tubes (TWTs) are used as wideband microwave amplifiers, they are similar in structure to klystrons but do not generate the same power levels. They have high gain and large bandwidth although they do suffer from relatively low efficiency and require high voltages. Figure 4.3-6 below shows the construction of a typical travelling wave tube. As in the Klystron there is an electron gun that consists of the heated cathode, modulating anode or grid, and the collector. The significant difference is that the interaction region is not just the gaps between the resonant cavities, but over the full length of the propagating structure within the beam. The propagating structure is usually a delay line, which is typically a slow-wave helix structure at lower power levels. The slow-wave helix structure can have bandwidths of over an octave. However it is limited to peak powers of a few kilowatts. For higher power TWTs, simple helix structures are not used. For powers up to 200 kW the ring-and-bar configuration is common. Greater powers require either the two-tape contra wound helix or the coupled cavity TWT. The coupled cavity TWT is by far the most common for high power, however does suffer from reduced bandwidth over the slow-wave helix design.

CollectorHeater

Cathode

RF in RF out

Attenuation

AnodeHelix

(interaction region)Electron beam

Figure 4.3-6 : Diagram of the principle parts of a TWT.

As a RF signal passes along the helix the electron beam only experiences the longer and slower wave components in the longitudinal direction. The RF signal travelling around one turn of the helix only moves forward by one pitch, i.e. its velocity in the direction of the electron beam is slowed down (hence the name slow-wave structure) so that its velocity is similar to the electron beam. Cumulative interaction between the RF energy in the helix and the DC energy in the electron beam causes velocity modulation of the electron beam as in the Klystron. Energy is passed from the electron beam to the RF signal causing amplification.

The whole structure has an axial magnetic field around it to focus the electrons into a long thin beam. The two most common methods for focusing the beam are the periodic-permanent magnet (PPM) method (lower peak power) and solenoid focusing (for higher peak power). Tubes that use solenoid focusing have the coils foil-wound as an integral part of the tube and succeed in reducing the magnetic field variations to less then 1%. Tubes


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using PPM do not achieve the same degree of focusing as the solenoid focusing TWTs, but they are smaller in size and do not require extra power supplies.

TWTs do suffer from low efficiency. This is due from the heat dissipated in the collector, electron gun, slow wave structure and body. Sufficient cooling must be available to keep these items cool. Normally either water, forced air or convection cooling is used. The efficiency can be improved in two ways, the first is using a depressed collector and the second is velocity resynchronization. Depressed collector involves operating the collector at lower potential than the body voltage. This causes the electrons to reduce their velocity and hence reduce their kinetic energy before they strike the collector. Therefore the heating effect in the collector is reduced and the TWT efficiency increases. For velocity resynchronisation the TWTs body is split about ¾ along its length with the second section set at a slightly higher potential than the rest of the body. The main problem with this is creating an insulation portion that does not introduce significant RF reflections and mismatches. Efficiency increases to levels of 50 to 60% can be achieved by using depressed collector and resynchronisation techniques.

The TWT beam is usually modulated via modulation of the cathode or control grid. For higher powers the slow-wave helix structure is replaced with a ring and bar configuration because it can cope with higher temperatures. Even higher powers require a coupled cavity configuration similar to the Klystron.

Modulation of the electron beam in the TWT can be either by cathode modulation or by using a control grid. Arcing can be a problem in TWTs, particularly those with a control grid requiring the power supplies to have fast acting energy diverting circuits to remove the high voltages should an arc begin to develop.

TWTs are coherent transmitter devices and thus suitable for amplifying certain types of coded frequency modulated waveforms such as FM chirps, however, since they are operated in class C with rapid rise and fall times (of the order of 10s of nano-seconds) they are not suitable for amplitude pulse shaping.

4.3.1.6 Solid state transmitter power amplifiers Solid state radars utilise numbers of solid state power transistors to amplify low power RF pulses for transmission. Each transistor is generally low gain and low output power when compared to vacuum tube devices so that the output of many transistors must be combined together to achieve adequate transmit power.

The low peak power of solid-state devices is due to the heating effect of the transistor during the pulse. Solid-state devices have poor thermal conductivity and hence when an RF pulse passes through the transistor its internal junction temperature increases very quickly. This excess heat must be dissipated otherwise the junction will burn out. Poor thermal conductivity limits the rate of heat dissipation, hence also the peak power that can be achieved. However, the transistors can be switched on for much longer periods of time.

In addition, as a result of the solid state technology and the need to combine multiple transistors; the rise and fall times of pulses generated by solid state radar transmitters is significantly slower than that of vacuum tube devices. This means that very short pulses are difficult to achieve.


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The very short pulse lengths and low duty cycles used in high power tube based transmitters thus simply cannot be achieved with solid-state transmitters. Therefore direct replacements where the radar pulse routine remains the same are not very common. Solid-state radar are typically new designs with the transmitters operating with low peak powers and longer pulse widths of the order of 10 to 200 µs long in order to get enough energy on the wanted targets.

However, in order to maintain range resolution the radar must make use of pulse compression waveforms such as LFM or NLFM chirps to modulate the long pulses. To overcome short range eclipsing, as described in 4.3.1.1, a short-range pulse is transmitted between the long-range pulses to maintain short-range detection. This extra pulse requires additional signal processing in the receiver. Low pulse compression time side-lobes for the long pulses and a high clutter cancellation ratio are a necessity and due to the extra complexity required in the radar receiver and signal processing, solid-state systems are generally new systems. Some TWT based transmitters such as Watchman also have the similar requirements with regard to pulse compression and short-range pulses.

Solid-state transmitters, however, do offer many advantages over more traditional radar transmitter technologies:

No hot cathodes, eliminating warm-up delay, increasing efficiency and operating life.

No high voltages required, thereby reducing power supply size.

Solid-state amplifiers have better MTBF in comparison to tube transmitters due to their soft fail nature.

Solid-state amplifiers generally operate in class C therefore are self-pulsing and require no pulse modulator.

Soft failure is possible. The large number of solid-state modules means the radar can still perform if some solid-state devices fail.

Active phased antenna arrays are possible with a different module available for each antenna element. Beam forming can be performed at low power on the input side of each active array module.

4.3.1.6.1 Solid state technologies In currently deployed radar systems in the L and S bands, there are several types of solid-state transistors in use, although there are new variants in development all the time. The main categories are silicon bipolar, MOSFET and Gallium Arsenide FET (GaAs). Depending on which frequency band is being used, solid-state transistor output RF power devices range in output power from 10W through to about 300W (although this is also increasing all the time). It is normal to combine multiple transistor stages into modules that are then combined in the radar transmitter to generate adequately large peak powers required for radars.


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Microwave Silicon Bipolar Junction Transistors (BJT) The Silicon BJT is the original solid-state RF power device originating from the 1960s, and is most common in L and S-band solid state radar transmitter systems, but less so at higher frequencies as device performance starts to fall. The fundamental properties of RF bipolar transistors remain similar to ordinary bipolar transistors through their operating frequency range.

Microwave bipolar transistors are usually complex hybrid circuits. Lower power stages use single-chip devices whereas higher power stages use multiple dies combined in parallel within the same hermetically sealed ceramic package. This package also includes internal matching circuitry to increase the terminal impedance of the die to that of the circuit. This technology is operated in class C (saturated output) to maximise efficiency and stability. Today this is a very mature technology providing good output power, reliability and performance.

Metal Oxide Silicon Field Effect Transistor (MOSFET) MOSFETs are commonly produced using a double-diffusion process. They have an insulted gate and hence conduct no DC current therefore are very easy to bias. Thermal runaway is prevented due the MOSFETs negative temperature coefficient causing drain current to decrease with temperature increase. Multiple MOSFETs can be connected in parallel to build up RF power devices. They will typically operate off 12V and 28V power supply rails.

Microwave Field Effect Transistors (FETs) Gallium arsenide (GaAs) FET transistors are commonly used for frequencies up to G/H-band (just below 10GHz). Although they are more limited in output power capability, due to the poor thermal conductivity of GaAs and its lower breakdown-voltage, GaAs FETs will operate at higher frequencies than silicon bipolar devices because GaAs has a higher electron bulk mobility and greater maximum electron drift velocity than silicon. They have been used in Active Array Multi-Function Radar designs where high duty ratios (up to 20% – 30%) are required in order to support the various radar tasks being simultaneously carried out.

GaAs FET transistors can be operated in class AB (near linear) however in radar designs have mostly been operated in class C (saturated) to maximise efficiency and stability.

4.3.1.7 Summary Several types of common radar transmitters have been described in the preceding section explaining the principles behind the key technologies in current radar transmitter design within the L and S bands. Power oscillators, driven tube power amplifiers and solid-state based transmitters have all been reviewed. Some of the transmitter designs such as cross-field amplifiers and klystron amplifiers will not be described any further because these systems do not exist within the densely populated areas of the UK.

The most common types of high power radar transmitters used within the UK have been identified as magnetron, TWT and solid-state based systems. With respect to the sell-off of spectrum, the study can only concern itself with civil radar. It will not be possible to free up spectrum from military radars in the same way to civil radar. Military radar within the L and S


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bands are also used for ATC purposes but not necessarily in fixed locations. Any free spectrum within the L and S bands available for commercial purposes would most likely come from making civil ATC radars more spectrally efficient. If this spectrum can be traded for new applications within the communication sector, the market will most likely be near the densely populated areas of the UK.

In the UK high power radars that commonly operate in the L and S bands use magnetron, TWT or solid-state based transmitter systems. Magnetron based transmitters are generally used for marine radars where freeing up spectrum would have little commercial value because it is not being used over densely populated areas in the UK. Civil radars that use these transmitters nearly all belong to the National Air Traffic Control (NATS). NATS radars are mainly TWT and solid-state based radars, they do still have some magnetron systems however generally these are being phased out. Therefore effort shall be focused towards TWT and solid-state based systems. As a result this study will focus more deeply on the application of coded waveforms with the following transmitter systems:

A TWT Terminal ATC Radar in S Band

A Solid State Terminal ATC Radar in S Band

There are examples of the above systems all owned by NATS. The TWT transmitter based system is the Watchman made by BAE SYSTEMS Insyte (formerly Plessey, Radar Division). The solid-state transmitter based system in S-band is the ASR-10SS made by Raytheon. Where commercial sensitivities allow parameters representative of these radar will be used in the analysis.

4.3.2 Review of basic performance limitation of current power devices

4.3.2.1 Introduction This section contains a more detailed analysis of the transmitter types within the UK identified in the previous section as being capable of coded waveform transmission. The TWT and solid-state types of radar transmitter are examined to determine the possible limitations of these transmitter types when used to produce high power coded waveforms.

4.3.2.2 Driven vacuum tubes

4.3.2.2.1 Klystrons and Travelling Wave Tubes Essentially similar problems occur for both klystrons and travelling wave tubes when they are to be used with spectrally efficient coded pulses. Therefore they shall both be considered together under the term driven tubes.

A typical characteristic of a driven tube is that its gain and hence it output power varies with frequency across it’s frequency band. As the drive level increases the variation in the output level across the frequency band reduces because the tube is going into saturation. Figure 4.3-7 shows the output power of a typical travelling wave tube against frequency for different input drive levels. The red lines represent the limits the tube operates under in normal conditions where it is being operated into compression. At this drive level the output


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power level is fairly flat across the frequency band. However if the tube were to be run unsaturated, there would be large variations across the frequency band. From Figure 4.3-7 it is clear that the output power of the tube varies with frequency and that this effect is minimised if the tube is driven more into compression.

Even when the tube is operated in compression some output power variation remains. This is especially true because the tube cannot be overdriven i.e. driven hard into compression. The effect of overdriving the tube increases the body current and can damage the tube. The electron beam becomes defocused and electrons hit the slow wave structure. This causes heating within the tube structure and will eventually lead to damage within the tube’s body and loss of vacuum. Therefore it is essential not to drive a high power vacuum tube too hard into saturation. Operating a tube in compression also has the effect of reducing the level of the in band transmitted noise because any noise sitting on top of the transmitted signal is also be heavily compressed.

Typical Output Power against Frequency for Watchman TWT

0

20

40

60

80

100

120

140

160

2.75 2.8 2.85 2.9 2.95 3 3.05

Frequency (GHz)

Pow

er (%

)

-10dB-8dB-6dB-4dB-2dB0.3Wlower limit (50.7kW)upper limit (69.04kW)

Figure 4.3-7 : Output power against frequency for TMD travelling wave tube used in the Watchman radar for different drive levels.

The compression characteristic of a high power vacuum tube can cause several problems with coded pulses. An amplifier operating in compression has different amounts of gain depending on the input drive level. Small input signals are amplified more than large input signals. This has the effect of distorting a coded pulse as it is amplified up through the radar transmitter. If the input pulse has shaped edges in order to make it more spectrally efficient, then the effect of the amplifier operating in compression will be to square up the edges of the pulse thereby making the transmitted pulse less spectrally efficient. This effect is illustrated in Figure 4.3-8 below.


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Time

Pow

er

Time

Pow

er

(a) (b)

Figure 4.3-8 : The effect of passing a shaped pulse through a saturated transmitter stage: a) Shaped pulse with cosine shaped edges, b) distorted pulse with more rectangular edges.

For any high power vacuum tube a phase change occurs when the pulse passes through the tube. As the pulse builds up from a small value the phase of the pulse starts spinning quite rapidly. This effect reduces when the tube starts to go into compression. Another reason that radar pulses have evolved to have very fast edges and to be operated in compression is to minimise this phase spinning effect. Overdriving the tube can also cause a phase change effect from overdriving the slow wave structure. The electron bunching (or velocity modulation) is so heavily modulated that effectively the electrical length of the electron beam is modulated. Therefore an optimum operating point for any radar transmitter tube is slightly into compression. There will always be a little phase change that will distort any carefully coded pulse driven through the system.

Consider a typical tube with modulator circuit, as shown in Figure 4.3-9. There are several supplies needed to drive a travelling wave tube. Three-phase mains electricity is converted to HT by the main inverter. Normally a switched HT supply is generated around 300V. This drives the transformer rectifier that steps up and rectifies the inverters output to about 16kV for the collector supply and 36kV for the cathode supply. There is a second inverter to generate supplies for the grid and heater. These are controlled through the floating-deck modulator that regulates the inverters output and controls the grid switching. An optical signal comes from the radar processor to tell the floating-deck modulator to tell when to switch the grid. There is also an overall TX control unit to keep at the timing in check. The ion pump also has its own supply.

The grid drive voltage has to be very stable. Any fluctuations on the grid drive voltage result in the grid modulating the phase of the signal passing through the tube amplifier. Similar modulation processes are also true for the body and cathode voltages and to a lesser extent the collector voltage. This effect is called phase pushing. Phase pushing is also caused by the variation in drive level. Various figures for phase pushing for different supply voltages used within the TWT are included in Table 4.3-1. Note Table 4.3-1 expresses phase or power change against percentage change of voltage on the specified electrode. This phase noise needs to be minimised to make the radar more spectrally efficient.

Power pushing is also a consideration. Any variation of the body, collector or grid voltage can cause variation in output power performance. The figures for the Watchman tube are also shown in Table 4.3-1.

Table 4.3-1 : Various pushing figures for a Watchman TWT tube.


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Phase Pushing Body 35°/1%

Phase Pushing Collector 0.5°/1%

Phase Pushing Grid 7°/1%

Phase Pushing Drive 6°/1%

Power Pushing Body 0.7dB/%

Power Pushing Collector 0.01dB/%

Power Pushing Grid 0.15dB/%

FloatingDeck

Modulatorand HeaterRegulator

Traveling Wave Tube

IsolationTransformer

AUX Inverter

TransformerRectifier

Main Inverter

InverterSupply

Tx Control andInterlocking

Ion PumpSupply

RF input RF output

GridDrive

CathodeSupply

CollectorSupply

Body

Mains

OpticalGridDrive

Figure 4.3-9 : Generic diagram of TWT power supplies and control circuitry.

Figure 4.3-9 shows that the body, collector and grid voltages are all EHT supplies; because they are such high voltages they are very difficult to keep constant, hence pushing will always be a problem to some degree for a high power vacuum tube. There will always be a phase change in the RF pulse as it passes through the tube. The phase change is very tube design specific, almost to the extent it will change with different tubes of the same design. This phase change will always distort any coded pulses passing through the system.

There is a natural tendency for a high power vacuum tube amplifier to have a large gain with a small input signal and a small gain with a large input signal due to the saturation effect of the class C stage. This coupled with the pulse-to-pulse phase changes mean every coded pulse through the TWT will get distorted. Therefore any noise generated by the


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transmitter will get amplified more when the RF pulse is low level, i.e. during the rise and fall times. The transmitter can be switched off or disconnected whilst the RF pulse is off to minimise noise generation. If the RF pulse edges are slower to make the pulse spectrally more efficient then the radar transmitter will generate more noise during the pulse edges. This gives rise to the term ‘rabbits ears’. When a pulse with noise on its edges is viewed in the frequency domain then it looks like it has a pair of rabbit ears attached to the pulse edges. This noise needs to be minimised to create a spectrally efficient pulse.

4.3.2.3 Solid state The low power output from solid-state transistors means several amplifying stages have to be combined to generate the high power required for a radar transmitter. Many solid-state systems use an amplifier combination approach similar to that shown in Figure 4.3-10.

input output

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ay s

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er

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ay c

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r

Figure 4.3-10 : Diagram showing a typical solid-state RF power module.

The arrangement of RF power transistor combination shown in Figure 4.3-10 is commonly thought of as a power amplifier module. A power amplifier module may typically have one driver stage driving two more driver stages that will each drive four-output stages. A combiner will then combine all the outputs in phase to generate a powerful module. The power amplifier module may have more or less amplifier stages than those shown above.

A radar transmitter will then combine several power amplifier modules together in order to generate a high peak power. A circulator or switched duplexer is used on the overall output before the feed to the antenna to protect the amplifiers from high-load VSWR. The peak powers produced are not as large as a vacuum device. Whereas a vacuum tube device might generate a peak power of up to 1 MW, a solid-state transmitter generally only delivers a peak output power of only around 10 - 20kW. To deliver the same energy on target as a vacuum tube transmitter then a solid-state transmitter will use much longer pulses. For current solid state devices the maximum pulse length is approximately 200µs, although this length is unlikely to be utilised since the required PRF of ~1000Hz for ATC applications


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would lead to a high duty cycle of around 20%. Current solid-state transmitter technology has a maximum duty cycle of around 1 to 10%. For comparison a high-power vacuum tube e.g. magnetron will have a duty cycle in the region of 0.001 to 0.01%, whereas a TWT transmitter will have duty cycle in the region of 1 to 5%.

The power amplifier structure used in the above example is known as the corporate-combined structure. These sort of solid-state systems replace high power vacuum tubes with a bulk set of solid-state modules. Solid-state power amplifiers can also be created using a space-combined structure where the outputs from power amplifier modules are combined in space. The active phased array transmitter is an example where each power amplifier drives a radiating element and the wave front is formed in space. Hybrids of both ideas also exist.

Class C bias is the most common for solid-state devices because this maximises the RF power output for a given power supply input. This results in a higher efficiency for the amplifier because no quiescent current is being drawn whilst the device is not driven. Also because the class C configuration is a self-bias then only one power supply is needed for the collector of the RF power transistors.

Figure 4.3-11 : Typical values of output power as a function of input power for a Class C Phillips BLS2731-50 S-band silicon bipolar microwave transistor currently available for radar applications. Note that the

linear region only exists for a small range if input powers. Note also that the device does not operate at all until a threshold input power is applied.

There are some characteristics of class C operation to consider when thinking about a radar amplifier. As the RF drive level is increased from zero, the device begins to draw collector current when the dc potential of the reversed-biased base-emitter junction is surpassed. The transfer characteristic of the device will remain linear with increased drive level until the device begins to saturate. The device will reach output power saturation where further increase in input power degrades the output power. This occurs when the device is thermally limited at the junction. The device will continue to draw collector current with increased drive. The optimum operating point is usually 0.5-0.75 dB into saturation.

A class-C bias device operating under normal chosen conditions will suffer from sensitivities in its output power and phase response due to variations in input drive level, collector voltage, temperature and load impedance. These sensitivities can degrade the output pulse


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envelope characteristics considerably if not considered within the design. The linear transfer region of the single stage device may only exist over a very small window of 1 to 3 dB. For a class C amplifier the last stage in the cascaded chain must be driven into saturation so not to degrade the output pulse. Any small reduction in drive level at the beginning of the cascaded chain could result in the output stage coming out of saturation because of the small operating range of all the cascaded stages. The drive level is usually held constant by a feedback path so to maintain the output pulse at a constant level across the frequency band and for the duration of the pulse.

Output power can vary by up to 0.2-0.9dB per dB of input drive change (depending on saturation level) or by 0.2-0.4dB per volt of collector voltage change from nominal values. Phase change across the device can vary by 10-13° per dB of input drive change or 0.5-1.5° per volt of collector voltage change from nominal values. Phase change during the pulse due to device junction heating effects can be 5-20°. It is not uncommon for the loading effects from the power combiners and circulators following the RF power transistor to vary by ±50%. The phase of this mismatch can cause the port-to-port characteristics of the device to vary dramatically. Any variations in port-to-port insertion phase can cause problems with combining inefficiencies. As shown in Figure 4.3-10 RF power amplifiers are normally combined in parallel so any variation in the phasing when the amplified signals meet in the combiner results in power dissipated across the load.

4.3.2.3.1 Module design The power amplifier module design as shown in Figure 4.3-10 will also contain matching circuitry to match the low input and output impedance of the RF transistors into the 50 ohms impedance of the rest of the microwave circuitry. The splitters and combiners used in the module must also provide isolation between the parallel amplifiers. Therefore if a device fails then the combiner will provide constant load impedance on the remaining devices. The amplifiers must also be isolated serially because the devices input and output impedances vary as the device passes through its cut-off, linear and saturation regions. The input impedance changes the most and can change from near infinite in the off state to well matched in the on state. For serially cascaded devices with no isolation then the varying input impedance appears as the load impedance of the previous stage, possibly sending the previous stage into oscillation. Use of hybrid combiners and splitters or 90° offsets can provide the isolation needed by making use of reflected signal phase cancellation techniques.

The power supply for the module must supply the peak current for the RF power transistors during the pulse. A large capacitor bank within the power amplifier module or local decoupling banks next to the RF power transistor provide this function. It is common to charge the capacitors with a constant current during the pulse repetition interval and then disconnect the power supply during the pulse and rely on stored charge to power the RF power devices during the pulse. This prevents any noise from the power supply from getting on the RF pulse. It also allows a much more constant load to be presented to the radars bulk power supply.

The errors due to the phase and amplitude sensitivity of the solid-state transistor amplifiers to power supply ripple can limit the MTI improvement factor obtained. The phase errors in cascaded stages will add. It is possible to reduce phase error due to power supply ripple by not synchronising the clocks within the power supplies for all the parallel stages assuming


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there are separate power supplies for each amplifier. Amplitude errors due to power supply ripple do not simply add due to saturation effects. Also of consideration is time jitter of the RF pulse, again this can add up through cascaded stages so has to be considered carefully.

As an alternative to combiner circuits within the transmitter, power combination can be achieved in space by using an active phased array antenna architecture. These typically consist of relatively low peak power amplifiers connected separately to each radiating element in an array. Located in the phased array module is a core-chip that controls the phase shifter, attenuator and switch the module between transmit and receiver duties. The advantage of the active phased array architecture is the module’s outputs are combined in space, which eliminates the losses associated with the combiner structures needed in the bulk transmitter designs. However, active phased array radar are significantly more complex systems typically only designed for military applications and are consequently much more expensive than is feasible for civil applications in the L and S-bands such as ATC.

4.3.2.3.2 Spectral emissions Solid-state power amplifier modules biased in class C typically show rise and fall times in the order of 10s of nanoseconds for a rectangular pulse, while this is slower than for TWT devices such rising and falling edges are still relatively fast. As already mentioned the output spectrum for a rectangular pulse is not very spectrally efficient. Solid-state radar transmitters usually consist of several serially cascaded stages. Using slower rise and fall times in order to increase the spectrum efficiency may be a problem to implement. The highly nonlinear effects of the class C amplifier stage will have the effect of speeding up the pulse edges and making the pulse more rectangular as it passes through the cascaded stages within the solid-state amplifier. In addition, if slow rising and falling edges were applied to the transmitter they could cause rapid phase and amplitude oscillations and much degraded transmitter performance.

In order for a solid-state transmitter to pass a more spectrally efficient pulse then it would need to behave more like a linear amplifier. For a cascaded chain of solid-state RF power transistors there is typically a 1-3dB window in input drive level for each device where it is linear; however it would be nearly impossible to make a cascaded chain where all the devices are operating in a linear region. The efficiency of this amplifier would also be very poor.

4.3.2.3.3 Investigation of the spectral performance of a real solid state radar transmitter system. Most of the analysis carried out to date has considered theoretical spectra calculated from radar waveform designs and some top level parameters of radar transmitter devices (including peak power, duty and rise/fall time). The transmitted spectra of practical real world transmitter systems, however, will not be the same as these theoretical spectra.

As a manufacturer of radar systems, BAE Systems Insyte is able to carry out experimental measurements on real transmitter systems. This section details an investigation into the actual transmitted spectra from a modern S-band solid state class C transmitter system compared to theoretical spectra for the same waveforms.

Phase and amplitude across the band of interest


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The first part of this investigation required measuring the amplitude and phase performance of the transmitter across it’s band of operation. The method used was based on “High PRF measurements” from an Agilent application note (5980-0798E).

Attenuator Pulse Modulator

TX

Network Analyser HP8753C

Figure 4.3-12 : Experimental set up to measure phase and amplitude performance across the band

The network analyser generated a swept frequency CW signal, which was slowly swept from 2.7GHz to 3.0GHz. This signal was pulse modulated to form 100µs pulses with a PRR of 800Hz before being applied to the input of the transmitter (TX). The high power output of the TX was attenuated before being sampled and analysed by the network analyser.

Within the network analyser the sampled signals are down converted to an IF, filtered and then down-converted a second time to a 4kHz IF. These signals are then digitized via an ADC and all other processing is done digitally. The greatest influence on the pulsed signal was caused by a digital IF BW filter which was set to 100Hz. As the analyser swept in frequency the IF BW filter tracked the source, selecting only a narrow section from the centre of the IF signal bandwidth. The amplitude and phase response of the transmitter device, at steps across the band, were stored and are presented in Figure 4.3-13 and Figure 4.3-14.


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2.7 2.75 2.8 2.85 2.9 2.95 3-1.4

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Frequency, GHz

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Figure 4.3-13 : Transmit power variation across the band 2.7 – 3GHz

2.7 2.75 2.8 2.85 2.9 2.95 3-4500

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Figure 4.3-14 : Phase change variation across the band 2.7 – 3GHz

In this way the network analyser is able to capture the spectral response across the entire band of interest; however, the measurements taken show the average response though out the pulse. They cannot supply data on how the amplifier performs at the beginning, middle or end of pulse. This is due to the IF BW filter and is therefore a limitation of the


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measurement. Note that for experimental reasons there are gaps in the measured responses around 2.8GHz and 2.95GHz, where data was not collected.

For this modern solid state transmitter it is clear from these measurements that there is only a small variation in transmitted power across the band and the phase shift from input to output is linear with a relatively shallow slope. Across a typical chirp bandwidth of 1MHz, this would cause only 13° of phase distortion, which is not significant.

Chirp spectrum measurements

The object of this exercise was to generate a LFM chirped waveform in a similar way to that produced for a practical Radar system, then capture frequency and time domain data for this waveform. The next step was to input the chirped waveform to the transmitter and capture frequency and time domain data at its output. In this way the effect the transmitter has on the spectrum can be evaluated by comparing it’s output spectrum to the input spectrum.

The chirped pulses were produced using an Anritsu Signal Generator (MG3694B). The generator’s modulation can be externally programmed using PC based software via its serial interface. The software enables the user to define the required modulation waveforms and allows simultaneous use of its AM, FM and pulse modulation modes to produce a LFM chirped pulse at S Band.

Modulation waveforms were generated for various pulse widths, with a fixed PRR of 1000Hz and two different sweep bandwidths as tabulated below:

Table 4.3-2 : Table of measured LFM pulse spectra

10µs 20µs 50µs 80µs 100µs 1.0MHz

2.5MHz The test equipment was set up as illustrated in Figure 4.3-15.


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ExtStandard

PC

Sig’ Gen’Anritsu

MG3694B

Serial Data

AM Mod’O/P

Pulse Mod’I/P

RF output Coupler

Sig’ Gen’IFR 2042

LO

RF

Oscilloscope

GPIB to USB

FM & AM

IF

Spectrum Analyser

Trigger

-20dB

Figure 4.3-15 : Test equipment setup

For data capture of the chirp wave from the Signal Generator the test equipment is connected as in figure 3 and set to work as follows.

The centre frequency and output power of the chirp waveform were manually set on the Anritsu signal generator, the modulation configuration data was then downloaded from the PC. A sample of the waveform was provided via a directional coupler to the RF input of a frequency mixer. The second signal generator is used to drive the mixer LO port and is tuned to enable visibility on the oscilloscope display of the LFM chirp frequency sweep (low to high) at IF, see Figure 4.3-16.The two signal generators are phase locked together using the external frequency standard (10MHz) from the Anritsu signal generator.

Figure 4.3-16 : 1.0MHz LFM chirp measured on oscilloscope

Spectrum data from the spectrum analyser was downloaded to the connected computer via the GPIB interface. This allowed the capture of the pulse spectra before input into the transmitter. In order to capture the transmitter output spectrum; these measurements were


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repeated with the coupler output connected to the transmitter input via an isolator and with the spectrum analyser input connected to the Transmitters output.

The transmitter operation in practice is limited to an 8% duty cycle although it will operate at a maximum of 10%. During the measurements the maximum duty cycle was 10% with the 100µs pulse, all the narrower pulses were running at 8% or less as the PRR was fixed at 1000Hz.

Measurements were taken at the signal generator output in both the frequency and time domain. This was performed for each pulse width tabulated in Table 4.3-2 at three points over the operational bandwidth; 2.700GHz, 2.825GHz and 2.900GHz.

For all these measurements, the Spectrum analyser was set to detect peak power and hold, the resolution bandwidth was set so that impulse response was not compromised. The output spectrum of the pulse at each frequency was measured over a ±22.5MHz band. This band was determined by the 55dB dynamic range of the spectrum analyser, the frequency response of the pulse was suppressed by the Spectrum Analyser LO noise at the extreme edges of this band.

The total 45MHz band was then broken down into separate measurements to improve the data resolution. The analyser was set to a 5MHz span and its centre frequency tuned in 4MHz steps over the bands, 2.700GHz ±22.5MHz, 2.825GHz ±22.5MHz and 2.900GHz ±22.5MHz with a data capture being made at each step.

The pulse rise time was also measured at the transmitter outputs, for both chirp bandwidths at each pulse width on 2.825GHz. An example time domain plot, illustrating the leading edge of the transmitted pulse is illustrated in Figure 4.3-17 below, note the slow rise time of 168ns and the overshoot before the transmitter settles, this is not something which is typically modelled and is one reason why real spectra differ from theoretical ones.

Figure 4.3-17 : Leading edge of the transmitter pulse output


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Frequency offset, MHz

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SS 80us 1.0MHz LFM TheoreticalSS 80us 1.0MHz LFM Sig GenSS 80us 1.0MHz LFM Tx

Figure 4.3-18 : Theoretical, Signal Generator and Transmitted spectra for 80µs LFM pulse

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SS 50us 1.0MHz LFM TxSS 80us 1.0MHz LFM TxSS 100us 1.0MHz LFM Tx

Figure 4.3-19 : 1.0MHz Measured LFM pulse spectra of various pulse lengths at 2.825GHz


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SS 80us 1.0MHz LFM Tx - 2.700GHzSS 80us 1.0MHz LFM Tx - 2.825GHzSS 80us 1.0MHz LFM Tx - 2.900GHz

Figure 4.3-20 : Measured Transmitted spectra for 80µs LFM pulses at various centre frequencies

Illustrated in Figure 4.3-18 are the theoretical and measured signal generator and transmitted spectra for 80µs LFM pulses. These are plotted on a linear frequency scale about the transmitter centre frequency. Of particular note is the observation that, while the measured transmitted spectrum matches fairly well with the theoretical spectrum within 1-2MHz of the centre frequency, beyond this region the measured transmitted spectrum rolls off much more slowly than the theoretical. This is as a result of real effects within practical transmitter systems which are not typically modelled in theoretical spectra. Additionally as the spectral skirts approach the -55dB noise floor of the spectrum analyser, the quality of the measurements visibly decreases. The asymmetry of the transmitted spectra is due to gain variations across the band.

Also of note are the differences between the signal generator output and the transmitted output. The signal generator output is symmetrical about the centre frequency but does not roll off very rapidly. This is because pulses output from the signal generator have very fast rise and fall times in order to ensure that the transmitter is cleanly triggered. If slow rising and falling edges were applied to the transmitter; this could cause rapid phase and amplitude oscillations and much degraded transmitter performance. At some points the transmitted spectrum is narrower than signal generator spectrum as a result of the slower rise and fall times.

Figure 4.3-19 is a comparison of the transmitted spectra of 50µs, 80µs and 100µs 1.0MHz LFM pulses. These measurements were made at 2.825GHz, which is around the centre of the transmitter’s operating band. Despite the different pulse lengths, there is little difference between the spectra for these pulses; this reflects the effect of the rise and fall times rather than the pulse lengths on the spectra. The spectrum of the longest 100µs pulse is slightly


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degraded in comparison to the shorter pulses since the transmitter was operating at its maximum rated duty of 10% (100µs at 1000Hz PRR).

Figure 4.3-20 illustrates the measured spectra of 80µs pulses, centred on different frequencies across the S-band (2.700GHz, 2.825GHz and 2.900GHz). Some variation is apparent from one frequency to the other, with different asymmetrical responses and different roll-offs in the skirts of the spectra. This reflects the phase and amplitude variations across the band and other real world transmitter effects.

Summary This investigation has demonstrated that solid state transmitters are capable of improved spectral efficiency compared to TWT based transmitters, which is mainly due to the inherently slower rise and fall times of the pulse edges rather than pulse length effects. However, there are real effects within practical transmitter systems which limit their spectrum efficiency and are not easily modelled when calculating theoretical spectra.

When considering improving pulsed radar spectral efficiency, one of the main transmitter performance parameters to consider is the shape of the leading and trailing edges of pulses, which, in class C transmitters, is controlled by the rise and fall times.

4.3.2.4 Spectrum differences between currently deployed TWT driven vacuum tubes and solid-state transmitters As previously stated, although there remain a number of magnetron based ATC radar, these are being phased out and the majority of deployed ATC TMA radar in the UK use either TWT (BAE SYSTEMS Watchman) or Silicon Bipolar Solid State transmitters (Raytheon ASR-10SS).

While these radar are used to fulfil the same role, their designs and technology limitations have lead to different implementations.

The TWT Watchman was designed to meet both military and civil ATC requirements (some military requirements exceed the civil requirements). In order to meet these detection and accuracy requirements the Watchman TWT system uses a transmitted bandwidth of 2.5MHz, resulting in a range resolution of 60m. This is achieved through the use of both 20µs long NLFM modulated pulses for long range detection and 0.4µs unmodulated short pulses for short range fill in detections. The small range cell size results in good range accuracy and performance in clutter (since a radar’s performance in cluttered environments improves with reduced range cell size). These pulse regimes are transmitted on two separate frequencies at different times to achieve the required frequency diversity against varying target returns.


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Figure 4.3-21 : Measured spectrum of a dual frequency diversity TWT based ATC system [1].

The Raytheon ASR-10SS employs a silicon bipolar solid state bulk transmitter and has a transmitted bandwidth of 1MHz using 50µs and 100µs chirp modulated long range pulses and 1µs short range pulses, resulting in a range resolution of 150m. These pulse regimes are transmitted on four separate frequencies (two pairs) at different times to achieve the required frequency diversity against varying target returns.

Figure 4.3-22 : Measured spectrum (Lower two frequencies) of S band civil ATC radar using various measurement bandwidths (RBw) [1].

Since the deployed ASR-10SS solid state systems use 1MHz transmitted bandwidth, compared to the 2.5MHz of the Watchman radar, these solid state systems are more spectrally efficient. In addition, as discussed previously, the rise time of the transmitted


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pulse from a solid-state system is about 5-10 times slower than that from a TWT transmitter. This combination of reduced bandwidth, slow rise time and longer pulses in the solid-state radar systems make them potentially much more spectrally efficient than the TWT systems.

It is worth noting, however, that solid-state radar systems do still need to make use of a short unmodulated pulse for short range detection, providing a time-averaged spectrum that is the composite spectrum of the long and short pulse. In addition the ASR-10SS implementation uses four frequencies to achieve frequency diversity against target variability rather than two, thereby somewhat increasing their spectral usage.

4.3.2.5 Summary As described above the main problem of using coded waveforms with radar transmitters is an issue of linearity. The above section has shown that current radar transmitter technologies capable of the high powers required for radar applications typically operate in class C configuration, which is a highly non-linear mode. Therefore many of the coding techniques proposed in previous studies, which rely on amplitude shaping and thus require linear operation, could not be used with existing transmitters simply because the coded waveforms would be corrupted. In addition, it would be close to impossible to design a linear radar transmitter capable of amplifying these coded pulses using current radar transmitter devices due to their very narrow range of linear operation.

Currently the best spectral efficiency is achieved by the newer solid state systems. These use long frequency modulated pulses, have slow rise and fall times and approach the best spectral efficiency that is achievable using current transmitter technology.

4.3.3 Review of proposed new techniques and their applicability to practical radar systems The 2004 study ‘Project AY4490 – A study into techniques for improving radar spectrum utilisation’ prepared for Ofcom by Qinetiq [2] documented many possible coded waveform techniques. As discussed above, many of the coded waveform techniques mentioned would simply not be possible to implement on existing radar systems. This is because of the non-linear nature of existing radar transmitters. Some of the ideas may be implemented on new radar systems. The following section revisits some of the ideas proposed in the Qinetiq study that might possibly be implemented on current or new radar systems.

Coded waveforms under consideration which reduce radar spectral bandwidth fall into two types:

Those that can be used to generate a radar-pulse with the minimum RF bandwidth whilst still meeting the minimum performance requirement of the radar.

Those that allow radars to operate with a waveform that minimises the interference to radars of a like type, thus improving spectral efficiency by reducing the frequency reuse distance.

Coded waveforms that have the potential to be used with current or new radar transmitters generally fall under two categories, these are essentially analogue or digital.


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4.3.3.1 Analogue waveforms

4.3.3.1.1 Linear Frequency Modulation (LFM) This is a very common pulse compression technique used in radars. The pulse waveform is frequency modulated with a linear frequency slope that changes in frequency from f1 to f2 (where the Bandwidth, BW = f2 – f1) over the duration of the pulse, this is commonly known as a chirp, Figure 4.3-23.

The frequency-modulated waveform is generated at low power by analogue or digital means and amplified by the radar transmitter. On reception the echo pulse is passed through a pulse compression filter. This effectively trades the bandwidth of the signal for processing gain and resolution with a resulting compressed pulse width of 1/BW. This processed pulse now effectively has the same range resolution as a shorter pulse of length 1/BW.

A limitation of this technique is that moving target echos are Doppler frequency shifted by an amount proportional to their speed relative to the radar, which leads to systematic errors in the calculated range of these targets. This range-Doppler coupling, however, is well understood and can be relatively easily corrected in practical systems.

Figure 4.3-23 : Linear Freqeuncy modulated pulse (LFM Chirp)


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Figure 4.3-24 : Compressed LFM pulse with sidelobes

LFM matched filtering results in relatively large time sidelobes (-13.2dB) that can be mistaken for target echoes or mask nearby weaker targets. In order to reduce these sidelobe levels, amplitude weighting of the received pulses is carried out before pulse compression. However, since it is not possible to apply the same amplitude weighting to the transmitted pulse due to non-linear transmitter operation, this does result in a mismatch process and a reduction in achievable SNR. This loss is traded against improved sidelobe levels.

4.3.3.1.2 Nonlinear Frequency Modulation (NLFM) NLFM is a technique where frequency is varied according to a non-linear function across the pulse, as shown in Figure 4.3-25. Less time is spent over some parts of the spectrum than others, reducing range sidelobes without amplitude weighting. This allows efficient generation of high power, non-distorted, waveforms by a class C transmitter. The pulse compression filter, now also a matched filter, has no mismatch losses. With careful selection of frequency modulation, low sidelobes of -35 to -40 dB or better can be obtained. NLFM waveforms also suffer range-Doppler coupling, but this is relatively easily corrected.


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Figure 4.3-25 : Nonlinear Freqeuncy modulated pulse (NLFM Chirp) with 40dB Taylor weighting

Figure 4.3-26 : Compressed NLFM pulse with reduced range sidelobes

4.3.3.1.3 Stepped Frequency Pulse Modulation An alternative method of generating the required resolution bandwidth is to transmit a train of separate unmodulated pulses, each at sequential frequencies spanning the required bandwidth. For the same pulse length, detection and resolution performance, this technique does not degrade so significantly at short ranges, when the target echo pulses are eclipsed, as for long FM modulated pulses or even the same length plain short pulses.

This means that potentially both long and short range detection, with sufficient range resolution, could be achieved using a sequence of stepped frequency pulses of a longer length than the typical ATC short range pulse. This may give some spectral efficiency benefits compared to the short unmodulated pulses typically used by ATC radar to enable


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short range detection, although the composite spectrum of the entire sequence is unlikely to be better than that of the long FM modulated pulses typically used for long range detection.

The major limitation of this technique is that in order to achieve the required maximum range and range resolution, the target must be illuminated with sufficient stepped frequency pulses to span the resolution bandwidth. With a typical ATC radar update period and beam width there may not be enough time to achieve this. In addition, by maintaining the same PRF between stepped pulses, the repeat period of the entire sequence is much longer, which introduces Doppler ambiguities and will affect the operation of Moving Target Filters in the radar system.

These limitations can be mitigated by reducing the pulse-to-pulse interval to allow more pulses on the target in a shorter repeat period, but this then leads to range ambiguities.

An additional limitation is that, at ATC radar frequencies, this technique also suffers significant range-Doppler coupling where targets with significant velocities are reported at the incorrect range. For this technique this effect is not easily corrected.

4.3.3.1.4 Analogue waveform summary Analogue LFM and NLFM waveforms are widely deployed in current radar systems. Among the reasons for this is that they offer very good performance against radar requirements for detection and range resolution, are relatively easy to generate, do not require linear transmitter devices and their properties are very well understood. In particular, while non-symmetrical FM chirp waveforms do suffer from a range-Doppler coupling, where moving target Doppler can result in range errors, this is relatively easily corrected.

In terms of their potential to interfere with neighbouring systems (radar and/or communications) there is a question whether it is preferable to suffer interference from short pulses with high peak powers, which increase the false alarm rate (but in radar may be mitigated with Pulse Pattern Correlators) or long pulses with lower peak powers which do not cause false alarms but do increase the effective system noise floor and thus desensitise radar over long ranges.

For typical ATC radar systems, which use long chirp modulated pulses, the issue is to use the minimum spectral extent in order to meet the system requirements.

In terms of spectral efficiency, the spectrum of these waveforms is a function of the resolution bandwidth, the pulse length and the transmitter characteristics. In general longer pulses are the most spectrally efficient since the proportion of the pulse occupied by the fast rising and falling edges is minimised. As described in section 4.3.1.1, such long pulses also require less peak power. The only limitation is the need for short unmodulated pulses to allow detection at the short ranges eclipsed by the long modulated pulses, however, these pulses are very short duration and do not have larger peak powers thus do not carry a lot of energy.

At the moment, given the limits of current transmitter technology e.g.

Non-linear, class C operation

Maximum duty cycle ~10%


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The long NLFM chirp pulses utilised in modern ATC radar systems are close to the best spectral efficiency possible in current radar systems. Any further improvements would require significant changes to radar transmitter technology.

4.3.3.2 Digitally coded waveforms Digitally coded pulse waveforms divide a long radar pulse, length T, into N sub-pulses, length τ, with each sub-pulse having a phase selected according to a (often pseudo-random) sequence. Since rate of change of phase with time equates to frequency, a resolution bandwidth increase is achieved. In general these are known as polyphase codes, however the special case of 2 phases (known as biphase or binary phase shift keyed (BPSK)) sequences and quadriphase (four phase states) are most commonly proposed for radar.

The pulse compression ratio for these waveforms is equal to N = T/τ. The power spectrum of such a random sequence, however, is the Fourier transform of its autocorrelation function and is approximately equal to that of a single rectangular segment τ [16].

Digitally coded waveforms have found wide use in communications applications. A selection of these waveforms, whose application to radar has been previously proposed, are discussed and summarised below with attention being paid to any radar performance or spectral efficiency issues:

4.3.3.2.1 Spectrally compact waveforms Coded waveforms exist which are much more spectrally compact than those already mentioned above for pulse compression radars. To maintain range resolution they are designed to contain all energy between the 3dB bandwidth. Descriptions of some of these types of waveforms follow below:

Constrained FM

Constrained FM uses a technique where the correct quadratic phase profile for a LFM waveform is generated in the frequency domain. The resulting spectrum is a constrained rectangular spectrum in the frequency domain; conversion to the time domain is performed for transmission. However this results in an amplitude shaped pulse in the time domain. Radar transmitters are generally class C devices and operate in saturation. This would cause distortion of the time domain signal and hence cause spectrum spread in the frequency domain.

Taylor quadriphase

Taylor quadriphase is a four-phase-coded pulse modulation that spectrally improves performance over normal binary phase coded waveforms. Quadiphase codes consist of two binary phase coded signals used in quadrature. The sub-pulses used are not rectangular, but half cosines at twice the width of the normal sub-pulse interval. This gives a phase change between sub-pulses of only ±π/2 radians. The sub-pulse width τ is measured from the half-power points of the half-cosine. The overlap of the cosine sub-pulses at half power points results in an uncompressed pulse that has constant amplitude except for the leading and trailing edges. The spectrum is more efficient than using rectangular sub-pulses


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because of the elimination of phase transients between the sub-pulses. The phase profile is now continuous. However, in order to maintain the amplitude shaping of the leading and trailing edges, very linear amplifiers are required. In addition the slow rise and fall times created from this type of scheme are liable to cause range accuracy problems.

4.3.3.2.2 Orthogonal and Closely Spaced waveforms Digital codes can be used to increase the orthogonal property between two radars. This is the ability for the radar to reject codes from nearby radar, i.e. the interfering signal does not correlate with its own wanted signals when the two radars are operating on the same carrier frequency. Examples of these orthogonal waveforms are maximum length sequences, Costa codes, LFM and Polyphase codes (Note the LFM waveforms here assume a pair of opposing frequency gradient LFM waveforms).

As indicated, orthogonal waveforms assume operation on identical centre frequencies and rely on their low code cross-correlation for interference rejection. This means that orthogonal interfering energy is distributed over many range cells whereas the wanted signal is concentrated in a single range cell. This reduces false alarm problems from neighbouring radar since the peak interference is not large. However, although the interfering energy is distributed over many range cells, this only serves to raise the effective noise floor of the radar detection process, which in turn desensitises the radar to wanted targets. Such interference can be more destructive than false alarms since the radar operator will be unaware of the problem and cannot compensate. Therefore, whilst these orthogonal low cross-correlation waveforms do provide a significant amount of isolation between waveforms their cross-correlation sidelobes have been found to not meet radar interference rejection requirements.

Further isolation can be obtained by putting spectrum distance between the waveform centre frequencies to generate a pair of closely spaced waveforms. As the frequency distance between centre frequencies increases then the correlation falls off. However, if the frequency distance between centre frequencies increases to more then the range resolution bandwidth then the correlation performance of identical and orthogonal waveforms become virtually indistinguishable and there is no benefit to using them. Orthogonal waveforms are thus very unlikely to be used for radar applications because there is virtually no performance increase over a pair of identical frequency spaced waveforms given the significant increase in complexity of using these waveforms with radar.

4.3.3.2.3 Digitally coded waveform summary As discussed above, there are a number of potential digitally coded waveform designs that have been proposed for use in radar. These waveforms, however, do suffer a number of drawbacks which had lead to them not being widely deployed in practical radar systems:

Doppler intolerance All the biphase codes share the common quality that they are sensitive to Doppler shift

generally showing significant degradation in performance (both detection, range resolution and accuracy).


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Certain polyphase codes do exhibit better Doppler tolerance for a broader range-Doppler coverage than do the biphase codes, and they can also exhibit relatively good sidelobe characteristics [16]. These Doppler tolerant polyphase codes typically resemble discrete approximations to analogue FM chirp waveforms.

Spectral efficiency Since the carrier frequency of radar is not necessarily fixed and is not usually a multiple

of 1/τ, a phase coded signal in general is discontinuous at the sub-pulse boundaries, where the phase change points are. These discontinuities can lead to spreading in the frequency domain.

The power spectrum of a random sequence is the Fourier transform of its autocorrelation function and is approximately equal to that of a single rectangular segment τ.

While orthogonal waveforms could in principle allow spectral re-use, their cross correlation sidelobes are still not sufficiently small to avoid radar desensitisation.

There appears to be no spectral efficiency benefit unless other techniques, such as Taylor quadriphase, a form of pulse shaping, requiring very linear transmitters are applied.

4.3.3.3 Summary Analogue LFM and NLFM waveforms are widely deployed in current radar systems. Among the reasons for this is that they offer very good performance against radar requirements for detection, range resolution and accuracy, are relatively easy to generate, do not require linear transmitter devices and their properties are very well understood.

As discussed, many of the digital phase codes are sensitive to Doppler shift generally showing significant degradation in performance (both detection and range resolution). Those waveforms that are less Doppler intolerant typically resemble discrete approximations to analogue FM chirp waveforms. In addition there appears to be very little if any spectral efficiency benefit unless more complex techniques are implemented, such as Taylor quadriphase, which require very linear transmitters and significantly more complex and costly radar systems.

For modern ATC radar systems, which use long analogue FM pulses, the issue is to use only the minimum spectral extent in order to meet the system requirements.

At the moment, given the limits of current transmitter technology (e.g. Non-linear, class C operation and Maximum duty cycle ~10%); the long NLFM chirp pulses utilised in the newest ATC radar systems are close to the best spectral efficiency possible in current radar systems.

Any further improvements would require changes to radar transmitter technology to allow more linear operation where certain amounts of pulse edge shaping would be possible.


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For existing deployed radar, however, there may be some opportunities, within the limits of their existing transmitter technology, to improve their spectral efficiency by adopting longer pulses and/or reduced resolution bandwidth transmissions which meet minimum requirements for ATC performance. However, it is important to note that, as mentioned in section 4.1, the benefits of improved individual radar spectrum efficiency will only be realised if the frequency planning of the whole radar network is optimised.

4.3.4 Study into the practical application of potentially productive approaches identified This report has examined a variety of coded waveforms and techniques that could be used to improve radar spectral efficiency. However, many of these techniques would be very difficult to implement on existing radar transmitters and would require significant changes to the transmitter technology.

This section will look at which techniques could be implemented on existing radar transmitters and also which techniques could be designed into new radar systems. Current NATS radars are slowly being replaced by solid-state systems that are much more spectrally efficient than TWT based radar. However, there is still a large number of deployed TWT radar in the UK (both at regional airports and in the MoD). Therefore a method of improving the spectral efficiency of these existing TWT based radar systems could lead to significant improvements if it allows more efficient frequency planning across the radar network.

One solution towards better spectral efficiency for existing TWT based radar (e.g. the Watchman radar) is to change the pulse length. There is scope within the NATS criteria for the Watchman radar to operate with a longer pulse length and still meet range resolution and range accuracy requirements. Therefore it is proposed to increase the Watchman pulse lengths to improve spectral efficiency. This idea is explored below.

4.3.4.1 Options for current radar systems

4.3.4.1.1 Reducing spectrum spread from Watchman radar The Watchman radar, which is widely deployed in regional airports and across numerous MoD sites, is the main candidate for spectral efficiency improvements.

In its current form it uses two pulse lengths, one for short range (0.4µs) and one for long range (20µs). The short-range pulse is not modulated whilst the long-range pulse has a 2.5MHz NLFM chirp. The long-range pulse is then compressed to 0.4µs in the radar receiver resulting in the same range resolution of 60m as the short range pulse. Two different pulses are needed so the Watchman radar can achieve radar coverage from 0.25 nautical miles to over 76 nautical miles with a probability of detection of 80%. Both the short range and long-range pulses have been modelled to show the theoretical spectrum spread. One-sided spectra are illustrated in Figure 4.3-27 and Figure 4.3-28 below; the red grid lines indicate the spectral width at the -20, -40 and -60dB points respectively. Note these pulse spectra are theoretical, although there has been consideration of the fast rising and falling edges generated by a TWT transmitter; in reality, after being amplified by a practical radar transmitter, the spectrum spread will be worse than these illustrations.


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6.523 16.91 61.77-120

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Figure 4.3-27 : Theoretical spectrum of 0.4µs plain pulse with 35ns rise&fall time.

1.625 6.977 18.42 63.18-120

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Figure 4.3-28 : Theoretical spectrum of 20µs NLFM 2.5MHz pulse with 35ns rise&fall time.

Comparing Figure 4.3-27 and Figure 4.3-28 reveals that the most spectrally inefficient radar pulse from the Watchman radar is the 0.4µs short-range pulse. Comparing the spectrum spread at the -60dB points shows a two-sided spectrum spread of 36.84MHz for the 20µs pulse compared with 123.54MHz for the 0.4µs pulse. As previously mentioned another consideration is that the longer pulse is on for much longer than the short-range pulse, so although the longer pulse is more spectrally efficient it could be interfering for longer. The peak power output is the same in each case.

The Watchman radar meets the detection criteria for NATS with ease under its current implementation. A solution to reduce the spectrum spread would be to increase the length of the short-range pulse. If the short-range pulse is increased to 1µs then the short-range pulse spectrum would reduce while the probability of detection, range resolution and accuracy could remain within the NATS limits. The chirp on the long-range pulse could also be reduced to 1MHz so that it compresses to the same pulse length as the short-range pulse and can therefore be processed by the same radar receiver processor. In terms of clutter performance and range accuracy, a resolution bandwidth of 1MHz giving a range cell size of 150m is close to the maximum acceptable for ATC applications.

2.67 14.52 39.64-120

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Figure 4.3-29 : Theoretical spectrum of 1µs plain pulse with 35ns rise&fall time.

Figure 4.3-29 illustrates the one sided spectrum spread for the Watchman radar short-range pulse when it has been increased to 1µs. The two sided spectrum spread at the -60dB bandwidth is now 79.28MHz. This is a 35.8% reduction in spectrum spread for the 1µs pulse when compared to the 0.4µs pulse at -60dB.


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Figure 4.3-30 below shows the result for the Watchman radar long-range pulse when the chirp has been reduced to a bandwidth of 1MHz. The spectrum spread at the -60dB point is now 33.35Hz. This is a 9.5% reduction in spectrum spread at the -60dB point compared to the 2.5MHz chirp.

0.7286 4.726 16.67 59.62-120

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Figure 4.3-30 : Theoretical spectrum of 20µs NLFM 1.0MHz pulse with 35ns rise&fall time.

For comparison, the theoretical spectral spreads for all four of the current and proposed Watchman pulse waveforms are tabulated in Table 4.3-3 and the spectral envelopes are illustrated in Figure 4.3-31, the red line indicates the ITU specification for the 0.4µs pulse with a 20dB per decade roll off, which the system is currently configured to meet.

Table 4.3-3 : Two sided spectral spreads in MHz at the various roll off points

Spectral width (MHz) at:

-20dB -40dB -60dB -80dB -100dB

Watchman 0.4µs plain pulse 13.05 33.81 123.54 492.68 499.45

Watchman 20µs NLFM 2.5MHz pulse 3.25 13.95 36.85 126.35 492.65

Watchman 1µs plain pulse 5.34 29.04 79.28 345.15 499.37

Watchman 20µs NLFM 1.0MHz pulse 1.46 9.45 33.35 119.25 487.05


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ITU mask for 0.4usWatchman 0.4us plain pulseWatchman 20us NLFM 2.5MHz pulseWatchman 1us plain pulseWatchman 20us NLFM 1.0MHz pulse

Figure 4.3-31 : Theoretical spectral envelopes of the current and proposed Watchman pulse waveforms

It is clear that the 0.4µs short pulse is indeed the least spectrally efficient while the 20µs 1MHz NLFM pulse is spectrally the narrowest. It is worth noting that these pulses all have the same peak power.

Radar spectral separation. While the spectral envelope of a waveform is an indicator of its spectral efficiency, it does not take into account the transmitted power and thus does not indicate the waveform’s interference potential.

The best metric for estimating spectral efficiency in this case is the frequency separation between neighbouring radar required to avoid interference with each other. If this frequency separation is significantly reduced by adopting new waveforms then this is an indication that the radar network could be re-planned to take advantage of these smaller spectral separations and thus occupy less overall spectrum.

A received signal level of 6dB below a radar’s noise floor has been proposed as sufficiently small to declare a signal as non-interfering. This is based on the effect of such a received signal on the Probability of Detection (PD) of the radar, which is reduced by ~1-2% at its maximum target range. This ratio of acceptable interference to noise is termed the ‘Protection Ratio’ and the actual signal level this corresponds to; the ‘Protection Level’

If the scenario of two neighbouring radar, each at height 10m, separated by range R, using identical waveforms is considered. The received signal at each radar directly from it’s neighbour may be calculated from the one-way radar equation :


4 Detailed Work Packages - Ofcom · The majority of S-band radars are in the frequency range 2.7 to...

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