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FEATURES

1

2

3

4

8

7

6

5

SHUTDOWNBYPASS

IN+IN-

VO-GNDVDD

VO+

D OR DGN PACKAGE(TOP VIEW)

DESCRIPTION

Audio Input

BiasControl

VDD

350 mW

6

5

7

VO+

VDD

1

2

4

BYPASS

IN -

VDD/2

CI

RI

CS

1 µF

CB

0.1 µF

RF

SHUTDOWN

VO- 8

GND

From System Control

3 IN+

-

+

-

+

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

350-mW MONO AUDIO POWER AMPLIFIER

• Fully Specified for 3.3-V and 5-V Operation• Wide Power Supply Compatibility 2.5 V - 5.5 V• Output Power for RL = 8 Ω

– 350 mW at VDD = 5 V, BTL– 250 mW at VDD = 3.3 V, BTL

• Ultra-Low Quiescent Current in ShutdownMode . . . 0.15 µA

• Thermal and Short-Circuit Protection• Surface-Mount Packaging

– SOIC– PowerPAD™ MSOP

The TPA301 is a bridge-tied load (BTL) audio power amplifier developed especially for low-voltage applicationswhere internal speakers are required. Operating with a 3.3-V supply, the TPA301 can deliver 250-mW ofcontinuous power into a BTL 8-Ω load at less than 1% THD+N throughout voice band frequencies. Although thisdevice is characterized out to 20 kHz, its operation was optimized for narrower band applications such as cellularcommunications. The BTL configuration eliminates the need for external coupling capacitors on the output inmost applications, which is particularly important for small battery-powered equipment. This device features ashutdown mode for power-sensitive applications with a quiescent current of 0.15 µA during shutdown. TheTPA301 is available in an 8-pin SOIC surface-mount package and the surface-mount PowerPAD MSOP, whichreduces board space by 50% and height by 40%.

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of TexasInstruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments.

PRODUCTION DATA information is current as of publication date. Copyright © –2004, Texas Instruments IncorporatedProducts conform to specifications per the terms of the TexasInstruments standard warranty. Production processing does notnecessarily include testing of all parameters.

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ABSOLUTE MAXIMUM RATINGS

DISSIPATION RATING TABLE

RECOMMENDED OPERATING CONDITIONS

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integratedcircuits be handled with appropriate precautions. Failure to observe proper handling and installationprocedures can cause damage.

ESD damage can range from subtle performance degradation to complete device failure. Precisionintegrated circuits may be more susceptible to damage because very small parametric changes couldcause the device not to meet its published specifications.

AVAILABLE OPTIONS

PACKAGED DEVICESMSOPTA SMALL OUTLINE (1) MSOP (1)

SYMBOLIZATION(D) (DGN)

–40°C to 85°C TPA301D TPA301DGN AAA

(1) The D and DGN packages are available taped and reeled. To order a taped and reeled part, add thesuffix R to the part number (e.g., TPA301DR).

over operating free-air temperature range (unless otherwise noted) (1)

UNIT

VDD Supply voltage 6 V

VI Input voltage –0.3 V to VDD +0.3 V

Continuous total power dissipation Internally limited (see Dissipation Rating Table)

TA Operating free-air temperature range –40°C to 85°C

TJ Operating junction temperature range –40°C to 150°C

Tstg Storage temperature range –65°C to 150°C

Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds 260°C

(1) Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratingsonly, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operatingconditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

PACKAGE TA ≤ 25°C DERATING FACTOR TA = 70°C TA = 85°C

D 725 mW 5.8 mW/°C 464 mW 377 mW

DGN 2.14 W (1) 17.1 mW/°C 1.37 W 1.11 W

(1) See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report(SLMA002), for more information on the PowerPAD package. The thermal data was measured on aPCB layout based on the information in the section entitled Texas Instruments Recommended Boardfor PowerPAD™ on page 33 of the before mentioned document.

MIN MAX UNIT

VDD Supply voltage 2.5 5.5 V

VIH High-level voltage SHUTDOWN 0.9 VDD V

VIL Low-level voltage SHUTDOWN 0.1 VDD V

TA Operating free-air temperature –40 85 °C

2

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ELECTRICAL CHARACTERISTICS

OPERATING CHARACTERISTICS

ELECTRICAL CHARACTERISTICS

OPERATING CHARACTERISTICS

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

at specified free-air temperature, VDD = 3.3 V, TA = 25°C (unless otherwise noted)

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

|VOD| Differential output voltage SHUTDOWN = 0 V, RL = 8 Ω, RF = 10 kΩ 5 20 mV

PSRR Power supply rejection ratio VDD = 3.2 V to 3.4 V 85 dB

IDD Supply current (see Figure 3) SHUTDOWN = 0 V, RF = 10 kΩ 0.7 1.5 mA

IDD(SD) Supply current, shutdown mode (see Figure 4) SHUTDOWN = VDD, RF = 10 kΩ 0.15 5 µA

|IIH| High-level input current SHUTDOWN, VDD = 3.3 V, VI = 3.3 V 1 µA

|IIL| Low-level input current SHUTDOWN, VDD = 3.3 V, VI = 0 V 1 µA

VDD = 3.3 V, TA = 25°C, RL = 8 Ω

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

PO Output power (1) THD = 0.5%, See Figure 9 250 mW

PO = 250 mW, f = 20 Hz to 4 kHz,THD + N Total harmonic distortion plus noise 1.3%AV = 2 V/V, See Figure 7

Maximum output power bandwidth AV = 2 V/V, THD = 3%, See Figure 7 10 kHz

B1 Unity-gain bandwidth Open loop, See Figure 15 1.4 MHz

Supply ripple rejection ratio f = 1 kHz, CB = 1 µF, See Figure 2 71 dB

AV = 1 V/V, CB = 0.1 µF,Vn Noise output voltage 15 µV(rms)RL = 32 Ω, See Figure 19

(1) Output power is measured at the output terminals of the device at f = 1 kHz.

at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise noted)

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

|VOD| Differential output voltage SHUTDOWN = 0 V, RL = 8 Ω, RF = 10 kΩ 5 20 mV

PSRR Power supply rejection ratio VDD = 4.9 V to 5.1 V 78 dB

IDD Supply current (see Figure 3) SHUTDOWN = 0 V, RF = 10 kΩ 0.7 1.5 mA

IDD(SD) Supply current, shutdown mode (see Figure 4) SHUTDOWN = VDD, RF = 10 kΩ 0.15 5 µA

|IIH| High-level input current SHUTDOWN, VDD = 5.5 V, VI = VDD 1 µA

|IIL| Low-level input current SHUTDOWN, VDD = 5.5 V, VI = 0 V 1 µA

VDD = 5 V, TA = 25°C, RL = 8 Ω

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

PO Output power THD = 0.5%, See Figure 13 700 mW

PO = 350 mW, f = 20 Hz to 4 kHz,THD + N Total harmonic distortion plus noise 1%AV = 2 V/V, See Figure 11

Maximum output power bandwidth AV = 2 V/V, THD = 2%, See Figure 11 10 kHz

B1 Unity-gain bandwidth Open loop, See Figure 16 1.4 MHz

Supply ripple rejection ratio f = 1 kHz, CB = 1 µF, See Figure 2 65 dB

AV = 1 V/V, CB = 0.1 µF,Vn Noise output voltage 15 µV(rms)RL = 32 Ω , See Figure 20

3

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PARAMETER MEASUREMENT INFORMATION

Audio Input

BiasControl

VDD

6

5

7

VO+

VDD

1

2

4

BYPASS

IN -

VDD/2

CI

RI

CS

1 µF

CB

0.1 µF

RF

SHUTDOWN

VO- 8

RL = 8 Ω

GND

3 IN+

-

+

-

+

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

Terminal Functions

TERMINALI/O DESCRIPTION

NAME NO.

BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be connected to aBYPASS 2 I 0.1-µF to 1-µF capacitor when used as an audio amplifier.

GND 7 GND is the ground connection.

IN- 4 I IN- is the inverting input. IN- is typically used as the audio input terminal.

IN+ 3 I IN+ is the noninverting input. IN+ is typically tied to the BYPASS terminal.

SHUTDOWN 1 I SHUTDOWN places the entire device in shutdown mode when held high (IDD~ 0.15 µA).

VDD 6 VDD is the supply voltage terminal.

VO+ 5 O VO+ is the positive BTL output.

VO- 8 O VO- is the negative BTL output.

Figure 1. Test Circuit

4

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TYPICAL CHARACTERISTICS

−50

−60

−80

−10020 100 1 k

−30

−20

f − Frequency − Hz

0

10 k 20 k

−10

−40

−70

−90

VDD = 5 V

VDD = 3.3 V

RL = 8 ΩCB = 1 µF

k SV

R−

Su

pp

ly V

olta

ge

Rej

ectio

n R

atio

− d

B

VDD − Supply Voltage − V

1.1

0.7

0.3

−0.1

0.9

0.5

0.1

3 4 62 5

I DD

(q)−

Su

pp

ly C

urr

ent −

mA

SHUTDOWN = 0 VRF = 10 kΩ

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

Table of Graphs

FIGURE

kSVR Supply voltage rejection ratio vs Frequency 2

IDD Supply current vs Supply voltage 3, 4

vs Supply voltage 5PO Output power

vs Load resistance 6

vs Frequency 7, 8, 11, 12THD+N Total harmonic distortion plus noise

vs Output power 9, 10, 13, 14

Open-loop gain and phase vs Frequency 15, 16

Closed-loop gain and phase vs Frequency 17, 18

Vn Output noise voltage vs Frequency 19, 20

PD Power dissipation vs Output power 21, 22

SUPPLY VOLTAGE REJECTION RATIO SUPPLY CURRENTvs vs

FREQUENCY SUPPLY VOLTAGE

Figure 2. Figure 3.

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VDD − Supply Voltage − V

0.15

0.1

0.053 43.5 4.5

0.35

2 5

0.2

0.25

0.3

5.52.5

0.4

0.45

0.5

I DD

(SD

)− S

up

ply

Cu

rren

t −

Aµ

SHUTDOWN = VDDRF = 10 kΩ

VDD − Supply Voltage − V

600

400

200

02.5 3.53 4 5.5

1000

2

P

4.5 5

O−

Ou

tpu

t Po

wer

− m

W

800

THD+N 1%

RL = 32 Ω

RL = 8 Ω

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

SUPPLY CURRENT (SHUTDOWN)vs

SUPPLY VOLTAGE

Figure 4.

OUTPUT POWERvs

SUPPLY VOLTAGE

Figure 5.

6

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RL − Load Resistance − Ω

300

200

100

016 3224 40 64

800

8

P

48 56

O−

Ou

tpu

t Po

wer

− m

W400

THD+N = 1%

VDD = 5 V

500

600

VDD = 3.3 V

700

f − Frequency − Hz

TH

D+N

−To

tal H

arm

on

ic D

isto

rtio

n +

No

ise

− % VDD = 3.3 V

PO = 250 mWRL = 8 Ω

20 1k 10k

1

0.01

10

0.1

20k100

AV =− 10 V/V

AV = −20 V/V

AV = −2 V/V

f − Frequency − Hz

TH

D+N

−To

tal H

arm

on

ic D

isto

rtio

n +

No

ise

− % VDD = 3.3 V

RL = 8 ΩAV = −2 V/V

20 1k 10k

1

0.01

10

0.1

20k100

PO = 50 mW

PO = 125 mW

PO = 250 mW

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

OUTPUT POWERvs

LOAD RESISTANCE

Figure 6.

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISEvs vs

FREQUENCY FREQUENCY

Figure 7. Figure 8.

7

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PO − Output Power − WT

HD

+N −

Tota

l Har

mo

nic

Dis

tort

ion

+ N

ois

e −

%

f = 20 HzVDD = 3.3 VRL = 8 ΩAV = −2 V/V

0.01 0.1 1

1

0.01

10

0.1

f = 1 kHz

f = 10 kHz

f = 20 kHz

PO − Output Power − W

TH

D+N

−To

tal H

arm

on

ic D

isto

rtio

n +

No

ise

− %

RL = 8 Ω

0.04 0.1 0.4

1

0.01

10

0.1

0.16 0.22 0.28 0.34

VDD = 3.3 Vf = 1 kHzAV = −2 V/V

f − Frequency − Hz

TH

D+N

−To

tal H

arm

on

ic D

isto

rtio

n +

No

ise

− % VDD = 5 V

PO = 350 mWRL = 8 Ω

20 1k 10k

1

0.01

10

0.1

20k100

AV =− 10 V/V

AV = −20 V/V

AV = −2 V/V

f − Frequency − Hz

TH

D+N

−To

tal H

arm

on

ic D

isto

rtio

n +

No

ise

− % VDD = 5 V

RL = 8 ΩAV = −2 V/V

20 1k 10k

1

0.01

10

0.1

20k100

PO = 50 mW

PO = 175 mW

PO = 350 mW

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISEvs vs

OUTPUT POWER OUTPUT POWER

Figure 9. Figure 10.

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISEvs vs

FREQUENCY FREQUENCY

Figure 11. Figure 12.

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PO − Output Power − WT

HD

+N −

Tota

l Har

mo

nic

Dis

tort

ion

+ N

ois

e −

%

f = 20 Hz

VDD = 5 VRL = 8 ΩAV = −2 V/V

0.01 0.1 1

1

0.01

10

0.1

f = 1 kHz

f = 10 kHz

f = 20 kHz

PO − Output Power − W

0.1 0.25 10.40 0.55 0.70 0.85

TH

D+N

−To

tal H

arm

on

ic D

isto

rtio

n +

No

ise

− %

RL = 8 Ω

VDD = 5 Vf = 1 kHzAV = −2 V/V

1

0.01

10

0.1

10

0

−20

−30

20

30

f − Frequency − kHz

40

−10

180

120

0

−120

−180

VDD = 3.3 VRL = Open

Gain

Phase

60

−60Op

en-L

oo

p G

ain

− d

B

Pha

se −

°

1 101 102 103 104

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISEvs vs

OUTPUT POWER OUTPUT POWER

Figure 13. Figure 14.

OPEN-LOOP GAIN AND PHASEvs

FREQUENCY

Figure 15.

9

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10

0

−20

−301

20

30

f − Frequency − kHz

40

−10

180

120

0

−120

−180

VDD = 5 VRL = Open

Gain

Phase

60

−60Op

en-L

oo

p G

ain

− d

B

Pha

se −

°

101 102 103 104

−0.5

−1

−1.5

−2

f − Frequency − Hz

−0.25

−0.75

−1.25

−1.75

0

0.5

Clo

sed

-Lo

op

Gai

n −

dB 0.25

0.75

130

120

140

Pha

se −

°

150

160

VDD = 3.3 VRL = 8 ΩPO = 0.25 WCI =1 µF

1

170

180

Gain

Phase

101 102 103 104 105 106

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

OPEN-LOOP GAIN AND PHASEvs

FREQUENCY

Figure 16.

CLOSED-LOOP GAIN AND PHASEvs

FREQUENCY

Figure 17.

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−0.5

−1

−1.5

−2

f − Frequency − Hz

−0.25

−0.75

−1.25

−1.75

0

0.5

Clo

sed

-Lo

op

Gai

n −

dB 0.25

0.75

130

120

140

Pha

se −

°

150

160

VDD = 5 VRL = 8 ΩPO = 0.35 WCI =1 µF

1

170

180

Gain

Phase

101 102 103 104 105 106

− O

utp

ut N

ois

e V

olta

ge

−µ

Vn

f − Frequency − Hz

20 1 k 10 k

10

1

100

20 k100

VO BTL

VDD = 3.3 VBW = 22 Hz to 22 kHzRL = 32 ΩCB =0.1 µFAV = −1 V/V

VO+

V(r

ms)

− O

utp

ut N

ois

e V

olta

ge

−µ

Vn

f − Frequency − Hz

20 1 k 10 k

10

1

100

20 k100

VDD = 5 VBW = 22 Hz to 22 kHzRL = 32 ΩCB =0.1 µFAV = −1 V/V

VO BTL

VO+

V(r

ms)

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

CLOSED-LOOP GAIN AND PHASEvs

FREQUENCY

Figure 18.

OUTPUT NOISE VOLTAGE OUTPUT NOISE VOLTAGEvs vs

FREQUENCY FREQUENCY

Figure 19. Figure 20.

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PO − Output Power − mW

200 4000

180

150

120

90

300

PD

− P

ow

er D

issi

pat

ion

− m

W

210

240

270

VDD = 3.3 VRL = 8 Ω

100 300

PO − Output Power − mW

200 600400 8000 1000 1200

VDD = 5 VRL = 8 Ω

400

320

240

160

720

PD

− P

ow

er D

issi

pat

ion

− m

W

480

560

640

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

POWER DISSIPATION POWER DISSIPATIONvs vs

OUTPUT POWER OUTPUT POWER

Figure 21. Figure 22.

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APPLICATION INFORMATION

BRIDGE-TIED LOAD

Power V(RMS)

2

RL

V(RMS) VO(PP)

2 2

(1)

RL 2x VO(PP)

VO(PP)

-VO(PP)

VDD

VDD

f(corner) 1

2RLCC (2)

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

Figure 23 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA301 BTL amplifier consistsof two linear amplifiers driving both ends of the load. There are several potential benefits to this differential driveconfiguration but power to the load should be initially considered. The differential drive to the speaker means thatas one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltageswing on the load as compared to a ground-referenced load. Plugging 2 × VO(PP) into the power equation, wherevoltage is squared, yields 4× the output power from the same supply rail and load impedance (see Equation 1).

Figure 23. Bridge-Tied Load Configuration

In a typical portable handheld equipment sound channel operating at 3.3 V, bridging raises the power into an 8-Ωspeaker from a single-ended (SE, ground reference) limit of 62.5 mW to 250 mW. In sound power that is a 6-dBimprovement—which is loudness that can be heard. In addition to increased power, there are frequencyresponse concerns. Consider the single-supply SE configuration shown in Figure 24. A coupling capacitor isrequired to block the dc offset voltage from reaching the load. These capacitors can be quite large(approximately 33 µF to 1000 µF) so they tend to be expensive, heavy, occupy valuable PCB area, and have theadditional drawback of limiting low-frequency performance of the system. This frequency limiting effect is due tothe high pass filter network created with the speaker impedance and the coupling capacitance and is calculatedwith Equation 2.

For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTLconfiguration cancels the dc offsets, eliminating the need for the blocking capacitors. Low-frequency performanceis then limited only by the input network and speaker response. Cost and PCB space are also minimized byeliminating the bulky coupling capacitor.

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RL

CCVO(PP)

VO(PP)

VDD

-3 dB

fc

BTL AMPLIFIER EFFICIENCY

VL(RMS)

VO IDD

IDD(RMS)

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

APPLICATION INFORMATION (continued)

Figure 24. Single-Ended Configuration and Frequency Response

Increasing power to the load does carry a penalty of increased internal power dissipation. The increaseddissipation is understandable considering that the BTL configuration produces 4× the output power of a SEconfiguration. Internal dissipation versus output power is discussed further in the thermal considerations section.

Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across theoutput stage transistors. There are two components of the internal voltage drop. One is the headroom or dcvoltage drop that varies inversely to output power. The second component is due to the sine-wave nature of theoutput. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from VDD.The internal voltage drop multiplied by the RMS value of the supply current, IDD(RMS), determines the internalpower dissipation of the amplifier.

An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the powersupply to the power delivered to the load. To accurately calculate the RMS values of power in the load and in theamplifier, the current and voltage waveform shapes must first be understood (see Figure 25).

Figure 25. Voltage and Current Waveforms for BTL Amplifiers

Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply aredifferent between SE and BTL configurations. In an SE application the current waveform is a half-wave rectifiedshape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, whichsupports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.The following equations are the basis for calculating amplifier efficiency.

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IDDRMS

2VP RL

PSUP VDD IDDRMS

VDD 2VP RL

Efficiency PL

PSUP

where

PL V

LRMS2

RL

Vp2

2RL

VLRMS

VP

2

(3)

Efficiency of a BTL Configuration VP2VDD

PLRL212

2VDD (4)

APPLICATION SCHEMATIC

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

APPLICATION INFORMATION (continued)

Table 1 employs Equation 4 to calculate efficiencies for three different output power levels. The efficiency of theamplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting in anearly flat internal power dissipation over the normal operating range. The internal dissipation at full output poweris less than in the half power range. Calculating the efficiency for a specific system is the key to proper powersupply design.

Table 1. Efficiency vs Output Power in 3.3-V 8-Ω BTL Systems

PEAK-to-PEAK INTERNALOUTPUT POWER EFFICIENCY VOLTAGE DISSIPATION(W) (%) (V) (W)

0.125 33.6 1.41 0.26

0.25 47.6 2.00 0.29

0.375 58.3 2.45 (1) 0.28

(1) High-peak voltage values cause the THD to increase.

A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in theefficiency equation to utmost advantage when possible. Note that in Equation 4, VDD is in the denominator. Thisindicates that as VDD goes down, efficiency goes up.

Figure 26 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of –10 V/V.

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Audio Input

BiasControl

VDD

350 mW

6

5

7

VO+

VDD

1

2

4

BYPASS

IN -

VDD/2 CS

1 µF

CB

2.2 µF

SHUTDOWN

VO- 8

GND

From System Control

3 IN+

-

+

-

+

CI0.47 µF

RI10 kΩ

RF50 kΩ

CF5 pF

COMPONENT SELECTION

Gain Setting Resistors, RF and RI

BTL Gain AV 2RFRI

(5)

Effective Impedance

RFRIRF RI (6)

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

Figure 26. TPA301 Application Circuit

The following sections discuss the selection of the components used in Figure 26.

The gain for each audio input of the TPA301 is set by resistors RF and RI according to Equation 5 for BTL mode.

BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring thevoltage swing across the load. Given that the TPA301 is a MOS amplifier, the input impedance is high;consequently, input leakage currents are not generally a concern, although noise in the circuit increases as thevalue of RF increases. In addition, a certain range of RF values are required for proper start-up operation of theamplifier. Taken together it is recommended that the effective impedance seen by the inverting node of theamplifier be set between 5 kΩ and 20 kΩ. The effective impedance is calculated in Equation 6.

As an example, consider an input resistance of 10 kΩ and a feedback resistor of 50 kΩ. The BTL gain of theamplifier would be -10 V/V, and the effective impedance at the inverting terminal would be 8.3 kΩ, which is wellwithin the recommended range.

For high-performance applications, metal film resistors are recommended because they tend to have lower noiselevels than carbon resistors. For values of RF above 50 kΩ, the amplifier tends to become unstable due to a poleformed from RF and the inherent input capacitance of the MOS input structure. For this reason, a smallcompensation capacitor, CF, of approximately 5 pF should be placed in parallel with RF when RF is greater than50 kΩ. This, in effect, creates a low-pass filter network with the cutoff frequency defined in Equation 7.

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−3 dB

fco

fco(lowpass) 1

2RFCF

(7)

Input Capacitor, CI

−3 dB

fco

fco(highpass) 1

2RICI

(8)

CI 1

2RIfco (9)

Power Supply Decoupling, CS

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

For example, if RF is 100 kΩ and CF is 5 pF then fco is 318 kHz, which is well outside the audio range.

In the typical application, an input capacitor, CI, is required to allow the amplifier to bias the input signal to theproper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequencydetermined in Equation 8.

The value of CI is important to consider as it directly affects the bass (low-frequency) performance of the circuit.Consider the example where RI is 10 kΩ and the specification calls for a flat bass response down to 40 Hz.Equation 8 is reconfigured as Equation 9.

In this example, CI is 0.40 µF, so, one would likely choose a value in the range of 0.47 µF to 1 µF. A furtherconsideration for this capacitor is the leakage path from the input source through the input network (RI, CI) andthe feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifierthat reduces useful headroom, especially in high gain applications. For this reason, a low-leakage tantalum orceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitorshould face the amplifier input in most applications, as the dc level there is held at VDD/2, which is likely higherthan the source dc level. It is important to confirm the capacitor polarity in the application.

The TPA301 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling toensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also preventsoscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved byusing two capacitors of different types that target different types of noise on the power supply leads. For higherfrequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramiccapacitor, typically 0.1 µF, placed as close as possible to the device VDD lead, works best. For filteringlower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the audiopower amplifier is recommended.

17

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Midrail Bypass Capacitor, CB

10CB 250 kΩ

1RF RI

CI (10)

USING LOW-ESR CAPACITORS

5-V VERSUS 3.3-V OPERATION

HEADROOM AND THERMAL CONSIDERATIONS

PdB 10LogPW 10Log 3500 mW –4.6 dB

TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

The midrail bypass capacitor, CB, is the most critical capacitor and serves several important functions. Duringstart-up or recovery from shutdown mode, CB determines the rate at which the amplifier starts up. The secondfunction is to reduce noise produced by the power supply caused by coupling into the output drive signal. Thisnoise is from the midrail generation circuit internal to the amplifier, appearing as degraded PSRR and THD + N.The capacitor is fed from a 250-kΩ source inside the amplifier. To keep the start-up pop as low as possible, therelationship shown in Equation 10 should be maintained, which ensures the input capacitor is fully chargedbefore the bypass capacitor is fully charged and the amplifier starts up.

As an example, consider a circuit where CB is 2.2 µF, CI is 0.47 µF, RF is 50 kΩ and RI is 10 kΩ. Inserting thesevalues into the Equation 10 we get: 18.2 ≤ 35.5, which satisfies the rule. Recommended values for bypasscapacitor CB are 2.2 µF to 1 µF, ceramic or tantalum low-ESR, for the best THD and noise performance.

Low-ESR capacitors are recommended throughout this application. A real (as opposed to ideal) capacitor can bemodeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes thebeneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance, the more thereal capacitor behaves like an ideal capacitor.

The TPA301 operates over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-Vand 3.3-V operation, as these are considered to be the two most common standard voltages. There are nospecial considerations for 3.3-V versus 5-V operation with respect to supply bypassing, gain setting, or stability.The most important consideration is that of output power. Each amplifier in TPA301 can produce a maximumvoltage swing of VDD– 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V asopposed to VO(PP) = 4 V at 5 V. The reduced voltage swing subsequently reduces maximum output power into an8-Ω load before distortion becomes significant.

Operation from 3.3-V supplies, as can be shown from the efficiency formula in Equation 4, consumesapproximately two-thirds the supply power for a given output-power level than operation from 5-V supplies.

Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortionas compared with the average power output. From the TPA301 data sheet, one can see that when the TPA301is operating from a 5-V supply into a 8-Ω speaker, 350-mW peaks are available. Converting watts to dB:

Subtracting the headroom restriction to obtain the average listening level without distortion yields:

–4.6 dB – 15 dB = –19.6 dB (15 dB headroom)

–4.6 dB – 12 dB = –16.6 dB (12 dB headroom)

–4.6 dB – 9 dB = –13.6 dB (9 dB headroom)

–4.6 dB – 6 dB = –10.6 dB (6 dB headroom)

–4.6 dB – 3 dB = –7.6 dB (3 dB headroom)

18

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TPA301

SLOS208E–JANUARY 1998 –REVISED JUNE 2004

Converting dB back into watts:

PW = 10PdB/10

= 11 mW (15 dB headroom)

= 22 mW (12 dB headroom)

= 44 mW (9 dB headroom)

= 88 mW (6 dB headroom)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for theamplifier system. Comparing the absolute worst case, which is 350 mW of continuous power output with 0 dB ofheadroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for thesystem. Using the power dissipation curves for a 5-V, 8-Ω system, the internal dissipation in the TPA301 andmaximum ambient temperatures is shown in Table 2.

Table 2. TPA301 Power Rating, 5-V, 8-Ω, BTL

MAXIMUM AMBIENTPEAK OUTPUT AVERAGE OUTPUT POWER DISSIPATION TEMPERATUREPOWER POWER (mW)(mW) 0 CFM

350 350 mW 600 46°C

350 175 mW (3 dB) 500 64°C

350 88 mW (6 dB) 380 85°C

350 44 mW (9 dB) 300 98°C

350 22 mW (12 dB) 200 115°C

350 11 mW (15 dB) 180 119°C

Table 2 shows that the TPA301 can be used to its full 350-mW rating without any heat sinking in still air up to46°C.

19

PACKAGE OPTION ADDENDUM

www.ti.com 24-Jan-2013

Addendum-Page 1

PACKAGING INFORMATION

Orderable Device Status(1)

Package Type PackageDrawing

Pins Package Qty Eco Plan(2)

Lead/Ball Finish MSL Peak Temp(3)

Op Temp (°C) Top-Side Markings(4)

Samples

TPA301D ACTIVE SOIC D 8 75 Green (RoHS& no Sb/Br)

CU NIPDAU Level-1-260C-UNLIM 301

TPA301DG4 ACTIVE SOIC D 8 75 Green (RoHS& no Sb/Br)

CU NIPDAU Level-1-260C-UNLIM 301

TPA301DGN ACTIVE MSOP-PowerPAD

DGN 8 80 Green (RoHS& no Sb/Br)

CU NIPDAU Level-1-260C-UNLIM AAA

TPA301DGNG4 ACTIVE MSOP-PowerPAD

DGN 8 80 Green (RoHS& no Sb/Br)

CU NIPDAU Level-1-260C-UNLIM AAA

TPA301DGNR ACTIVE MSOP-PowerPAD

DGN 8 2500 Green (RoHS& no Sb/Br)

CU NIPDAU Level-1-260C-UNLIM AAA

TPA301DGNRG4 ACTIVE MSOP-PowerPAD

DGN 8 2500 Green (RoHS& no Sb/Br)

CU NIPDAU Level-1-260C-UNLIM AAA

TPA301DR ACTIVE SOIC D 8 2500 Green (RoHS& no Sb/Br)

CU NIPDAU Level-1-260C-UNLIM 301

TPA301DRG4 ACTIVE SOIC D 8 2500 Green (RoHS& no Sb/Br)

CU NIPDAU Level-1-260C-UNLIM 301

(1) The marketing status values are defined as follows:ACTIVE: Product device recommended for new designs.LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.PREVIEW: Device has been announced but is not in production. Samples may or may not be available.OBSOLETE: TI has discontinued the production of the device.

(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availabilityinformation and additional product content details.TBD: The Pb-Free/Green conversion plan has not been defined.Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement thatlead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used betweenthe die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weightin homogeneous material)

(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

PACKAGE OPTION ADDENDUM

www.ti.com 24-Jan-2013

Addendum-Page 2

(4) Only one of markings shown within the brackets will appear on the physical device.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on informationprovided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken andcontinues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device PackageType

PackageDrawing

Pins SPQ ReelDiameter

(mm)

ReelWidth

W1 (mm)

A0(mm)

B0(mm)

K0(mm)

P1(mm)

W(mm)

Pin1Quadrant

TPA301DGNR MSOP-Power PAD

DGN 8 2500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

TPA301DGNR MSOP-Power PAD

DGN 8 2500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

TPA301DR SOIC D 8 2500 330.0 12.4 6.4 5.2 2.1 8.0 12.0 Q1

PACKAGE MATERIALS INFORMATION

www.ti.com 14-Jul-2012

Pack Materials-Page 1

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

TPA301DGNR MSOP-PowerPAD DGN 8 2500 358.0 335.0 35.0

TPA301DGNR MSOP-PowerPAD DGN 8 2500 364.0 364.0 27.0

TPA301DR SOIC D 8 2500 367.0 367.0 35.0

PACKAGE MATERIALS INFORMATION

www.ti.com 14-Jul-2012

Pack Materials-Page 2

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