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886 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 2, FEBRUARY 2013

High-Efficiency Single-Input Multiple-OutputDC–DC Converter

Rong-Jong Wai, Senior Member, IEEE, and Kun-Huai Jheng

Abstract—The aim of this study is to develop a high-efficiencysingle-input multiple-output (SIMO) dc–dc converter. The pro-posed converter can boost the voltage of a low-voltage input powersource to a controllable high-voltage dc bus and middle-voltage out-put terminals. The high-voltage dc bus can take as the main powerfor a high-voltage dc load or the front terminal of a dc–ac inverter.Moreover, middle-voltage output terminals can supply powers forindividual middle-voltage dc loads or for charging auxiliary powersources (e.g., battery modules). In this study, a coupled-inductor-based dc–dc converter scheme utilizes only one power switch withthe properties of voltage clamping and soft switching, and the cor-responding device specifications are adequately designed. As a re-sult, the objectives of high-efficiency power conversion, high step-up ratio, and various output voltages with different levels can beobtained. Some experimental results via a kilowatt-level prototypeare given to verify the effectiveness of the proposed SIMO dc–dcconverter in practical applications.

Index Terms—Coupled inductor, high-efficiency power conver-sion, single-input multiple-output (SIMO) converter, soft switch-ing, voltage clamping.

I. INTRODUCTION

IN ORDER to protect the natural environment on the earth,the development of clean energy without pollution has the

major representative role in the last decade [1]–[3]. By dealingwith the issue of global warning, clean energies, such as fuelcell (FC), photovoltaic, and wind energy, etc., have been rapidlypromoted. Due to the electric characteristics of clean energy, thegenerated power is critically affected by the climate or has slowtransient responses, and the output voltage is easily influencedby load variations [4]–[6]. Besides, other auxiliary components,e.g., storage elements, control boards, etc., are usually requiredto ensure the proper operation of clean energy. For example, anFC-generation system is one of the most efficient and effectivesolutions to the environmental pollution problem [7]. In addi-tion to the FC stack itself, some other auxiliary components,such as the balance of plant (BOP) including an electronic con-trol board, an air compressor, and a cooling fan, are requiredfor the normal work of an FC generation system [8], [9]. In

Manuscript received March 19, 2012; revised May 7, 2012; acceptedJune 14, 2012. Date of current version September 27, 2012. This work wassupported in part by the National Science Council of Taiwan under GrantNSC 101-3113-E-155-001. Recommended for publication by Associate EditorM. Vitelli.

The authors are with the Department of Electrical Engineering, YuanZe University, Chung Li 32003, Taiwan (e-mail: [email protected];[email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2012.2205272

other words, the generated power of the FC stack also shouldsatisfy the power demand for the BOP. Thus, various voltagelevels should be required in the power converter of an FC gen-eration system. In general, various single-input single-outputdc–dc converters with different voltage gains are combined tosatisfy the requirement of various voltage levels, so that its sys-tem control is more complicated and the corresponding costis more expensive. The motivation of this study is to designa single-input multiple-output (SIMO) converter for increasingthe conversion efficiency and voltage gain, reducing the controlcomplexity, and saving the manufacturing cost.

Patra et al. [10] presented a SIMO dc–dc converter capableof generating buck, boost, and inverted outputs simultaneously.However, over three switches for one output were required. Thisscheme is only suitable for the low output voltage and powerapplication, and its power conversion is degenerated due to theoperation of hard switching. Nami et al. [11] proposed a newdc–dc multi-output boost converter, which can share its totaloutput between different series of output voltages for low- andhigh-power applications. Unfortunately, over two switches forone output were required, and its control scheme was compli-cated. Besides, the corresponding output power cannot supplyfor individual loads independently. Chen et al. [12] investigateda multiple-output dc–dc converter with shared zero-current-switching (ZCS) lagging leg. Although this converter with thesoft-switching property can reduce the switching losses, thiscombination scheme with three full-bridge converters is morecomplicated, so that the objective of high-efficiency power con-version is difficult to achieve, and its cost is inevitably increased.

This study presents a newly designed SIMO converter witha coupled inductor. The proposed converter uses one powerswitch to achieve the objectives of high-efficiency power con-version, high step-up ratio, and different output voltage levels.In the proposed SIMO converter, the techniques of soft switch-ing and voltage clamping are adopted to reduce the switchingand conduction losses via the utilization of a low-voltage-ratedpower switch with a small RDS(on) . Because the slew rate of thecurrent change in the coupled inductor can be restricted by theleakage inductor, the current transition time enables the powerswitch to turn ON with the ZCS property easily, and the ef-fect of the leakage inductor can alleviate the losses caused bythe reverse-recovery current. Additionally, the problems of thestray inductance energy and reverse-recovery currents withindiodes in the conventional boost converter also can be solved,so that the high-efficiency power conversion can be achieved.The voltages of middle-voltage output terminals can be appro-priately adjusted by the design of auxiliary inductors; the outputvoltage of the high-voltage dc bus can be stably controlled by asimple proportional-integral (PI) control. This study is mainly

0885-8993/$31.00 © 2012 IEEE

WAI AND JHENG: HIGH-EFFICIENCY SINGLE-INPUT MULTIPLE-OUTPUT DC–DC CONVERTER 887

Fig. 1. System configuration of high-efficiency single-input multiple-output(SIMO) converter.

organized into five sections. Following the introduction, the con-verter design and analyses are given in Section II. In Section III,the design considerations of the proposed SIMO converter arediscussed in detail. Section IV provides rich experimental re-sults to validate the effectiveness of the proposed converter inpractical applications. Finally, some conclusions are drawn inSection V.

II. CONVERTER DESIGN AND ANALYSES

The system configuration of the proposed high-efficiencySIMO converter topology to generate two different voltage lev-els from a single-input power source is depicted in Fig. 1. ThisSIMO converter contains five parts including a low-voltage-sidecircuit (LVSC), a clamped circuit, a middle-voltage circuit, anauxiliary circuit, and a high-voltage-side circuit (HVSC). Themajor symbol representations are summarized as follows. VFC(iFC ) and VO1 (iO1) denote the voltages (currents) of the inputpower source and the output load at the LVSC and the auxiliarycircuit, respectively; VO2 and iO2 are the output voltage and cur-rent in the HVSC. CFC , CO1 , and CO2 are the filter capacitorsat the LVSC, the auxiliary circuit, and the HVSC, respectively;C1 and C2 are the clamped and middle-voltage capacitors inthe clamped and middle-voltage circuits, respectively. LP andLS represent individual inductors in the primary and secondarysides of the coupled inductor Tr , respectively, where the primaryside is connected to the input power source; Laux is the auxil-iary circuit inductor. The main switch is expressed as S1 in theLVSC; the equivalent load in the auxiliary circuit is representedas RO 1 , and the output load is represented as RO 2 in the HVSC.

The corresponding equivalent circuit given in Fig. 2 is used todefine the voltage polarities and current directions. The coupledinductor in Fig. 1 can be modeled as an ideal transformer in-cluding the magnetizing inductor Lmp and the leakage inductorLkp in Fig. 2. The turns ratio N and coupling coefficient k of

Fig. 2. Equivalent circuit.

this ideal transformer are defined as

N = N2/N1 (1)

k =Lmp

(Lkp + Lmp)= Lmp/LP (2)

where N1 and N2 are the winding turns in the primary and sec-ondary sides of the coupled inductor Tr . Because the voltagegain is less sensitive to the coupling coefficient and the clampedcapacitor C1 is appropriately selected to completely absorb theleakage inductor energy [13], the coupling coefficient could besimply set at one (k = 1) to obtain Lmp = LP via (2). In thisstudy, the following assumptions are made to simplify the con-verter analyses: 1) The main switch including its body diodeis assumed to be an ideal switching element; and 2) The con-duction voltage drops of the switch and diodes are neglected.

A. Operation Modes

The characteristic waveforms are depicted in Fig. 3, and thetopological modes in one switching cycle are illustrated in Fig. 4.

1) Mode 1 (t0–t1) [Fig. 4(a)]: In this mode, the main switchS1 was turned ON for a span, and the diode D4 turnedOFF. Because the polarity of the windings of the cou-pled inductor Tr is positive, the diode D3 turns ON. Thesecondary current iLs reverses and charges to the middle-voltage capacitor C2 . When the auxiliary inductor Lauxreleases its stored energy completely, and the diode D2turns OFF, this mode ends.

2) Mode 2 (t1–t2) [Fig. 4(b)]: At time t = t1 , the main switchS1 is persistently turned ON. Because the primary inductorLP is charged by the input power source, the magnetiz-ing current iLmp increases gradually in an approximatelylinear way. At the same time, the secondary voltage vLs

charges the middle-voltage capacitor C2 through the diodeD3 . Although the voltage vLmp is equal to the input volt-age VFC both at modes 1 and 2, the ascendant slope ofthe leakage current of the coupled inductor (diLkp /dt) atmodes 1 and 2 is different due to the path of the auxiliarycircuit. Because the auxiliary inductor Laux releases its


Fig. 3. Characteristic waveforms of high-efficiency SIMO converter.

stored energy completely, and the diode D2 turns OFF atthe end of mode 1, it results in the reduction of diLkp /dtat mode 2.

3) Mode 3 (t2–t3) [Fig. 4(c)]: At time t = t2 , the main switchS1 is turned OFF. When the leakage energy still releasedfrom the secondary side of the coupled inductor, the diodeD3 persistently conducts and releases the leakage energyto the middle-voltage capacitor C2 . When the voltageacross the main switch vS 1 is higher than the voltageacross the clamped capacitor VC 1 , the diode D1 conductsto transmit the energy of the primary-side leakage induc-tor Lkp into the clamped capacitor C1 . At the same time,partial energy of the primary-side leakage inductor Lkp istransmitted to the auxiliary inductor Laux , and the diode

D2 conducts. Thus, the current iL aux passes through thediode D2 to supply the power for the output load in theauxiliary circuit. When the secondary side of the coupledinductor releases its leakage energy completely, and thediode D3 turns OFF, this mode ends.

4) Mode 4 (t3–t4) [Fig. 4(d)]: At time t = t3 , the main switchS1 is persistently turned OFF. When the leakage energyhas released from the primary side of the coupled induc-tor, the secondary current iLS is induced in reverse fromthe energy of the magnetizing inductor Lmp through theideal transformer, and flows through the diode D4 to theHVSC. At the same time, partial energy of the primary-side leakage inductor Lkp is still persistently transmittedto the auxiliary inductor Laux , and the diode D2 keeps toconduct. Moreover, the current iL aux passes through thediode D2 to supply the power for the output load in theauxiliary circuit.

5) Mode 5 (t4–t5) [Fig. 4(e)]: At time t = t4 , the main switchS1 is persistently turned OFF, and the clamped diode D1turns OFF because the primary leakage current iLkp equalsto the auxiliary inductor current iL aux . In this mode, theinput power source, the primary winding of the coupledinductor Tr , and the auxiliary inductor Laux connect inseries to supply the power for the output load in the aux-iliary circuit through the diode D2 . At the same time, theinput power source, the secondary winding of the coupledinductor Tr , the clamped capacitor C1 , and the middle-voltage capacitor (C2) connect in series to release theenergy into the HVSC through the diode D4 .

6) Mode 6 (t5–t6) [Fig. 4(f)]: At time t = t5 , this mode beginswhen the main switch S1 is triggered. The auxiliary in-ductor current iL aux needs time to decay to zero, the diodeD2 persistently conducts. In this mode, the input powersource, the clamped capacitor C1 , the secondary windingof the coupled inductor Tr , and the middle-voltage capac-itor C2 still connect in series to release the energy into theHVSC through the diode D4 . Since the clamped diode D1can be selected as a low-voltage Schottky diode, it willbe cut off promptly without a reverse-recovery current.Moreover, the rising rate of the primary current iLkp is lim-ited by the primary-side leakage inductor Lkp . Thus, onecannot derive any currents from the paths of the HVSC,the middle-voltage circuit, the auxiliary circuit, and theclamped circuit. As a result, the main switch S1 is turnedON under the condition of ZCS and this soft-switchingproperty is helpful for alleviating the switching loss. Whenthe secondary current iLS decays to zero, this mode ends.After that, it begins the next switching cycle and repeatsthe operation in mode 1.

Remark 1: In general, a dc–dc converter operated at the con-tinuous conduction mode (CCM) can provide a low ripple cur-rent for protecting the energy source. In the proposed SIMOconverter, it is operated at the CCM due to the design of theauxiliary inductor. The coupled inductor is charged by the in-put power source when the main switch is turned ON, and thecoupled inductor releases its energy to the auxiliary inductorwhen the main switch is turned OFF until the energy balance ofthe coupled inductor and the auxiliary inductor is established.


Fig. 4. Topological modes: (a) Mode 1 [t0−t1 ]; (b) Mode 2 [t1−t2 ]; (c) Mode 3 [t2−t3 ]; (d) Mode 4 [t3−t4 ]; (e) Mode 5 [t4−t5 ]; (f) Mode 6 [t5−t6 ].

As can be seen from Fig. 3, the primary current of thecoupled inductor is positive during one switching cycle. ThisCCM operation is helpful to extend the lifetime of the inputenergy source.

B. Voltage Gain Derivation

Since the magnetizing inductor voltage vLmp is equal to theinput power source VFC at the mode 2, the voltage vLmp can berepresented as

vLmp=VFC . (3)

Due to the relation of vLs=NvLp=VC 2 , the voltage VC 2 canbe represented as

VC 2 = NVFC . (4)

By using the voltage-second balance [14], the relation of theaverage voltage across the magnetizing inductor Lmp of the

coupled inductor Tr to be zero can be represented as

VFCd1TS + vLmp(1 − d1)TS = 0. (5)

From (5), one can obtain

vLmp = [−d1/(1 − d1)]VFC . (6)

Since the voltage of the clamped capacitor VC 1 is equal tothe negative voltage of magnetizing inductors voltage vLmp atmodes 3 and 4, the voltage VC 1 can be expressed via (6) as

VC 1 = −vLmp = [d1/(1 − d1)]VFC . (7)

According to Kirchhoff’s voltage law, the output voltage VO2can be obtained as

VO2 = VFC + VC 1 + VC 2 − vLs. (8)

By using the voltage-second balance [14], the relation of theaverage voltage across the secondary winding vLs to be zero


can be expressed by (4) and (8) as

(NVFC)d1TS + (VFC + VC 1 + VC 2 − VO2)(1 − d1)TS = 0.(9)

From (4)–(9), the voltage gain GVH of the proposed SIMOconverter from the LVSC to the HVSC can be given as

GVH =VO2

VFC=

N + 11 − d1

. (10)

For calculating the discharge time of the auxiliary inductor atmodes 1 and 6, the corresponding time interval can be denotedas dxTs = [(t6 − t5) + (t1 − t0)]. By using the voltage-secondbalance [14], the relation of the average voltage across the aux-iliary inductor Laux to be zero can be represented as

(VFC − vLmp − VO1)(1 − d1)TS + (−VO1)dxTS = 0. (11)

The voltage gain GVL of the proposed SIMO converter fromthe LVSC to the auxiliary circuit can be obtained by (6) and (11)as

GVL =VO1

VFC=

11 − d1 + dx

. (12)

Because the diode current iD2 is equal to the current iLaux ,the average value of the diode current iD2 can be calculatedfrom the third graph in Fig. 3 as

iD2(avg) =1TS

[12iLaux(max)(1 − d1)TS +

12iLaux(max)dxTS

]

(13)where TS is the converter switching cycle, iLaux(max) is themaximum current of the auxiliary inductor and can be expressedas

iLaux(max) =(

VO1

Laux

)dxTS . (14)

By substituting (14) into (13), one can obtain

iD2(avg) =VO1

2LauxdxTS (1 − d1 + dx). (15)

Because the average current of the diode D2 is equal to thecurrent iO1 , it yields

iD2(avg) =VO1

RO1. (16)

From (15) and (16), the duty cycle dx can be rewritten as

dx =−(1 − d1) +

√(1 − d1)2 + [8Laux/(RO1TS )]

2. (17)

By substituting (17) into (12), the voltage gain GVL of theproposed SIMO converter from the LVSC to the auxiliary circuitcan be rearranged as

GVL =VO1

VFC=

2(1 − d1) +

√(1 − d1)2 + [8Laux/(RO1TS )]

.

(18)

III. DESIGN CONSIDERATIONS

In a FC generation system, except for the FC stack itself,some other auxiliary components, such as the BOP includingan electronic control board, an air compressor, and a cooling

Fig. 5. Voltage gain GVH with respect to duty cycle d1 under different turnsratios.

fan, are required. To verify the effectiveness of the proposedSIMO converter topology, a 12-V power supply VFC is takenas the input source to imitate a FC stack, and a 24-V batterymodule is utilized for the output load in the auxiliary circuit.Moreover, the maximum battery floating charge voltage is setat 28 V, and the allowable charging power is 100 W (RO 1 =7.84 Ω). In addition, the desired output voltage VO2 is setat 200 V in the HVSC, and the maximum output powerin the HVSC is 1.1 kW (RO 2 = 36.36 Ω). Define theminimum and maximum output powers in the auxiliarycircuit and the HVSC as (P1 min , P1 max ) and (P2 min ,P2 max ), respectively. In the case of resistive loads, onecan, respectively, obtain the minimum and maximum resis-tances connected at the auxiliary circuit and the HVSC as(RO1 min = G2

VLV 2FC/P1 max , RO1 max = G2

VLV 2FC/P1 min )

and (RO2 min = G2VHV 2

FC/P2 max , RO2 max = G2VHV 2

FC/P2 min ) according to (10) and (12). Furthermore, this con-verter is operated with a 100-kHz switching frequency(fS = 100 kHz), and the coupling coefficient could be simplyset at one (k = 1) because the proposed circuit has a goodclamped effect.

By substituting N = 1 – 7 into (10), the curve of the voltagegain GVH with respect to different duty cycles d1 is depictedin Fig. 5. Moreover, the curve of the voltage gain GVL withrespect to different duty cycles d1 can be represented as Fig. 6by substituting Laux = 1 – 7μH, TS = 10 μs, and RO 1 = 7.84Ω into (18). By analyzing Fig. 5, the turns ratio of the coupledinductor can be selected as N = 5 when the operational condi-tions are VO2 = 200V and VFC = 12 V (i.e., GVH = 16.67),so that the corresponding duty cycle can be obtained as d1 =0.64. This value is reasonable in practical applications. As canbe seen from Fig. 3, the relation of dx < d1 should be satis-fied. According to (17), the limit for Laux can be calculatedas Laux < 0.5d1RO1TS . By considering VFC = 12 V, VO1 =28 V, and d1 = 0.64, the value of the auxiliary inductor can beobtained as Laux = 2 μH from Fig. 6.


Fig. 6. Voltage gain GVL with respect to duty cycle d1 under different auxil-iary inductor values.

Because the voltage across the main switch S1 at the mode 3can be represented as vS1 = VC 1 + VFC = [1/(1 − d1)]VFC ,and the voltage relation between the LVSC and the HVSCaccording to (10) can be given by VFC = [(1 − d1)/(N +1)]VO2 , the voltage across the main switch S1 can berewritten as

vS1 =VO2

(N + 1). (19)

By analyzing (19), the switch voltage vS1 is not related tothe input power source VFC and the duty cycle d1 if the valuesof the output voltage VO2 and the turns ratio N are fixed. Thus,the switch voltage vS1 can be clamped at 33.3 V by substitutingN = 5 and VO2 = 200 V into (19). As long as the input voltagein the LVSC is not higher than the output voltage in the HVSC,the proposed SIMO converter can be applied well to the inputpower source with voltage variations. To consider the ring phe-nomena caused by the circuit stray inductance and the parasiticcapacitance of the main switch S1 , the main switch S1 with a75-V voltage rating is selected in the proposed SIMO converter.

In this study, the clamped diode D1 should be a fast conduc-tive device. Because the clamped voltage of the diode D1 is thesame as the one of the main switch S1 , a low-voltage Schot-tky diode can be adopted to conduct promptly with lower con-duction loss and reverse-recovery current. During the mode 2,the diode D2 turns OFF, and its voltage can be represented byvD2 = VO1 + [Laux(diD2/dt)]. Since the ascendant slope ofthe diode current (diD2/dt) is equal to zero at the mode 2,the voltage vD2 can be rearranged as vD2 = VO1 . Accord-ing to VO1 = 28 V, the voltage stress on the diode D2 issmaller than 100 V. Thus, a low-voltage Schottky diode alsocan be selected to reduce the conduction loss and the reverse-recovery current. The voltage across the diodes D3 and D4 ,which can be expressed by vD3,D4 = VO2 − (VC 1 + VFC) =[N/(N + 1)]VO2 , is 166.67 V by considering N = 5 and

VO2 = 200 V. Thus, the diodes (D3 and D4) with 200 V voltageratings can be selected.

When the main switch S1 is turned ON during modes1, 2, and 6, the voltage across the magnetizing inductor isvLmp = VFC = 12V. Moreover, the relation between the cur-rent and voltage of the magnetizing inductor can be representedas vLmp = LP (di/dt). If the current range di is designed as30 A and dt = d1TS = 6.4 μs, one can calculate the value ofLP as 2.56 μH. In order to manufacture the couple inductoreasily, the number of winding turns in the primary side of thecoupled inductor is N1 = 2, and its measured inductor valueis LP = 3 μH. Because the ratio of the primary and secondaryinductors in the coupled inductor is square proportional to theturns ratio (N = 5), the value of LS can be determined as 75 μH,and the winding turns in the secondary side of the coupled in-ductor is N2 = 10 in this study.

In this study, an EE-55 core with the magnetic flux density390 mT, the maximum magnetic flux 138 μWb, the cross sec-tion area 354 mm2 , and the designed air gap 1.5 mm is adoptedas the magnetic core of the coupled inductor. According to themagnetic circuit law [14], the air resistance can be calculatedas 3.4 MΩ. By considering the maximum output power 1.1 kWwith the corresponding conversion efficiency 85%, the maxi-mum input current is about 107.84 A, and the maximum mag-netizing current is about 122.84 A. The produced magnetic fluxcan be obtained as 72.25 μWb by using the primary windingturns (N1 = 2), maximum magnetizing current (122.84 A), andair resistance (3.4 MΩ). Thus, the saturation phenomenon of themagnetizing current can be prevented by the designed magneticcore of the coupled inductor.

In the proposed SIMO converter, the electric charge variationΔQ1 of the filter capacitor for the auxiliary circuit can be repre-sented as ΔQ1 = (VO1/RO1)(d1 − dx)TS = CO1ΔVO1 , andthe voltage ripple of VO1 can be rearranged as (ΔVO1/VO1) =(d1 − dx)/(RO1CO1fS ). By substituting d1 = 0.64, RO 1 =7.84 Ω, TS = 10 μs, and Laux = 2 μH into (17), the duty cy-cle dx can be calculated as 0.11. If one sets the voltage rip-ple of VO1 to be less than 1%, the value of CO1 should beselected over 67.6 μF by substituting d1 = 0.64, dx = 0.11,RO 1 = 7.84 Ω, fS = 100 kHz, and VO1 = 28 V into thefunction of CO1 = (d1 − dx)/[(RO1fS )(ΔVO1/VO1)]. More-over, the electric charge variation of the filter capacitor ΔQ2for the HVSC can be expressed as ΔQ2 = (VO2/RO2)d1TS =CO2ΔVO2 , and the ripple of the output voltage VO2 can be re-arranged as (ΔVO2/VO2) = d1/(RO2CO2fS ). By substitutingfS = 100 kHz, d1 = 0.64, RO 2 = 36.36 Ω, and VO2 = 200Vinto the function of CO2 = d1/[(RO2fS )(ΔVO2/VO2)], thevalue of CO2 should be chosen over 17.6 μF with the constrainton the output voltage ripple to be less than 1%. According tothe previous consideration, the values of CO1 = 100 μF andCO2 = 20 μF are adopted in the experimental prototype.

Due to a high-switching frequency (fS = 100 kHz) in theproposed SIMO converter, the factors of lower equivalent seriesresistance and faster dynamic response should be considered inthe design of the clamped capacitor C1 and the middle-voltagecapacitor C2 for reducing the capacitor voltage ripples. In thisstudy, metalized-polyester film capacitors are adopted for C1


and C2 for satisfying the fast charge and discharge property.In order to further minimize the current and voltage ripplesimposed to the main switch S1 and the diodes D3 and D4 , thecutoff frequencies of the LP − C1 and LS − C2 filters are takento be at least ten times smaller than the switching frequency[15]. According to the previous consideration, the values ofC1 and C2 are, respectively, chosen as 85 and 10μF in theproposed SIMO converter so that the corresponding resonantfrequencies are f01 ≡ 1/(2π

√LP C1) ∼= 9.97 kHz and f02 ≡

1/(2π√

LS C2) ∼= 5.81 kHz.In this study, the dc voltage feedback control is used to solve

the problem of the output voltage of the HVSC varied withload variations, and a digital-signal-processor TMS320F2812manufactured by Texas Instruments is adopted to achieve thisgoal of feedback control. In this feedback scheme, conven-tional PI control without detailed mathematical dynamic modelis utilized, and its formula can be represented as Vcom1 =kpVerr + ki

∫ t

0 Verr dt, where kp and ki are proportional andintegral gains, respectively; Verr = Vcmd − VO2 is the voltagetracking error, in which Vcmd is the voltage command. The valueof kp = 5 is designed according to the amount of initial track-ing error, and the value of ki = 0.05 is selected based on theamount of steady-state error. The driving signal T1 is generatedby comparing the control signal Vcom1 with the carrier wavevtri1 .

Remark 2: According to (19), the switch voltage vS1 can beeffectively clamped at a predetermined value if the values of theoutput voltage VO2 and the turns ratio N are fixed. As can beseen from Fig. 4(a) and (f), the action of the auxiliary circuit ishelpful to share the switch current iS1 . For the proposed SIMOconverter with larger output powers and more than two outputs,the increase of the input current will raise the current stress onthe main switch. Fortunately, one can use the parallel MOSFETto reduce the conduction loss of the main switch.

IV. EXPERIMENTAL RESULTS

A prototype with the following specifications is designed inthis section to verify the design procedure given in Section III.

Input power source VFC = 12 VAuxiliary output voltage VO1 = 24 – 28 VDesired output voltage Vcmd = 200 VSwitching frequency fS = 100 kHz;Coupled inductor LP = 3 μH; LS = 75 μH; N1 :

N2 = 2:10; k = 0.97; EE-55core;

Switch S1 : IRFP2907 (75 V/209 A × 2,RDS(on) = 4.5 mΩ); TO-247AC;

Diode D1 , D2 : Schottky diodeSR2060CT, (60 V/20 A × 2);D3 , D4 : Schottky Barrier Rec-tifiers MBR20200CT, (200 V/20 A × 2);

Auxiliary inductor Laux = 2 μH;

Capacitor C1 = 85 μF/100 V;C2 = 10 μF/250 V;CO1 = 100 μF/35 V;CO2 = 20 μF/250 V;

Weight 1.5 kg;Volume 2560 cm3 (16 cm × 16 cm ×

10 cm).Note that, a power supply with the electric specification of

15 V, 100 A, and 1.5 kW, manufactured by the Riye ElectricCompany, is taken as the input power source. Moreover, a batterywith the electric specification of 12 V and 60 Ah, manufacturedby the GS Battery Company, is adopted, and two batteries areconnected in series to take as the output load in the auxiliarycircuit. The proposed SIMO converter only has one input powersource, and the use of battery in the auxiliary circuit is just anexample for the load RO1 in Fig. 1, not a power-supply source forthe proposed SIMO converter. In real applications, the batterymodule in the auxiliary circuit can supply its power for otherperipheral circuits, e.g., the BOP including an electronic controlboard, an air compressor, and a cooling fan in an FC generationsystem.

The practical photograph of the entire experimental setup isdepicted in Fig. 7, where the front side and the reverse side ofthe converter prototype are given in Fig. 7(a) and (b), respec-tively; the entire experimental equipment is shown in Fig. 7(c).The experimental voltage and current responses of the proposedSIMO converter operating at 200 and 800 W output powersare depicted in Figs. 8 and 9, respectively. From Figs. 8(a) and9(a), the switch voltage vS1 is clamped at 37.5 V that is muchsmaller than the output voltage of the HVSC (VO 2 = 200 V),and the main switch turned ON under the condition of ZCS isobvious. As can be seen from Figs. 8(b) and 9(b), the primarycurrent iLkp and the secondary current iLs of the coupled in-ductor change by turns, and the charge and discharge currents ofthe auxiliary inductor iL aux are obvious. In Figs. 8(c)–(d) and9(c)–(d), the reverse-recovery currents can be removed owingto the selection of Schottky diodes for D1 and D2 . Accordingto Figs. 8(e)–(f) and 9(e)–(f), the voltages across the diodes D3and D4 are smaller than the output voltage of the HVSC (VO 2 =200 V), and the reverse-recovery currents can be alleviated bythe leakage inductor of the coupled inductor. Note that, the halfcycle oscillations in the measured waveforms of vS1 , vD1 , vD3 ,and vD4 with light load (200 W output power) are mainly causedby the resonant phenomena between the circuit stray inductanceand the parasitic capacitance of the power switch and diodes. Byobserving Figs. 8(g) and 9(g), the output voltage of the HVSCcan be stably adjusted by a conventional PI voltage controllerto be VO2 = 200 V when the voltage of the input power sourceis VFC = 12 V. Moreover, the output voltage of the auxiliarycircuit VO1 does not exceed the maximum floating charge volt-age of the battery module. The previous experimental resultsagree well with those obtained from theoretical analyses givenin Section II.

The experimental responses of the proposed SIMO con-verter under the conditions of output power variations and var-ied input voltages are given in Fig. 10. From Fig. 10(a), the


Fig. 7. Practical photograph of entire experimental setup: (a) front side of con-verter prototype; (b) reverse side of converter prototype; (c) entire experimentalequipment.

output voltage of the HVSC can be stably controlled to beVO2 = 200 V under the total output power variation between193 and 635 W; the output voltage of the auxiliary circuit VO1just changes slightly, and its voltage still does not exceed themaximum floating charge voltage of the battery module. Ascan be seen from Fig. 10(b) and (c), the output voltage of theHVSC (VO2 = 200 V) of the proposed SIMO converter operat-ing at 1-kW total output power is insensitive to the conditionof varied input voltages [i.e., the voltage of the input powersource changes from VFC = 12 V to VFC = 13 V in Fig. 10(b)or from VFC = 12 V to VFC = 11 V in Fig. 10(c)] owing to thePI closed-loop voltage control. The experimental conversion ef-ficiency of the proposed SIMO converter is depicted in Fig. 11.On the experimental system, the converter efficiency is evalu-ated via Power Analyzer PA4400 A equipment, manufacturedby the AVPower Company. The bandwidth of the PA4400 A isdc to 500 kHz, and the accuracy of the measured power is within±0.1%. From the experimental results, the maximum conver-sion efficiency of the proposed SIMO converter is over 95%, andits average conversion efficiency exceeds 91%. The objective ofhigh-efficiency SIMO power conversion can be verified. For ver-ifying the ability of the proposed SIMO converter with both re-sistive loads connected at the auxiliary circuit and the HVSC, theexperimental output voltages and currents due to the fixed resis-tance RO 1 = 6.5 Ω and the HVSC power change between 125 W(RO 2 = 320 Ω) and 750 W (RO 2 = 53.3 Ω) are given in Fig. 12.From Fig. 12, the output voltage of the HVSC still can be sta-bly controlled to be VO2 = 200V under the total output powervariation between 221 and 858 W. Moreover, the output voltageof the auxiliary circuit VO1 changes from 25 to 26.5 V underthe power delivered by the HVSC from 125 to 750 W, and itsvoltage still does not exceed the designed range between 24 and28 V.

In order to further examine the possibility of the exten-sion of two output ports to three output ports, the proposedSIMO framework in Fig. 1 can be reconfigured as Fig. 13.The major difference between Figs. 1 and 13 is the intro-duction of an additional auxiliary circuit composed of L′

aux ,D′

2 , C ′O1 , and R′

O1 into the original framework. The corre-sponding output voltage and current at the third output portare labeled as V ′

O1 and i′O1 in Fig. 13, respectively. In thisexperiment, a 12-V battery module is utilized for the out-put load in another auxiliary circuit, and its maximum bat-tery floating charge voltage is set at 14 V. It means thatthe voltages of three output ports are set at VO1 = 24 – 28 V,V ′

O1 = 12 – 14 V, and VO2 = 200 V when the voltage of the in-put power source is VFC = 12 V. The values of D′

2 and C ′O1

are the same as D2 and CO1 . Because the design of the aux-iliary inductor is related to the duty cycle dx , the value ofL′

aux is reselected as 4.5μH. The experimental responses ofthe proposed SIMO converter with three output ports are de-picted in Fig. 14. The voltages of the input power source andthree output ports are shown in Fig. 14(a), and their corre-sponding currents are given in Fig. 14(b). From Fig. 14(a)and (b), it is obvious that three output voltage levels satisfythe designed values. As can be seen from Fig. 14(c), the out-put voltage of the HVSC also can be stably controlled to


Fig. 8. Experimental voltage and current responses of SIMO converter with 200-W output power.

be VO2 = 200 V under the total output power variation be-tween 298 and 741 W owing to the PI closed-loop voltagecontrol. Thus, one can conclude that the output ports of theproposed SIMO converter can be extended to satisfy the de-signer’s requirement.

The performance comparisons of the proposed SIMOconverter with similar research in the announcedworks [11], [12], [16]–[19] are summarized in Table I.As can be seen from the tabular data, the circuit frameworksin previous works [12], [16]–[19] belong to the buck-typedc–dc converter. As far as our knowledge goes, few boost-typeSIMO converter design similar to the one in this study has

been reported. For the multi-output boost converter in [11],over two switches for one output were required, and thecorresponding experimental conversion efficiency was notgiven. The major difference between the proposed SIMOconverter and the converter in [11] is the application of theproposed SIMO converter to the requirement of multiplevoltage levels with one common ground, and the applicationof the converter in [11] to the active front of dc–ac multilevelinverters. Although the maximum conversion efficiency of theprevious multiple-output converter in [19] is slightly higherthan the one in the proposed SIMO converter, the amounts ofpower switches in [19] are over double than the requirement


Fig. 9. Experimental voltage and current responses of SIMO converter with 800-W output power.

in the proposed converter. It will result in the increase ofmanufacturing cost. Thus, the proposed SIMO converterindeed performs high-conversion efficiency under akilowatt-level output power with less power switchesthan other announced works [11], [12], [16]–[19].

In a standard boost converter, the voltage stress of the out-put diode should be equal to the output voltage, which is largein high-voltage applications. The price of the diode with high-voltage stress is rather higher than that of the diode with low-voltage stress. Besides, the reverse-recovery loss is significantdue to the use of a high-voltage-rating diode. For protectingthe energy source, the main inductor to be operated at CCM

is usually larger than the conventional one, so that the coilloss will be increased and it will limit the application with alarge input current. Because the clamped voltage of the diodeD1 is the same as the one of the main switch S1 in the pro-posed SIMO converter, a low-voltage Schottky diode can beadopted to conduct promptly with lower conduction loss andreverse-recovery current. Then, a low-voltage Schottky diodealso can be selected for the diode D2 to reduce the conductionloss and the reverse-recovery current because the voltage stressof the diode D2 is equal to the low-output voltage in the aux-iliary circuit. Moreover, the voltage across the diodes D3 andD4 , which can be expressed by vD3,D4 = [N/(N + 1)]VO2 , is


Fig. 10. Experimental input/output voltage and current responses of SIMO converter due to varied input voltage and output power. (a) Output power variationbetween 193 and 635 W. (b) Input voltage change from VFC = 12 V to VFC = 13 V with 1-kW output power. (c) Input voltage change from VFC = 12 V toVFC = 11 V with 1-kW output power.

Fig. 11. Conversion efficiency of SIMO converter for VFC = 12 V, VO 1 =24–28 V, and VO 2 = 200 V under different powers.

always lower than the output voltage VO2 in the HVSC. Asfor the coupled inductor in the proposed SIMO converter, itis designed with lower turns in the primary side to reduce thecoil loss, and works well for the application with a large in-put current. Thus, the limits on passive elements of a stan-dard boost converter can be relaxed in the proposed SIMOconverter.

Although the objective of a dual-output converter could beachieved by combining two traditional boost converters to fullyregulate the outputs, its manufacturing cost is inevitably in-creased, and the corresponding conversion efficiency is degen-erate due to the problem of reverse-recovery currents within the

Fig. 12. Experimental output voltages and currents due to fixed resistanceRO 1 = 6.5 Ω and HVSC power change between 125 W (RO 2 = 320 Ω) and750 W (RO 2 = 53.3 Ω).

Fig. 13. System configuration of high-efficiency SIMO converter with threeoutput ports.


Fig. 14. Experimental responses of SIMO converter with three output ports. (a) Input and output voltages with 885-W output power. (b) Input and output currentswith 885-W output power. (c) Output power variation between 298 and 741 W.

TABLE IPERFORMANCE COMPARISONS OF SIMO CONVERTER WITH OTHER

ANNOUNCED WORKS

output diodes, and large conduction and switching losses as thepower switches with high-voltage stress and hard switching. Inthis study, the proposed SIMO converter utilizes the propertyof the coupled inductor to make the switch working at ZCS, toraise the voltage gain, and to alleviate the reverse-recovery cur-rent, so that higher conversion efficiency and voltage gain canbe obtained in comparison with the combination of two tradi-tional boost converters. Moreover, it is easy to extend the outputterminals of the proposed SIMO converter by adding auxiliarycircuits, not a complete structure of traditional boost convertercircuit. As long as the well design of auxiliary inductors ac-cording to (18) and Fig. 6, the regulation of output voltagesalso can be achieved. As a result, the functions of the proposedSIMO converter with high-conversion efficiency, high-voltage

gain and low-manufacturing cost are superior to the combinationof traditional single-input single-output converters.

V. CONCLUSION

This study has successfully developed a high-efficiencySIMO dc–dc converter, and this coupled-inductor-based con-verter was applied well to a single-input power source plustwo output terminals composed of an auxiliary battery moduleand a high-voltage dc bus. The experimental results reveal thatthe maximum efficiency was measured to exceed 95%, and theaverage conversion efficiency was measured over 91%. The pro-posed SIMO converter is suitable for the application requiredone common ground, which is preferred in most applications.However, it is not appropriate to be used as the active frontfor dc–ac multilevel inverters. This limitation is worthy to beinvestigated in the future research.

The major scientific contributions of the proposed SIMO con-verter are recited as follows: 1) this topology adopts only onepower switch to achieve the objective of high-efficiency SIMOpower conversion; 2) the voltage gain can be substantially in-creased by using a coupled inductor; 3) the stray energy canbe recycled by a clamped capacitor into the auxiliary batterymodule or high-voltage dc bus to ensure the property of voltageclamping; 4) an auxiliary inductor is designed for providing thecharge power to the auxiliary battery module and assisting theswitch turned ON under the condition of ZCS; 5) the switchvoltage stress is not related to the input voltage so that it is moresuitable for a dc power conversion mechanism with differentinput voltage levels; and 6) the copper loss in the magnetic corecan be greatly reduced as a full copper film with lower turns. This


high-efficiency SIMO converter topology provides design-ers with an alternative choice for boosting a low-voltagepower source to multiple outputs with different voltage levelsefficiently. The auxiliary battery module used in this study alsocan be extended easily to other dc loads, even for differentvoltage demands, via the manipulation of circuit componentsdesign.

ACKNOWLEDGMENT

The authors would like to thank the Referees and theAssociate Editor for their valuable comments and helpfulsuggestions.

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Rong-Jong Wai (M’99–SM’05) was born in Tainan,Taiwan, in 1974. He received the B.S. degree in elec-trical engineering and the Ph.D. degree in electronicengineering from Chung Yuan Christian University,Chung Li, Taiwan, in 1996 and 1999, respectively.

Since 1999, he has been with Yuan Ze University,Chung Li, Taiwan, where he is currently a Yuan-ZeChair Professor with the Department of Electrical En-gineering, the Dean of the Office of General Affairs,the Chief of the Environmental Protection and Sani-tation Office, and the Director of the Electric Control

and System Engineering Laboratory. He is a chapter-author of Intelligent Adap-tive Control: Industrial Applications in the Applied Computational IntelligenceSet (Boca Raton, FL: CRC Press, 1998) and the coauthor of Drive and In-telligent Control of Ultrasonic Motor (Tai-chung, Taiwan: Tsang-Hai, 1999),Electric Control (Tai-chung, Taiwan: Tsang-Hai, 2002), and Fuel Cell: NewGeneration Energy (Tai-chung, Taiwan: Tsang-Hai, 2004). He has 30 inventivepatents and has authored more than 130 conference papers and 140 internationaljournal papers. His biography was listed in Who’s Who in Science and Engineer-ing (Marquis Who’s Who) in 2004–2011, Who’s Who (Marquis Who’s Who)in 2004–2011, and Leading Scientists of the World (International Biographi-cal Centre) in 2005, Who’s Who in Asia (Marquis Who’s Who), Who’s Who ofEmerging Leaders (Marquis Who’s Who) in 2006–2011, and Asia/Pacific Who’sWho (Rifacimento International) in volumes VII–XI. His research interests in-clude power electronics, motor servo drives, mechatronics, energy technology,and control theory applications. The outstanding achievement of his researchis for contributions to real-time intelligent control in practical applications andhigh-efficiency power converters in energy technology.

Dr. Wai received the Excellent Research Award in 2000, and the Wu Ta-You Medal and Young Researcher Award in 2003 from the National ScienceCouncil, Taiwan. In addition, he was the recipient of the Outstanding ResearchAward in 2003 and 2007 from the Yuan Ze University, Taiwan; the ExcellentYoung Electrical Engineering Award and the Outstanding Electrical Engineer-ing Professor Award in 2004 and 2010 from the Chinese Electrical EngineeringSociety, Taiwan; the Outstanding Professor Award in 2004 and 2008 from theFar Eastern Y. Z. Hsu–Science and Technology Memorial Foundation, Taiwan;the International Professional of the Year Award in 2005 from the InternationalBiographical Centre, U.K.; the Young Automatic Control Engineering Award in2005 from the Chinese Automatic Control Society, Taiwan; the Yuan-Ze ChairProfessor Award in 2007 and 2010 from the Far Eastern Y. Z. Hsu–Science andTechnology Memorial Foundation, Taiwan; the Electric Category-Invent SilverMetal Award in 2007, the Electronic Category-Invent Gold and Silver MetalAwards in 2008, the Environmental Protection Category-Invent Gold MetalAward in 2008, and the Most Environmental Friendly Award in 2008 from theInternational Invention Show and Technomart, Taipei, Taiwan; the UniversityIndustrial Economic Contribution Award in 2010 from the Ministry of Eco-nomic Affairs, Taiwan.

Kun-Huai Jheng was born in Changhua, Taiwan,in 1986. He received the B.S. degree in electricalengineering from the National Yunlin University ofScience and Technology, Yunlin, Taiwan, 2009, andthe M.S. degree in electrical engineering from YuanZe University, Chung Li, Taiwan, in 2012.

His research interests include battery module andpower electronics.

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